(navigation image)
Home American Libraries | Canadian Libraries | Universal Library | Community Texts | Project Gutenberg | Children's Library | Biodiversity Heritage Library | Additional Collections
Search: Advanced Search
Anonymous User (login or join us)
Upload
See other formats

Full text of "TM 11-666 Antennas And Radio Propagation"

DEPARTMENT OF THE ARMY TECHNICAL MANUAL 

T M 11-666 



ANTENNAS 
AND 
RADIO 
PROPAGATION 




DEPARTMENT OF THE ARMY FEBRUARY 1953 



United States Government Printing Office 
Washington : 1953 



DEPARTMENT OF THE ARMY 
Washington 25, I). C, 9 February 1953 
TM 11-666 is published for the information and guidance of ail concerned. 
[AG 413.44 (20 Oct 52)] 

By order of the Secretary of the Army: 

Official: ,1. LAWTOX COLLIN'S 

WM. E. BERGIN ( 'kit f of Staff , United States Army 
Major General, USA 
The Adjutant General 

Distribution: 

Active Army: 

Tech Svc (1); Tech Svc Bd (i): AFF B<i (ea Svc Tost Sec) (1); 
AFF (5); AA Comd (2); OS Maj Comd (:>)• Base Comd (,'{); MDW 
(5); Log Comd (2); A (10); CHQ (2), FT (2); I'SMA (8); C&GSC 
(3); PMS&T 11 (1); Gen Dep (2); Dep 1 1 (20); RTC (2) except 11 
(100); POE (1), OSD (3); Lab 1 I :2>; 4th & 5th Eeh Maint Shops 
11 (2); Mil Dist (1); T/O & E's, 1 1-7N (2); I 1 15N (2). 

NG: Same as Active Army. 

Army Reserve: Same as Active Army. 

For explanation of distribution formula, see SR 310 90 1. 



ELECTRONIC FUNDAMENTALS SERIES 



The manuals on electronic fundamentals form a progressive series of 
educational texts which present the theory and application of electronics 
for the military services. The series starts with the basic electrical funda- 
mentals and extends to the most recent technical concepts, as applied to 
telegraphy, telephony, radio, loran, facsimile, radio direction finding, radar, 
meteorological radio, television, and other military equipments. See SR 
310-20-4 for published, available manuals in this series. 



Hi 



CONTENTS 



CHAPTER 1. INTRODUCTION Paragraph, Pagt 

Section I. The electromagnetic wave 1-6 1 

II. Wave propagation 7-9 10 

77 J. Summary and review questions 10, 11 14, 15 

CHAPTER 2. MODES OF PROPAGATION 

Section I. Ground-wave propagation 12-18 17 

77. The ionosphere 19-25 24 

777. Sky-wave propagation 26-33 37 

IV. Summary and review questions 34,35 51,53 

CHAPTER 3. HALF-WAVE AND QUARTER-WAVE ANTENNAS 

Section I. Basic theory _ .. 36-43 54 

77. Transmission lines 44-46 67 

777. Basic feeder systems 47-51 79 

IV. Basic radiation patterns 52-55 89 

V. Practical half-wave antennas 56-68 97 

VI. Grounded antennas 69-79 118 

VII. Summary and review questions 80, 81 135, 

137 

CHAPTER 4. LONG-WIRE ANTENNAS 82-93 139 

5. DRIVEN AND PARASITIC ARRAYS 

Section I. Introduction 94-97 174 

II. Driven arrays 98-104 177 

777. Parasitic arrays 105-107 198 

IV. Summary and review questions 108, 109 208, 

209 

CHAPTER 6. RADIO DIRECTION FINDING ANTENNAS- 110-115 211 

INDEX 221 




F igure 1 . Assorted antennas and their uses. 



CHAPTER I 
INTRODUCTION 



Section I. THE ELECTROMAGNETIC WAVE 



1. G 



enera 



a. The study of antennas and wave propagation 
is essential to a complete understanding o: ; radio 
communication, radar, loran, and other electronic 
systems. In such systems, energy in the form of 
radio or electromagnetic waves is generated by 
electronic equipment and led to an anterna by 
means of special transmissio n lines. The antenna 
radiates this energy out into space at the speed of 
light (approximately 186,030 miles per second). 
Receiving antennas, when placed in the path of 
the traveling wave, absorb part of this energy 



the power of the transmitter, the distance between 
transmitter and receiver, and the sensitivity of 
the receiver. The ability of the earth's atmos- 
phere to conduct the energy to its destination, 
together with the nature of the terrain between 
the sending and the receiving points, may, however, 
be responsible for the frequency selected; inter- 
fering signals may make reception impossible at 
a desired time, and the amount of noise present 
and transmission line losses ma y combine to make 
an otherwise good signal unintelligible. To under- 
stand the proper importance of all these factors it 
is necessary first to investigate the nature of the 



MEDIUM 
[EARTH'S ATMOSPHERE) 



PORTION or RADIO W«VE 
INTERCEPTED BY RECEIVING 
ANTENNA 



TRANSMITTING ' 
ANTENNA 




TRANSMISSION 
LINE 



TRANSMITTER 



— — TRANSMISSION 

LINE 



Figure 2. Simple radio communications network. 



and send it to the receiving equipment by means 
of a transmission line. Thus, the components 
required for successful transmission of intelligence 
by means of radio waves are a transmitter and 
transmission line, a transmitting antenna, a 
medium through which the waves travel (for 
example, the atmosphere surrounding the earth), 
a receiving antenna, another transmission line, 
and suitable receiving equipment. Figure 2 is a 
block diagram showing the arrangement of these 
components. 

b. The ability to obtain successful communica- 
tion by means of radio waves depends chiefly on 



radio wave and the factors affecting its successful 
propagation. 

2. Wave Motion 

a. Heinrich Hertz, in 1887, demonstrated that 
electromagnetic energy could be sent out int/ft 
space in the form of radio waves. It is known, 
of course, that an induction field exists about any 
wire carrying an electric current. Another field, 
called the radiation field, exists, becomes detached 
from the wire, and travels through space to make 
radio communication possible. Attempts have 



1 



been made to illustrate the action involved by 
comparing it to the waves formed by dropping a 
stone on the smooth surface of a pond. Although 
the analogy is not exact, it does serve a useful 
purpose because it makes comparison with a well 
known physical action. Figure 3 shows such an 
event taking place, the waves appearing to be a 
series of regularly alternating crests and hollows 
moving radially outward in all directions from the 
point of disturbance on the surface of the water. 

b. Figure 4 presents a graphical analysis of this 
action, showing how the stone imparts its energy 
to the water surface. A of figure 4 shows the 
falling stone just an instant before it strikes the 
water. Its energy has been derived from the 
gravitational pull exerted on it by the earth, and 
the amount of this energy depends not only on the 
weight of the stone but also on the height from 
which it has been dropped. B of figure 4 shows 
the action taking place at the instant the stone 
strikes the surface, pushing the water around it 
upward and outward, thereby imparting an initial 
velocity to the mass of water at this point of con- 
tact. In C of figure 4 the stone has sunk deeper 
into the water, which closes violently over it, 
causing some spray, while the leading wave has 
moved radially outward because of its initial 
velocity. An instant later, the stone will have 
sunk out of sight, leaving the water agitated 



(D of fig. 4) . Here the leading wave has continued 
to move outward, and is followed by a series of 
waves of gradually diminishing amplitude; mean- 
while, the agitation in the immediate vicinity of 
the original point of contact gradually subsides. 
Note that the leading wave has amplitude (note 
A, fig. 4) and wavelength (note B, fig. 4) corre- 
sponding to 1 complete cycle. 

c. Of course, this action fails to compare with 
that of electromagnetic radiation, since a con- 
tinuous wave motion is not imparted to the surface 
of the water by a dropped stone,' the action just 
described being that of a damped oscillation. 
But suppose a string bo be attached to the stone 
of figure 4 so that its upward and downward 
motion caa be controlled from above. Before the 
event shown in C of figure 4, takes place, the stone 
is pulled sharply upward after its initial fall to the 
position shown in .,4 of figure 5, and then lowered 
again to the position shown in J? of figure 5. 
The result of these upward and downward motions 
is to reinforce the diminishing amplitude of the 
succeeding waves, and, if the timing and the down- 
ward force of the 3(;one are exactly right, waves of 
constant amplitude continue to travel outward 
from the disturbing source at a velocity, v, 
determined by the product of the frequency, /, 
of the downward motion (those that impart 
energy to the medium) multiplied by the measured 




TL 14431 S . 



Figure S. Formation of waves in mooter. 



2 



FALLING OBJECT 



SURFACE OF WATER 



A 



L EADING WAVE FORMS AT INSTANT 




POINT Of" ORIGINAL DISTURBANCE 




NOTES: A AMPLITUDE OF LEADING WAVE. 

B CORRESPONDS TO 1 CYCLE OF OSCILLATION. 



Figure 4. How a falling stone im parts wave motion to a water surface. 



wavelength X (lambda), of the resulting waves. 
(Since L is used conventionally as the sjonbol for 
inductance and I for dimensional length, the 
equivalent Greek letter, X, is used for the length 
of waves.) Thus: v—\j. If the frequency and 
the velocity of propagation are known, then the 
wavelength may be determined: 

\=v/j 

d. This same type of action takes pla,ce at the 
antenna of a radio transmitter, the medium in this 
case being the free space about the antenna 
instead of a water surface, and the disturbing 
source a fluctuating induction field in place of a 
moving stone. The preceding formulas (c above) 
hold for this wave motion and for all types of 
wave motion, whether it be of water waves, sound 
waves in air, or light and electromagnetic waves 
in free space. It should be noted, however, that 
the term free space is used to denote the unob- 
structed medium through which radio waves 
travel. Free space implies that the source (trans- 
mitter and antenna in the case of radio waves) is 
surrounded by nothing except ordinary air or 




TM 066-3 

Figure 5. Formation of continuous waves in water. 

3 



vacuum. The presence of trees, hills, lakes, or 
other aspects of the local terrain modifies in each 
case the effective radiation of an electromagnetic 
wave. And thus, although the surface of the 
earth, or the air near its surface, is not considered 
to be free space, the formula in c above still will 
hold in terms of properly modified factors. 

e. Like light and radiant heat, radio waves 
are a form of radiant energy propagated through 
space at a speed of nearly 300,000,000 meters 
(186,000 miles) per second. Thus, a wave alternat- 
ing at a frequency of 1,000,000 cycles per second 
has a wavelength of approximately 982 feet. 

Since: X ==»//= 186,000/1,000,000 

X=0.186 mile 
Converting to feet: X=0.186 x 5,280 

X = 982 feet. 

Then one half-wavelength or X/2 = 982/2 =491 feet. 
3. Radiation from a Half-wave Antenna 

a. Although nothing has been said about the 
characteristics of a half-wave antenna, it is con- 
venient to use this element in describing the 
mechanism of radiation. Simply stated, a hal:'- 
wave antenna is one which is approximately a 
half-wave long at the operating frequency. For 
example, at 30 megacycles, a half-wave antenna :s 
approximately 16 feet long. When power is 
delivered to such an antenna, two fields are set 
up by the fluctuating energy: one the induction 
field, which is associated with the stored energy 
(par. 6), and the other the radiation field, which 
moves out into space at nearly the speed of lighl,. 
At the antenna, the intensities of these fields are 
large and are proportional to the amount of power 
delivered to the antenna. At a short distance 
from the antenna, and beyond, only the radiation 
field prevails. This radiation field is made ud 
of an electric component and a magnetic compo- 
nent at right angles to each other in space and 
varying together in intensity. 

b. Figure 6 shows the manner in which the 
radiation field is propagated away from the 
antenna. The electric and magnetic components 
are represented here by separate sets of flux lines, 
which are at right angles to each other and to the 
radial direction of propagation. The picture is 
representative of any plane containing the antenna, 
and applies for only a single instant of time. The 
magnetic flux lines are shown as circular lines 
having the axis of the antenna as their axis, so 
that they appear in the illustration as dots and 



crosses. The electric flux lines also are closed, or 
endless, lines in the present case, and consist 
essentially of arcs of circles lying in planes con- 
taining the antenna and joined in the manner 
shown. These electric flux lines reverse direction 
at precisely the places where the magnetic flux 
lines reverse, and their density varies along the 
radial direction in the way that the magnetic flux 
density varies. In fact, the electric flux density 
is everywhere proportional to the magnetic flux 
density. 

c. The correct configuration of flux lines for a 
single instant of time is shown in figure 6. As 
time passes, these flux lines expand radially with 
the velocity of light, and new flux lines are created 
at the antenna to replace those that travel 
outward. Thu"«, oscillating electric and magnetic 
fields are produced along the path of travel. The 
frequency of the oscillating fields is the same as the 
frequency of the antenna current, and the magni- 
tudes of both fields vary continuously with this 
current. The variations in the magnitude of the 
electric component (called the El field) and those 
of the magnetic component (called the H field) 
are in time phase, so that at every point in space 
the time-varying magnetic field induces a differ- 
ence in voltage, which is the electric field. The 
electric field also varies with time, and its varia- 
tion is equivalent to a current, even though it is 
not associated with a movement of charge. This 
is called the displacement current, and it estab- 
lishes a magnetic field in precisely the way that a 
conduction current does. Thus, the varying mag- 
netic Jield produces a varying electric yield, and the 
varying electric yield, through its associated dis- 
placement current, sustains the varying magnetic 
yield. Each yield supports the other, and neither 
can be propagated by itself, without setting up the 
other. 

d. The mechanism by which the flux lines of the 
radiation field become separated from the antenna 
and are radiated out into space can be understood 
by considering the movement of the charges which 
pass back and forth along the length of the antenna 
as a result of the driving current. At an instant 
of time when positive charges are distributed 
along one half of the antenna, and negative 
charges along the other half, electric flux lines 
originate on the positive charges and terminate 
on the negative charges. These flux lines follow 
paths in space such as those indicated in A of 
figure 7. As time passes, the separated charges 
again come together, bringing the two ends of the 



4 




Figure 6. Electric and magnetic fields about an antenna. 



flux lines together, as in B of figure 7. When the 
unlike charges meet, they seem to cancel each 
other, and the flux lines attached to them collapse 
and cease to exist. Thus, the two ends of the 



flux lines become joined, creating closed lines 
which are snapped free from the antenna and 
propagated outward, as shown in C and D of 
figure 7. 





c 



o o 








Figure 7. Creation of closed electric lines at an antenna. 



5 




6 



4. TH« Wave Front 

Actually, the radiation field may be conceived 
of as a series of concentric expanding surfaces 
following one after the other. On dach of these 
surfaces, the radiation field is at a constant phase 
of vibration and is called a wave front. Figure 
3 illustrates this concept. If the surface of the 
water about the point of disturbance is examined, 
water waves are observed to be radiating outward, 
increasing in circumference as they travel. At 
a distance from the origin, the circular pattern 
becomes less apparent and, when a small trans- 
verse section of the disturbed surface is studied, 
only arcs of relatively large circles of the wave 
front appear, and these seem to be transverse 
straight lines. Similar effects occur with radio 
waves that are radiated from an antenna, so that, 
at great distances from the antenna, a small por- 
tion of the wave front may be taken to be a 
transverse plane surface (fig. 8). From the point 
of view of the observer, the wave fronts march 
past, varying sinusoidally in direction and mag- 
nitude, as shown by the graphs at the top of the 
picture. 



sounce 

TRANSMITTING 

ANTENNA 
' (VERTICAL) 



SIGNAL VOLTAGE 




ELECTRIC HELD 
COMPONENT 



RECEIVING 
"ANTENNA 



PLANE 
TRANSVERSE 
SURMCE 

MAGNETIC / \ 

FIELD 
COMPONENT 



Figure 9. Components of electromagnetic wave. 



a. E and H Fields. In considering the radio 
signal path between a transmitting and a receiving 
antenna, the concept of a moving wave front 
becomes important. This path is assumed to 
follow a straight line in the direction of travel. In 
traveling this way, the radio wave may be 
described as moving electromagnetic field, having 
velocity in the direction of travel, and with com- 
ponents of electric intensity and magnetic inten- 
sity arranged at right angles to each other (fig. 9). 
This figure indicates that the instantaneous ampli- 
tudes and phase relations of the electric field, E, 
and the magnetic field, H, change with time 
according to the frequency of the wave. In order 
to visualize this action, imagine a receiving antenna 
in the path of the oncoming wave (fig. 10). Now 
consider that the entire wave is moving at a con- 
stant speed in the direction indicated. The 
intensities of both the electric and the magnetic 
fields will be maximum at the same instant as the 
crest of the wave passes the antenna, and mini- 
mum at the same instant the zero point is reached, 
but at all times the fields are perpendicular to each 
other. It should be kept in mind that figure 10 
shows only one transverse section of the entire 
wave front, which fills all the space shown in the 
figure. 

b. Directional Conventions. The direction of the 
electric or magnetic component of an electromag- 
netic wave at any instant is determined by the 
conventions described below. It should be noted, 
however, that the electric and magnetic compo- 
nents of a radiated field are in phase in respect to 
time and 90° out of phase in respect to space, but 
that the electric and magnetic components of the 
induction field are 90° out of phase in respect to 
both time and space. Therefore, for the duration 



DECEIVING ANTENNA • 




Figure 10. E and H fields of radiated wave. 



7 



of a given electric induction field, the magnetic 
induction field is first in one direction at right 
angles to the increasing electric field and then in 
the opposite direction for the decreasing electric 
field. 

(1) Electric Jield. The direction of any elec- 
tric field is, like that of voltage, taken to 
be from plus to minus; that is, an arrow 
representing the direction of an electric 
field carries the plus sign at its tail and 
the minus sign at its head. This con- 
vention is based on the arbitrary selec- 
tion of a unit positive charge as a test 
charge, which in such a field would move 
from plus to minus. 

(2) Magnetic field (radiated). If the radiated 
field is moving away from an observer 
with the arrows indicating direction of 
the electric field pointing downward, the 
direction of the magnetic field is given by 
arrows pointing to the left. This con- 
vention is based on the generation of & 
magnetic field by current defined as 
electron flow — that is, moving negative 
charges — and determined by use of the 
left-hand rule. 

(3) Magnetic field (induction). To an observ- 
er looking outward from a transmitting 
antenna, an increasing electric field of 
downward direction would be accompa- 
nied by a magnetic field with arrows 
pointing to the left, and a decreasing 
electric field of downward direction 
would be accompanied by a magnetic 
field with arrows pointing to the right. 
By the same token, the electric induction 
field reverses direction for the duration 
of a given magnetic field. 

5. Polarization 

In describing the principal characteristics of ft 
wave front, the electric field component is taken 
as the point of reference. For example, the inten - 
sity of a radio wave usually is measured in terms of 
the strength of the electric field, and the orienta- 
tion of the wave in space usually is described in 
terms of the direction of wave travel and by the 
direction of the electric field. The direction of 
the electric field, in particular, determines the: 
polarization of the wave. Thus, as shown in (X) of 
figure 11, when the plane containing the electric: 
field and the direction of travel is vertical, the 



wave is said to be polarized vertically; if the plane 
is horizontal, as in <§) of figure 11 the wave is said 
to be polarized horizontally. This polarization of 
the wave front is an important factor in the 
efficient transmission and reception of signals. 
Thus, if a single-wire antenna is used to extract 
energy from a passing radio wave, maximum pick- 
up will result when the antenna is so placed 
physically that it lies in the same direction as the 
electric field component. Therefore, a vertical 
antenna (one perpendicular to the ground) should 
be used for the efficient reception of vertically 
polarized waves (those transmitted from a verti- 
cal antenna), and a horizontal antenna should be 
used for the reception of horizontally polarized 
waves (those transmitted from a horizontal an- 
tenna). In both cases, the direction of travel is 
taken to be parallel to the earth's surface. 

6. Field Intensity 

a. The conventional measure of the field inten- 
sity of a radiated wave is, as mentioned in the 
previous paragraph, a measure of the intensity of 
the electric field. This intensity usually is ex- 
pressed in volts, millivolts, or microvolts per 
meter, which is a measure of the dielectric stress 
produced by the electric field, or the voltage 
induced in a conductor 1 meter long when held so 
that it lies in the direction of the electric field and 
is at right angles to the direction of propagation 
and to the direction of the magnetic field. As the 
distance between the transmitting antenna and 
the receiving antenna becomes greater and greater, 
the field intensity of the radiated wave falls off 
proportionately to the distance. For example, if 
the received field intensity of a signal is 50 milli- 
volts per meter at a distance of 25 miles, then at 
50 miles, or twice the distance, the field intensity 
is one-half as much, or 25 millivolts. Thus, 
electric field intensity varies inversely with the 
distance from the transmitting antenna. 

b. In order to understand this variation of the 
field intensity of a radiated wave, consideration 
must be given to the relationship between the 
power radiated and the field intensity. As was 
known before the era of electronics, the power 
of a radiated wave, such as a light wave, falls off 
as the square, of the distance between the source of 
light and the point of measurement. Thus, if a 
lighted candle is placed in the center of a dark 
room, the intensity of its light will vary accord- 
ing to the square of the distance from it. In other 



8 



DIRECTION OF TRAVEL 




VERTICAL POLARIZATION 



DIRECTION OF TRAVEL 




(5) HORIZONTAL POLARIZATION 

Figure 11. Vertical and horizontal polarization. 



words, a sheet of paper held perpendicular to the 
rays of a candle and a certain distance from it will 
be lighted four times as brightly as one held at 
twice this same distance. The reason for this is 
illustrated in figure 12, where it may be seen that 
the same amount of light that is intercepted by the 
paper at A must be used to cover four times as 
great an area at B, which is twice as far from the 



candle as A. This law is expressed by the 
formula: 

A—4irr 2 

where A is the area of any portion of the surface of 
a sphere of radius, r, having its center at the 
source of the light. If r in figure 12 is equal to 1 
foot, the area of the sheet at A is then 4ir (l) 2 or 



9 



4*-, whereas the area of the sheet at B is equal to 
4x (2) a or 16*-. The amount of light intercepted 
it A is equivalent to the amount intercepted at B, 
which has an area four times as great. This same 
law holds for the field intensity of electromagnetic 
waves when the power intercepted in a unit area 
is considered. However, since electric power ex- 
pressed in terms of the voltage present is propor- 
tional to. E* (because P=E 2 jR), then the square 
of Che voltage falls off as the square of the distance, 
or voltage itself falls off as the distance. 




2, » 

Tuees-7 

Figure IS. How light waves decrease in intensity as the 
square of the distance. 



e. This analysis of field strength and the man- 
ner in which it decreases at a distance from the 
radiating antenna supplies the basis for under- 
standing the relationship between the various 
fields and their components present at a trans- 
mitting antenna. As mentioned in paragraph 3a, 
too major fields are set up about a transmitting 
antenna when alternating current power is applied. 
The radiation field persists to great distances , and 
the manner in which the power and intensi ty of 



this field are related to distance has been described 
above. The induction field, on the other hand, 
is present only in the immediate area of the trans- 
mitting antenna, since its power falls off as the 
fourth power of the distance. Close by the an- 
tenna, the induction field is stronger than the 
radiation field, but, at a distance equal to a few 
wavelengths of the frequency transmitted, the 
induction field is so small as to be negligible. The 
induction field is the field usually associated with 
the voltage and current in any conductor, and as 
such not only is the electric component at right 
angles to the magnetic component (as it is about 
any inductance carrying alternating current), but 
also the usual phase shift of 90° between voltage 
and current prevails. The result is that the in- 
duction field is made up of two fields separated 
in space by 90° and in time by 90°, or (as some- 
times expressed) two fields in space quadrature and 
in time quadrature. The electric and magnetic 
components of the radiation field, as has been 
shown, are also in space quadrature but are in 
time phase. 

d. Perhaps some notion of the characteristics 
of the induction field may be gained from con- 
sidering that its components are the stored energy 
usually associated with a resonant circuit — the 
energy stored alternately by an inductor and a 
capacitor. Thus, the energy in the electrostatic 
field is the energy in the magnetic field (oscillating 
continuously from one to the other), and this 
energy in either case is returned to the antenna 
circuit. Therefore, the power in the radiation 
field is the power delivered to a transmitting an- 
tenna, less the heat loss in the antenna itself. 



Section II. WAVE PROPAGATION 



7. The Atmosphere 

The study of wave propagation is concerned 
chiefly with the properties and effects of the actual 
medium through which radio waves must travel 
in following any given path between a transmit- 
ting antenna and a receiving antenna. Since the 
atmosphere about the earth is not uniform, chang- 
ing with a change in height or geographical lo- 
cation, or even with a change of time (day, night, 
season, year), the lack of uniformity appreciably 
influences the passage of electromagnetic waves 
through it, thereby adding many new factors to 
complicate what at first might seem to be a rel- 
atively simple communication problem. A knowl- 



edge of the composition of the earth's atmosphere 
is extremely important in solving this problem, 
and therefore, for purposes of understanding wave 
propagation, various layers of the atmosphere 
have been distinguished. These are the tropo- 
sphere, the stratosphere, and the ionosphere. Their 
relative positions are shown in figure 13. 

a. The Troposphere. The troposphere is that 
portion of the earth's atmosphere extending from 
the surface of the earth to heights of about 1 1 km 
(kilometers), or approximately 61 H miles. The 
temperature in this region varies appreciably with 
altitude. 

b. The Stratosphere. The stratosphere is that 
portion of the earth's atmosphere lying between 



10 



IhIigh 

I IN KILO- 

I METERSlH^OSMIC RAYS 

350) 



OORPUSCLESMMCORPUSCULAR RAIHATlON [ 
ELECTPONSI 



'eight 

IN 
MILES 



I TEMPERATURE! 



[BAROMETRIC] 

I PRESSURE 



20C 



NIGHTTIME SHORT-WAVE 
REFLECTION F2 REGION 




* - _ -40° C 
: 25 - -55°C 

r — -so°c 
sea' +,0 ° c 

LEVEL 



[BASALTIC layer] 

Figure 13. Layers of the earth's atmosphere. 



1P§ 

TM 666-188 Jl 



11 



the troposphere and the ionosphere. The tem- 
perature in this region is considered to be almost 
constant, and, hence, it is called the isothermal 
region. 

c. The Ionosphere. Besides the usual varia- 
tions of moisture content and temperature, and 
the variations in density associated with a change 
in elevation, the atmosphere is distinguished 
principally by the variation in amount of ioniza- 
tion present. The ionosphere accordingly is de- 
fined as that portion of the earth's atmosphere 
above the lowest level at which ionization affects 
the transmission of radio waves. The ionization 
of this layer is large compared with that near the 
surface of the earth. It extends from about 50 
kilometers to 250 kilometers above the earth. 
The ionosphere is itself composed of several layeni, 
as shown in figure 13, where ionization occurs at 
different levels and intensities. The properties 
of these layers and their effects upon the propa- 
gation of electromagnetic waves are treated ii 
detail in the next chapter. 

8. Frequency Classifications 

An understanding of the effects of the atmos- 
pheric layers described above is complicated 
further by variations in the frequency of the 
transmitted wave. The characteristics of low- 
frequency propagation are different from those of 
high-frequency propagation, and therefore, for 
ease of identification, the frequencies of propaga- 
tion usually are classed in ranges, as shown in 
table I. The choice of a given frequency as the 
point of division between classes, such as thai 
between the very-high frequencies and the ultra- 
high frequencies, is more or less arbitrary and is 
agreed upon for convenience. Note that in thio 
manual only those radio frequencies below SO mc 



Table I. Radio-Frequency Clatsificationt 



Freqtwaor (mc) 


Description 


Abbrevia- 
tion 


Below .03 . 


Very-low frequency 


vlf 


.03 to .3 


Low frequency 


If 


.3 to 3.0 


Medium frequency 


mf 


3.0 to 30 


High frequency 


of 


30 to 300 


Very-high frequency 


vhf 


300 to 3,000 


Ultrahigh frequency 


uhf 


3,000 to 30,000 


Superhigh frequency 


Hhf 


30,000 to 300,000 


Extremely-high fre- 


ehf 




quency. 





(megacycles) per satond are considered. The special 
considerations involved in the use of higher fre- 
quencies may be found in TM 11-667, Higher- 
frequency Techniques, and in TM 11-673, Gen- 
eration and Transmission of Microwave Energy. 

9. Propagation in the Atmosphere 

There are two principal ways in which radio 
waves travel from a transmitter to a receiver: by 
means of ground waves which travel directly from 
the transmitter to the receiver, and by means of 
sky waves which travel up to the electrically 
conducting layers of the earth's atmosphere (the 
ionosphere), and are reflected by them back to 
the earth. Long-distance radio transmission takes 
place principally by means of sky waves, and short- 
distance transmission and all ultrahigh-frequency 
transmission take place by means of ground waves. 
Some forms of transmission consist of combina- 
tions of both. The propagation of the ground 
wave is in part affected by the electrical charac- 
teristics of the earth (soil or sea), and by diffraction, 
or bending, of the wave with the curvature of the 
earth. These characteristics differ in different 
localities, but under most conditions they are 
practically constant with time. Sky-wave propa- 
gation, on the other hand, is variable, since the 
state of the ionosphere is always changing and 
this, therefore, affects the reflection, or the refrac- 
tion, of the waves. 

a, Reflection. 

(1) The reflection of a radio wave is like that 
of any other type of wave. For instance, 
when a beam of light falls on the surface 
of a mirror, nearly all of it is turned back 
or reflected. As with light waves, the 
efficiency with which reflection of radio 
waves occurs depends on the material of 
the reflecting medium. Large, smooth, 
metal surfaces of good electrical conduc- 
tivity (such as copper) are very efficient 
reflectors of radio waves, reflecting nearly 
all of the energy carried by the incident 
waves. The surface of the earth is itself 
a fairly good reflector of radio waves, 
particularly for waves that are incident 
at small angles from the horizontal; and 
the ionosphere, even though it is not a 
surface such as a mirror, is also a fairly 
good reflector of radio waves. 

(2) Figure 14 shows a planar wave front re- 
flected from a smooth surface. It should 



12 



REFLECTING 
SURFACE 



TM666-40 



Figure 14- Reflection of a planar wave front. 



be remembered here that, as in the re- 
flection of light, the angle of incidence is 
always equal to the angle of reflection. 
However, the incident wave front, A-Al. 
is reversed by the reflecting surface and 
appears at B-Bl 180° out of phase. 
The reason for this is that point X of the 
incident wave reaches the reflecting sur- 
face before point Y and is reflected to 
point X\ during the time it takes for 
point Y on the wave front to move to 
the point of reflection Fl. The parallel 
arrows indicate this change in phase. 
Refraction. 

(1) If a beam of light shines on a smooth 
surface of water, some of the light will be 
reflected and the remaining portion will 
penetrate the water, as shown in figure 
15. The phenomenon by which light 
waves penetrate the water in the manner 
shown is called refraction, and can be 
observed readily by examining a glass of 
water into which a spoon is immersed. 
If viewed from an angle, the spoon ap- 
pears broken or bent at the point where 
it enters the surface of the water. The 
reason for this is that light waves travel 
at a slower speed through water than 
through air. Thus, in figure 15, the 
direction of travel of the refracted light 
is different from that of the light beam 
incident on the surface of the water. 
Figure 16 shows how this change of di- 
rection of the light beam occurs. The 
parallel lines in this figure, which look 
like steps of a ladder, represent wave 
fronts of a beam of light incident on the 
surface of the water. It will be recalled 



that a wave front is a surface of equal 
phase perpendicular to the direction of 
travel of the wave. In the case of water 
waves previously discussed, the crests 
of adjacent circular expanding ripples 
would correspond to the wave fronts 
shown in the figure. 
(2) Consider the wave front, ^4-^11 (fig. 16), 
one portion of which is arriving at the 
surface of the water. Since the speed of 
light is less in water than it is in air, the 
point marked A. will advance the dis- 
tance, di, in a given length of time, 
whereas the point marked Al will travel 
a greater distance, d 2 , in the same length 
of time, since it is still passing through a 
faster medium. As a result, the wave 
front will be turned into a new direction, 
and the beam will follow this new direction . 
Note that refraction occurs only when 
the wave or beam of light approaches 
the new medium in an oblique direction; 
if the whole wave front arrives at the new 
medium at the same moment, it is slowed 




x TM 966-8 

Figure 15. Reflection and ref raction of a light beam. 

13 



, direction of travel 



INCIDENT LIGHT WAVES 




DIRECTION OF 
REFRACTED WAVE 



TM66(-4I 

Figure 16. Bending of a wave front by refraction. 

up uniformly and no bending occurs. 
The amount any wave is refracted, or 
bent as it passes from one medium to 
another is called the refractive index. 
This index depends on the relative densi- 
ties of the two media and is actually a 
ratio which compares the velocity of an 
electromagnetic wave through a perfect 
vacuum to its velocity through a denser 
medium such as the earth's atmosphere. 
c. Diffraction. If a beam of light in an other- 
wise blacked-out room shines on the edge of an 
opaque screen, it will be observed that the screen 
will not cast a perfectly outlined shadow. The 
edges of the shadow are not outlined sharply be- 
cause the light rays are bent around the edge of 
the object and decrease the area of total shadow. 
The diffraction or bending of a light wave around 
the edge of a solid object is slight. The lower the 
frequency of the wave, or the longer the wave- 
length, the greater the bending of the wave. 
Thus, radio waves are more readily diffracted than 
light waves, and sound waves more than radio 




F igure 1 7. Diffraction of waves around solid object. 

waves. A and B of figure 17 illustrate this phe- 
nomenon and help to explain why radio waves of 
the proper frequency can be received on the far 
side of a hill or other natural obstruction, and why 
sound waves can be heard readily from around 
the corner of a large building. Diffraction is an 
important consideration in the propagation of 
radio waves at a distance, because the largest ob- 
ject to be contended with is the bulge of the earth 
itself, which prevents a direct passage of the wave 
from the transmitter to the receiver. This prob- 
lem is analyzed in detail in paragraphs 12 to 18 
on ground-wave propagation. 



Section III. SUMMARY AND REVIEW QUESTIONS 



10. Summary 

a. Electromagnetic waves are a form of radis.nt 
energy propagated into space at nearly the speed 
of light, 300,000,000 meters or 186,000 miles per 
second. 

b. A radio wave may be described as a moving 
electromagnetic field having velocity in the direc- 
tion of travel. 

c. The wavelength of a radio wave in free space 
is equal to its velocity divided by its frequency 
(X=t>//). 



d. When power is delivered to an antenna, two 
fields are set up proportional to this power, an 
induction field and a radiation field. 

e. At a distance from the antenna, only the 
radiation field prevails. 

/. The radiated field has both electric and mag- 
netic field components. Each field component 
supports the other, and neither can be propagated 
by itself without setting up the other. 

g. The wave front of an electromagnetic wave 
is an expanding spherical surface all the points of 
which are of the same phase. 



14 



h. The electric and magnetic field components 
of a wave front are at right angles to each other 
and to the direction of travel. 

i. The direction of the components of a radiated 
field are determined by convention : If the field is 
moving away from an observer with the direction 
of the electric field given as downward, the direc- 
tion of the magnetic field is to the left. 

j. The polarization of an electromagnetic field 
is determined by the direction of its electric com- 
ponent. 

k. When the direction of the electric field is 
vertical, the wave is said to be polarized vertically; 
when the electric field is horizontal, the wave is 
said to be polarized horizontally. 

I. Vertical antennas should be used for the 
transmission and reception of vertically polarized 
waves, and horizontal antennas for horizontally 
polarized waves. 

m. The power in a radiated field at any point 
is measured in terms of the intensity of its electric 
field, which is expressed as volts, millivolts, or 
microvolts per meter. This intensity is a measure 
of the dielectric stress produced by the electric 
field, or the voltage induced in a conductor 1 meter 
long so held that it lies in the direction of the 
electric field and is at right angles to the direction 
of the magnetic field and to the direction of 
propagation. 

n. The power of any radiated wave falls off as 
the square of the distance between the source and 
the point of measurement. 

o. The intensity of the electric field of radiated 
electromagnetic waves falls off proportionally as 
the distance. 

p. At a distance of approximately 10 wave- 
lengths, or less, of the frequency transmitted, the 
power in the induction field is so small as to be 
negligible. 

q. The induction field is made up of two fields 
(the electrostatic and the magnetic) separated in 
space by 90° and in time by 90° — that is, in space 
quadrature and in time quadrature. 

r. The electric and magnetic components of the 
radiation field are in space quadrature and in time 
phase. 

s. The energy in the induction field is the stored 
energy usually associated with a resonant circuit, 
being returned each cycle to the antenna circuit, 
except for slight losses. 

t. The power in the radiation field is the power 
delivered to the antenna less the heat losses in the 
antenna circuit itself. 



u. The atmosphere is composed of three regions, 
named in order of their relative heights — the 
troposphere, the stratosphere, and the ionosphere. 

v. The ionosphere is defined as that portion of 
the earth's atmosphere above the lowest level at 
which ionization affects the transmission of radio 
waves. It is composed of several layers in which 
ionization occurs at different levels and intensities. 

w. The propagation of radio waves depends on 
the frequency of the wave, the location and height 
of the antenna, and the ability of the earth's at- 
mosphere to conduct the wave. 

x. Long-distance transmission takes place prin- 
cipally by means of sky waves, and short-distance 
transmission and all ultrahigh-frequency trans- 
mission take place by means of ground waves. 

y. Sky-wave propagation is variable, depending 
on the reflection or refraction of the wave from 
the ionosphere; ground-wave propagation is more 
constant, depending on the characteristics of the 
terrain, on reflection from the earth's surface, and 
on the diffraction of the wave around the curvature 
of the earth. 

z. Radio waves reflected from the surface of the 
earth, from large metal objects, and from the 
iorosphere suffer a change of phase polarization of 
the wave front. 

aa. Refraction is the bending of the beam of a 
radio wave when it passes from a medium of one 
density to that of another. 

ab. The refractive index is a ratio that com- 
pares the velocity of an electromagnetic wave 
through a perfect vacuum to its velocity through 
a denser medium. 

ac. Diffraction is the bending of a wave around 
the edges of a solid object; the lower the frequency, 
or the longer the wavelength, the greater is the 
bending of the wave. 

1 1 . Review Questions 

a. Describe an electromagnetic wave. 

b. What is the velocity of propagation of a 
radiated wave in free space? 

c. Describe simple wave motion in water. 

d. Give the formula for the length of any wave 
in terms of its velocity and frequency. 

e. When did Hertz first demonstrate the radia- 
tion of an electromagnetic wave? 

j. What two fields are set up about a trans- 
mitting antenna? 

g. Describe the components of the radiated 
field. 



15 



h. What is the conventional determination of 
the direction of the components of the radiated 
field? 

i. What is a vertically polarized wave? A hori- 
zontally polarized wave? 

j. What type of antenna should be used for 
receiving vertically polarized waves? 

k. What is the unit used in measuring the in- 
tensity of a radiated field? 

I. What is the meaning of the expression micro- 
volts per meter? 

m. Give a possible explanation of the propaga- 
tion of radio waves away from an antenna. 

n. What is the relationship between power and 
distance for a radiated wave? 

o. How are the components of the induction 
field related in time and space? The components 
of the radiated field? 

p. How do the power and the intensity of the 



electrostatic field vary with distance? Of the 
magnetic field? Of the radiated field? 

q. Describe the storage action of the induction 
field. 

r. How is the power in the radiation field 
related to the total power delivered to the 
antenna? 

s. Name the major regions of the atmosphere 
and describe the characteristics of each. 
t. Define the ionosphere. 

u. What major factors affect the propagation of 
radio waves? 

v. What type of transmission uses sky waves? 
Ground waves? 

w. What effect does reflection have on the 
phase of a wave front? 

x. What is refraction? 

y. Define the refractive index. 

z. What is diffraction? How does it vary with 
frequency? 



16 



CHAPTER 2 
MODES OF PROPAGATION 



Section I. GROUND-WAVE PROPAGATION 



1 2. Types of Ground Waves 

a. Ground-wave propagation refers to those 
types of radio transmission that do not make use 
of reflections from the ionosphere. Therefore, the 
field intensity of ground waves depends on other 
factors — the transmitter power, the characteristics 
of the transmitting antenna, the frequency of the 
waves, the diffraction of the waves around the 
curvature of the earth, the electrical characteristics 
(conductivity and dielectric constant) of the local 
terrain, the nature of the transmission path, and 
also, local meteorological conditions such as the 
distribution of the water vapor content of the at- 
mosphere. Most of the received ground-wave 



field intensity can be accounted for in terms of 
certain of these factors. Moreover, the earth it- 
self is a semiconductor, and, upon contact with 
its surface, some of the energy of the radiated 
wave is absorbed and rapidly dissipated in the 
form of heat. Thus, the losses suffered by ground- 
wave transmission are sometimes excessive and its 
use generally is limited to moderate-distance com- 
munication (up to several hundred miles) at low 
frequencies and to exceptional high-frequency 
applications. 

b. Figure 18 shows the way in which ground 
waves take a direct or reflected course from the 
transmitter to the receiver, or may be conducted 
by the surface of the earth and also refracted in 



TROPOSPHERIC PATH ' , 






Figure 18. Possible- routes for ground waves. 



17 



the troposphere. The resulting ground wave, 
therefore, can be considered as being composed of 
one or more of the following components: the 
direct wave, the ground-reflected wave, the surface 
wave, and the tropospheric wave. 



5j000 

4,800- 

4.600- 

4,400 

4,200 

4,000- 

3,800- 

3^600 

3,400 



1 3. Direct-Wave Component 

a. The direct wave is that component of the en- 
tire wave front wnich travels directly from the 
transmitting antenna to the receiving antenna. 



3,200-- 

£ 3,000-- 

u! 2,800-- 
X 2JB00-- 
£ 2400- - 
O 2,200-- 

• 2000- r 
w (,900- - 

6 isoo-ir 
5 1700 - 

1600- - 

x £00-- 

2 1400- L 

£ 1300- - 

< \200- r 

* 1,100-7 
ui 1,000 - - 
Z 900- 

o 80 °- 
5 700- 

t 600-±- 
2 

«j> 500- 
< 

K 400 



300- r 

200 -i r 
150-11- 



100- 



50-- 



Ui 



2 

Z 

UJ 

o 
z 
< - ■ 
I- 
«o 

a 

h- 
X 
(9 

V) 
i 

U. 

o 

I 
UJ 



:_ .---go 



200 
190 
-180 
•170 
160 
150 
■140 
•130 
120 
•110 
100 
■90 
SO 
70 
SO 
■ SO 
40 
30 



-5p00 
•4,800 
-4)600 
-4,400 
-4£00 
-4j000 
-3yB00 
3j600 
-3,400 
Jj200 
3,000 

2600 £ 
4-2/400 5 
■2£00 z 

2J0OO g 

- POO o 
WOO ^ 

■ 1,700 > 

-^600 g 

- 1,500 < 

- 1,400 1- 

- 1,300 o 

- (200 ui 

- 1,100 1 

" 1,000 t 

• 900 5 

-800 z 

"700 * 

-600 5 
> 

-J- 500 ui 
o 

UI 

r 400 k 

:r300 
4r 200 

4i- 150 



:: — -- too 



--10 



--50 



-"-0 
TM 666- 14 



Figure 19. Nomograph — antenna heights and line-of-sight distance. 



18 



This component of the ground wave is limited 
only by the distance to the horizon, or line of sight, 
from the transmit tor phis the small distance 
added by the atmosphere diffraction of the wave 
around the curvature of the earth. The total .limit- 



ing distance, then, is computed easily by assum- 
ing an earth with a radius 4/3 times its proper 
radius. Such an earth would have a larger cir- 
cumference and, hence, a longer distance to the 
horizon. This distance can be extended by in- 




FREQUENCY IN MEGACYCLES 

TM 666-11 



Figure 20. Attenuation of po wer with frequency and distance. 




19 



creasing the height of either the transmitting or 
the receiving antenna, and so effectively extend- 
ing the horizon. The nomograph of figure 19 
gives an approximation of the line-of-sight trans- 
mission range without any mathematical calcula- 
tions. Simply lay a straightedge on the chart so 
that it is alined with both the receiving-antenna 
height scale and the transmitting-antenna height 
scale, which are arranged on the two outside ver- 
tical lines of the chart. The transmission range 
then is indicated on the center vertical line at the 
point where the straightedge crosses it. An in- 
stance is shown by the dotted line — If the trans- 
mitting-antenna height is 30 feet and the receiv- 
ing-antenna height is 100 feet, the line-of-sight 
transmission range is approximately 20 miles. 

b. The electric field intensity of a direct wave 
varies inversely with the distance of transmission, 
as described in paragraph 6. This inverse-dis- 
tance attenuation is shown by the graph in figure 
20. For instance, the chart shows that for a 10- 
megacycle wave, 1 watt of radiated power suffers 
an attenuation of more than 80 db (decibels) below 
1 watt over a 25-mile direct-wave path. This may 



be converted into terms of electric field intensity 
(microvolts per meter) by referring to figure 21. 
On this latter chart, note that at 10 megacycles a 
loss of 80 db below 1 watt results m an electric 
field intensity of about 200 microvolts per meter. 
Also, note that the attenuation increases as the 
frequency of transmission is increased, but that 
attenuation is constant when distance is read in 
terms of wavelength, since the higher the fre- 
quency, the shorter the distance represented by a 
given number of wavelengths. Furthermore, the 
direct wave is not affected by the ground or by the 
earth's surface, but is subject to refraction in the 
tropospheric air between the transmitter and the 
receiver. This refraction is particularly important 
at very high frequencies and is explained in greater- 
detail in paragraphs 17 and 18. 

14. Ground-Reflected Component 

a. The ground-reflected component, as its name 
indicates, is the portion of the radiated wave that 
reaches the receiving antenna after being reflected 
from the ground or from the sea. For comnumi- 



'V "V 

s\s ^ V 

IN 180* OUT 90* OUT 

PHASE OF PHASE OF PHASE 




DIRECT-WAVE COMPONENT PLUS A PHASE 
LAG DUE TO GREATER DISTANCE TRAVELED 

Figure 22. Comparison of waie fronts of direct and reflected waves. 



20 



cation between points lower than a few thousand 
feet and separated by a distance of several miles, 
the ground-reflected wave takes on an importance 
comparable to the direct wave as a means of propa- 
gation. Upon reflection from the earth's surface, 
the reflected wave undergoes a phase reversal of 
180°, as noted previously, and this fact is impor- 
tant in determining the effect of its combining 
with the direct-wave component upon arrival at 
the point of reception. Since the reflected com- 
ponent travels a longer time in reaching its des- 
tination, a phase displacement over and above the 
180° shift caused by reflection will result. 

b. In figure 22, it may be seen that the waves 
start out with fronts of equal phase and continue 
in phase up to the point of reflection of the ground 
component. Beyond this point, corresponding 
wave fronts are 180° out of phase, plus whatever 
small phase displacement results from the rela- 
tively longer path of the reflected component. 
Where the reflected component strikes the ground 
at a small angle of incidence, the time lag will be 
small. The reflected wave will arrive at the re- 
ceiving antenna nearly 180° out of phase with the 
direct wave, and a cancelation of signal energy 
will result. This cancelation effect can be mini- 
mized by increasing the height of either antenna, 
since this tends to decrease the phase difference 
between the direct and the reflected components, 
and thus decreases the degree of signal voltage 
cancelation. 

1 5. Surface- Wave Component 

a. The surface wave is that component of the 
ground wave that is affected primarily by the con- 
ductivity and dielectric constant of the earth and 
is able to follow the curvature of the earth. When 
both transmitting and receiving antennas are on, 
or close to, the ground, the direct and ground- 
reflected components of the wave tend to cancel 
out, and the resulting field intensity is principally 
that of the surface wave. The surface-wave com- 
ponent is not confined to the earth's surface, how- 
ever, but extends up to considerable heights, 
diminishing in field strength with increased height. 
Because part of its energy is absorbed by the 
ground, the electric intensity of the surface wave 
is attenuated at a much greater rate than inversely 
as the distance. This attenuation depends on the 
relative conductivity of the surface over which the 
wave travels. Table II shows the relative con- 
ductivity for various types of surface. The best 



type of surface for surface- wave transmission is 
sea water; this accounts for the fact that relatively 
long-distance radio contacts have been established 
across the ocean. The electrical properties of the 
underlying terrain that determine the attenuation 
of the surface-wave field intensity vary little from 
time to time, and therefore, this type of transmis- 
sion has relatively stable characteristics. 

Table II. Propagation Characteristics of Local Terrain 



Type of surface 



Sea water 

Large bodies of fresh water 

Wet soil 

Flat, loamy soil 

Dry, rocky terrain 

Desert 

Jungle 



Relative 
conductivity 


Dielectric 
constant 


Good 


80 


Fair 


80 


Fair _ _ 


30 


Fair 


15 


Poor 


7 


Poor . 


4 


Unusable 





b. The surface-wave component generally is 
transmitted as a vertically polarized wave, and it 
remains vertically polarized at appreciable dis- 
tances from the antenna. This polarization is 
chosen because the earth has a short-circuiting 
effect on the electric intensify of a horizontally 
polarized wave but offers resistance to this com- 
ponent of the vertical wave. The ground cur- 
rents of the vertically polarized surface wave do 
not short-circuit a given electric field but rather 
serve to restore part of the energy used to the 
following field (fig. 23). The better the conduct- 
ing surface, the more energy returned and the less 
energy absorbed. Since no surface is a perfect 
conductor or perfect ground (one returning all of 
the energy) the loss retards the grounded edge 
of any given wave front, causing it to bend for- 
ward in the direction of travel so that the successive 
wave fronts have a forward tilt (fig. 23). The 
conducting surface of the earth is then, in effect, 
a part of a waveguide, and the tilt has the effect 
of propagating the energy in the direction of wave 
travel. Poor conducting surfaces cause high loss 
and greater tilt and, finally, total absorption of 
the energy. Table III shows the variation in 
angle of tilt from the vertical for frequencies 
from 20 kc (kilocycles) to 20 mc, propagated over 
sea water and over dry ground. As frequency 
increases, the angle of tilt increases. At 20 mc, 
the tilt is negligible over sea water, being little 
more than 1°, but over dry ground it is 35°, 
effecting a considerable change in polarization. 



21 



This tilting of the electric vector of an electro- 
magnetic wave is not to be confused with the 
bending of a wave, or diffraction, which is a 
phenomenon associated with a wave front striking 
the edge of a solid object, the greatest bending 
taking place at the lowest frequencies. 



Table III. Angle of Tilt Versus Frequency 



Frequency (mc) 


Angle of tilt over 
sea water 


Angle of tilt over 
dry ground 


0.02 


0°2. 5' 


4°18' 


.20 


. 0°8' 


13°30' 


2.00 


0°25' 


32° 12' 


20.00 


1°23' 


35° 



DIRECTION OF TRAVEL 

mil ml mil 

<^ ^-"^^ EARTH 

*" GROUND CURRENTS 

TM 668 -IS 

Figure 23. Tilting of electric vector of wave front. 

1 6. Frequency Characteristics of Ground Waves 

a. The frequency characteristics of the ground 
wave determine in large part the particular com- 
ponent of the wave that will prevail along any 
given signal path. When the conductivity of the 
earth is high and the wave frequency below 30 mc, 
the surface wave is the predominant component, 
except in the case of plane-to-plane or plane-to- 
ground transmission, in which the direct wave and 
ground-reflected waves are the principal means of 
communication. At frequencies greater than 10 
mc and less than 30 mc, the dielectric constant of 
the terrain determines the surface-wave trans- 
mission characteristics, the signal being largest for 
the higher dielectric constants and for the lower 
frequencies. At frequencies greater than 30 mc, 
the losses suffered by the surface wave become so 
excessive that transmission is usually possible only 
by means of the direct wave. However, at fre- 
quencies where the ground-wave field intensity is 
largely determined by the surface-wave com- 
ponent, vertically polarized radiation is superior to 
horizontally polarized radiation, except in heavily 
wooded or jungle areas. In such areas, horizontal 
polarization provides better gain, even at distances 
and frequencies where the surface-wave component 
normally would predominate, because most of the 



foliage grows vertically and absorbs vertically 
polarized energy. Above 30 mc, where the direct 
wave is the predominant component, the difference 
between vertical and horizontal polarization is 
negligible. 

b. These variations of the ground-wave compo- 
nents with frequency may be summarized, as 
follows: 

(1) The low-frequency band (0.03 to 0.3 mc) 

is used lor moderate-distance ground 
wave communication. In this band, the 
ground losses of a vertically polarized 
wave are small, and the wave is able to 
follow the curvature of the earth for 
several hundred miles. 

(2) The medium-frequency band (0.3 to 3 
mc), which includes the standard, broad- 
cast frequencies — amplitude modula- 
tion — is used for moderate-distance com- 
munication over land and for long- 
distance communication over sea water 
up to 1,000 miles. 

(3) The high-frequency band (3 to 30 mc) is 

used for short-distance communication. 
At these frequencies, the dielectric con- 
stant of the earth plays a greater part in 
the decrease of surface-wave field in- 
tensity and is the chief factor causing 
attenuation at very-high frequencies. 

(4) The very-high-frequency band (30 mc 
and over) is used for line-of -sight com- 
munication. At these frequencies, the 
direct-wave component is increasingly 
important. The direct-wave range there- 
fore can be extended greatly by increasing 
the height of transmitter and receiver 
antennas. Thus, it should be noted that 
whereas the distance range of the ground 
wave at low frequencies can be effectively 
increased only by increasing radiation 
power, the distance range of frequencies 
of 30 mc and higher can be effectively 
increased by increasing antenna heights 
as well as by increasing radiation power. 

17. Tropospheric-Wave Component 

a. The tropospheric wave is that component of 
the entire wave front which is refracted in the 
lower atmosphere by relatively steep gradients 
(rapid changes in respect to height) in atmospheric 
humidity and sometimes by steep gradients in 
atmospheric density and temperature. At heights 



22 



of a few thousand feet to a mile or so, huge masses 
of warm and cold air exist near each oilier, causing 
abrupt temperature differences and changes in 
density. The resulting troposphenc refraction 
and reflection make communication, possible over- 
distances far greater than can be covered by the 
ordinary ground wave. A of figure 24, shows the 
downward bending of a wave front which enters 
a layer of air the dielectric constant and moisture 
content of which decrease with height above the 
surface of the earth; B of figure 24 shows the up- 
ward bending of the same wave for the opposite 
condition. Since the amount of refraction in- 
creases as the frequency increases, fcropospherie 
refraction is more effective at, the higher fre- 
quencies, providing interesting communication 
possibilities at 50 mc and above. 



DIRECTION OF 
PROPAGATION 



LOW DIELECTRIC CONSTANT 

{LOW MOiSTUXe COMTCKT) 




HIGH DKLEC1 R! 



SURFACE OF EARTH 



DIRECTION OF 
PROPAGATION 



-HIGH Offl-CCTRIC CCK'-.l 



| 

SURFACE OF EARTH 



Figure S/ f . Refradimi in iht i v r . . 

b. One common cn " ' ' • r < 1 T 

is temperature Lave --.o 1 < . ' i > j - ■ 

results from seven* i .' i - > . t i > > 

overrunning a cold" r n,as- r! . , >♦ > i 

mass heated by cor i^o M > i i » col j.i* 

surface air after sun ' <.i ' j h it 



above a cloud layer by reflection of the sun's rays 
from the upper surface of the clouds. Tropo- 
spheric- propagation effect depends on weather 
■ 'Oii«i lions, and since these vary from minute to 
rnnute, they can cause fading or variable field 
intensity. In tropospheric-wave communication, 
the receiving and the transmitting antennas should 
have the same type of polarization, since the 
tropospheric wave maintains essentially the same 
; clanzaiion throughout its travel. 

1 8. Abnormal Effects of the Troposphere 

a. Propagation characteristics of the tropo- 
sphere vary under special weather conditions, and 
-ri Home parts of the world such conditions may 
t, - ! ong periods. In the tropics and over 
i - > re- of water, temperature inversions are 
h tost continuously at heights up to 
<•-., particularly in the range from 100 to 
,v v hen the boundary of the inversion is 
^ r ! ht.rj.iy, waves traveling horizontally or 
> -ngles of elevation become trapped by 

• ^' hog layer of air and continue to be bent 
> >' » aV the earth. Figure 25 shows how such 

i v> Hve follows a duct, the upper and lower 
v 1 - *. " "c 1 "h are formed by the boundary and the 
\ i waves are guided along within this 
i >> ha oh the same manner as in a metallic 
i< u "> t>nd since attenuation in a waveguide 
i^ti tK energy does not fall off inversely as 
- , liit- J the distance. Thus, the waves fol- 
h< fit ature of the earth for distances far 
1 .a 4 fl > >ptical horizon of the transmitter, and 
•> 1 . *ihtief may consistently reach distances 
i » o'i sands of miles. 

' >>■> •i-»tti.»;-,f ducts also are formed by the 
* 1 hue effect of two layers of air with sharply 
*. '.perature inversions. The refraction 

> L upper boundary bends the wave down, 
t .ip Tvfr>iOtio7i from the lower boundary bends 

> * m, effectively trapping the energy within 
•.fie :ayev The height of the duct determines the 
.uinmvmn frequency, and if this height is only a 
f-v: f ', 1 above the surface of the earth, or from 
•:».)>. uulary to boundary, transmission may be pos- 

• '•■■]•■ i-snlv sit the ultrahigh or superhigh frequencies. 

i \ , the height and the dielectric char- 

> •> u the layer are suitable for vhf trans- 
<ti [hiwever, a necessary feature of duct 

> . •>-, »Ti is that the angle of approach of the 
u % h ; be approximately half a degree or 
*» i for the wave to be trapped. In addi- 



HOT DRY AIR 



SHARPLY DEFINED BOUND- 
ARY BETWEEN HOT AND 
I COOL AIR 



TRANSMIT TMsl 
ANTENNA 




TM 666-16 



Figure 25. Transmission by means of iropospheric duct. 



tion, both the transmitting and the receiving 
antennas must be insids the duct, if communica- 
tion is to be established by this means. A trans- 
mitting antenna above the duct, as on a tower or 
promontory, will not operate into the duct, and 



no signals by this means will be received at the 
receiving antenna. Moreover, a receiving antenna 
below a duct will receive no signals from an air- 
plane flying in or above the duct, even though 
line-of-sight conditions prevail. 



Section li. THE IONOSPHERE 



19. General 

In the early days of radio, mathematical 
physicists reasoned that it would be impossible 
to receive radio signals at very great distances 
because of the attenuation resulting from the 
absorption of the energy by the earth. When it 
was found experimentally that signals could be 
received across the Atlantic Ocean, the work of the 
physicists was questioned. Their result was 
correct, of course, for the problem to which it 
applied — namely, the propagation of ground waves 
around a curved earth surrounded by free space, 
as has been noted in the section of ground waves*. 
Obviously, some other means of propagation had 
to exist. The experimental evidence of trans- 
Atlantic communication proved only that the 
assumption of an earth surrounded by nothing but 
free space was unjustifiable in this eonnectior.. 
It was then suggested by both Heaviside and Ken- 
nelly, the one an English scientist and the e ther a:i 
American, that the earth actually is surrounded 
by an electrified layer which acts as a reflector and 
prevents the escape of the wave into free space by 
bending it back toward the earth. Such a layer 
also could form the source of the electric currents 
in the upper atmosphere which had been suggested 
as the cause of changes in the magnetic field of the 
earth during magnetic storms. Later, when it 
was shown that not only one, but several such 



layers actually did exist, and that these layers 
consisted of ionized gases of the atmosphere, the 
name ionosphere was suggested for the region in 
which the layers were found. 

20. Formation olf ionosphere 

As indicated in figure 13, the earth's atmosphere 
extends up to a distance of over 250 miles. How- 
ever, the density of the gases that compose this 
atmosphere decreases with height, so that above 
250 miles the air particles are so rare as to be prac- 
tically nonexistent. Also indicated in figure 13 is 
the constant state of bombardment to which the 
atmosphere is subjected by radiation and particle 
showers from the sun and by cosmic rays, the 
source of which is not yet known. The radiation 
from the sun include* not only the light rays that 
we see, but also all the components of the entire 
spectrum, ranging from infrared rays to ultraviolet 
rays, and particle showers composed of positrons 
and electrons moving at almost the speed of light. 
As these different forms of radiation approach the 
atmosphere of the earth, they reach certain critical 
levels where the gases are of such density as to be 
particularly susceptible to ionization by their 
action, and at these levels ionized layers are 
formed. The upper layers of air appear to be 
affected most by particle radiation, although the 
predominant source of ionization is the ultraviolet 



24 



radiation. To understand how this ionization 
takes place, a brief review of the mechanics of 
gases is necessary. 



TV®© /\ 



:us ^' / 



NUCLEUS - 
ECTRON 



HYDROGEN ATOM HELIUM ATOM 

A 8 

TM666- !? 

Figure 26. Structure of hydrogen and helium atoms. 

a. Matter. All matter (solids, liquids, and 
gases) is made up of fundamental units called 
molecules. The molecule, the smallest subdivision 
of a substance that exhibits all its characteristic 
properties, is constructed of atoms of the elements 
comprising the substance. 

(1) The atom is made up of a central part 
called the nucleus, around which mi- 
nute particles or charges of electricity 
called electrons circulate (fig. 26), some- 
what as the planets revolve around the 
sun. In the normal or neutral atom, the 
electrical charge of each electron is 
balanced by an equal charge of opposite 
kind associated with the nucleus. The 
kind of charge represented by the electron 
is conventionally called negative, and 
that associated with the nucleus is called 
positive. 

(2) In the atoms of many substances, such as 
gases, one or more of the outer electrons 
are bound so loosely to the nucleus that 
they can be detached from the atom by 
suitable means, thus leaving the atom as 
a whole with a net positive charge. When 
this occurs, electrical activity becomes 
evident. 

b. Ionization. It has been found that energy 
in the form of electromagnetic radiation is capable 
of dislodging some of the loosely bound electrons 
from their atoms, provided that the radiation is 
of the proper wavelength and energy. When a 
number of such events happens in any gas, the 
gas is said to be ionized, since it has atoms lacking 
their normal quota of electrons, and free electrons 
unas^ociated with any atom. Atoms lacking in 
their normal quota of electrons are called positive 
ions, and electrons unassociated with any atom 



are called negative ions. The term ion is, in fact, 
applied to any elemental particle that has an 
electric charge. 

(1) Although a few ions may exist in any gas, 
external energy must be applied to the 
atom in order to produce an abundance of 
ions. A state of ionization is said to exist 
when all or a large proportion of the parti- 
cles in the gas are positive and negative 
ions. The external energy necessary for 
ionization can come from several sources, 
the most important being from an impact 
with another particle, from cosmic waves 
such as light waves, X-rays, gamma rays, 
and ultraviolet rays; from chemical re- 
action, or by the application of heat. 

(2) The natural ionization in air caused by 
any of the energies listed above produces 
approximately 2 ions per cubic centimeter 
per second at atmospheric pressure. At 
higher altitudes such as in the ionosphere, 
the rate of ionization is approximately 
100 times as great. In the upper atmos- 
phere, where the pressure is low, condi- 
tions are excellent for ionization to take 
place. The sun constantly gives off 
ultraviolet rays of the proper wavelength 
to ionize the gas particles of the upper 
atmosphere. 

c Recombination. The atoms and ions in a gas 
are in constant motion, and frequent collisions 
take place between them. When an electron col- 
lides with a positive ion, it may combi* with the 
ion and form again a neutral atom the gas. 
The process of recombination goes on coi antly, 
so that an atom, once it has been ioniz ., does 
not remain so indefinitely. The time that it takes 
for recombination, or deionization, depends on 
several .factors, but principally on the average 
distance between the particles of the gas. If only 
a few particles are present, as in the upper atmos- 
phere, collisions will not occur very frequently, 
and the particles remain ionized for relatively long 
periods. 

d. Source of Ionization — the Sun. Although the 
sun is composed of the same elements that are to 
be found on the earth, these elements exist in such 
a violent state of solar activity as to remain con- 
stantly in a molten or gaseous state. Probably 
because of intense internal stresses and the play 
of atomic forces on a gigantic scale, the sun con- 
stantly emits huge amounts of energy in the form 
of heat, particles, and electromagnetic waves. 



25 



Figur.i 27, Aolir eruptun. 



Eruptions at the surface of the sun have been 
noted to shoot immense clouds of hot gase? to 
distances of half a million miles above the surface 
(fig. 27). Another disturbance of the sun's sur- 
face is the appearance of sunspots, which l ave 
particular effects on the amount of ultraviolet 



radiation, and hence on the extent of ionization 

caused by this radiation. 

( 1 ) Effect of sunspots . During periods of high 
simspot activity, the extent of ionization 
of the various layers is greater than the 
average. The sunspots are dark areas 





Figur*. 28. Svmpots 



26 



which appear on the disk of the sun (fig. 
28), and although their relative darkness 
would seem to indicate lower tempera- 
tures and lower ultraviolet radiation, 
they have bright gaseous clouds about 
them, and the processes involved in the 
formation of the sunspots probably pro- 
duce vast amounts of ultraviolet energy. 
The sunspots usually appear in groups, 
and follow a more or less definite cycle of 
activity, with an average time interval of 
11.1 years between the maxima of two 
consecutive cycles. Magnetic storms on 
the earth abo are related to the presence 
of large sunspots. 
(2) Dellinger fade. Bright visible flares on 
the sun's disk instantaneously produce 
great effects on the ion density of the 
various ionospheric layers. This effect 
is known as the Dellinger fade, or sudden 
atmospheric disturbance. Great in- 
creases are noted in the ionization pro- 
duced at low levels of the ionosphere as 
the result of these flares. 
e. Formation of an Ionized Layer. When ultra- 
violet radiation of a particular wavelength enters 
the earth's atmosphere from above, the waves 
penetrate to a depth at which the critical density 
of the medium is sufficient to absorb the larger 
portion of the incident energy in the process of 
producing ionization. Before reaching this depth, 
the air particles are dispersed so sparsely that only 
a relatively few molecules of the rarefied gases be- 
come ionized. Upon reaching the level of critical 



density, a greater proportion of molecules are ion- 
ized, but the wave energy becomes so attenuated 
by the process of ionization that it is too weak to 
produce further ionization. The higher the fre- 
quency of the waves, the further they will pene- 
trate the atmosphere before reaching this level of 
critical density. Thus, an ionized layer is formed 
with the greatest intensity of ionization at its 
center and lesser intensities at its edges. How- 
ever, both the greater rate of recombination and 
the attenuation of the waves serve to decrease the 
extent of ionization in the lower or atmospherically 
denser part of an ionized region. 

21 . Ionosphere Layers or Regions 

By means of ionospheric sounding, it has been 
determined that there are four distinct layers of 
the ionosphere, which are called, in order of in- 
creasing heights and intensities, the D, E,F1, and 
F2 layers. The relative distribution of these layers 
about the earth is indicated in figure 29. As may 
be seen in this figure, the four layers are present 
only during the daytime, when the sun is directed 
toward that portion of the atmosphere. During 
the night, the Fl and F2 layers seem to merge into 
a single F layer, and the D and E layers fade out, 
because of the recombination of the ions compos- 
ing them. However, the actual number of layers, 
their heights above the earth, and the relative intensity 
of ionization present in them all vary from hour to 
hour, fivm day to day, from month to month, from 
season to season, and from year to year. 

a. D Region. At somewhat more than twice the 




Figure B9. Ionotphere layer* and region*. 



27 



highest altitude ever reached by man or between 
heights of 30 to 55 miles (50 to 90 kilometers) 
above the surface of the earth, is the first region of 
pronounced ionization,' known as the D region. In 
comparison with the conditions existing in the 
layers at greater heights, the amount of ionization 
in the D region is not extensive and has little effect 
in bending the paths of high-frequency radio waves. 
The chief effects of the ionization in this region are 
to cause a weakening or an attenuation of the field 
intensity of high-frequency radio waves as the 
transmission path of such waves crosses through 
this region, and to cause complete absorption of 
low- and medium-frequency radio waves. The D 
region exists only during the daylight hours, and 
its intensity follows the variation of the sun's al- 
titude, becoming most dense at noon, and fading 
out shortly after sunset because of the rapid re- 
combination of ions at this atmospheric height. 
It is chiefly responsible for the fact that the inten- 
sity of high-frequency waves is lower when the 
transmission path lies in sunlit hours than when 
the path traverses these regions during darkness. 

b. E Layer. At heights between 55 and 90 
miles (90 to 145 km) lies the second region in 
order of height, called the E layer. This layer 
sometimes is known as the Kennelly-Heaviside 
region after the names of the men who first pro- 
posed its existence. The layer height varies some- 
what with the season, lower heights occurring 
when the sun is in that latitude, probably because 
the solar ultraviolet radiation penetrates farther 
into the earth's atmosphere when the sun is more 
directly overhead. Since the rate of recombina- 
tion is large at this atmospheric level, the intensity 
of ionization of the E layer follows the sun's alti- 
tude variations closely, reaching a maximum at 
about noon, and fading to such a weak level during 
the night as to be practically useless as an aid to 
high-frequency radio communication. The num- 
ber of electrons per unit volume in this layer is 
usually great enought to refract back to earth 
radio waves at frequencies as high as 20 mc. 
Thus, the E layer is of great importance to radio 
transmission for distances less than approximately 
1,500 miles. For longer distances, transmission by 
this means is rather poor, because the necessarily 
k. vertical angle of departure of the wave from 
the ground results in greater absorption of the 
wave in the ionized region. At distances greater 
than 1,500 miles, better transmission can be ob- 
tained by means of the F, F\, and F2 layers. 

c. F Layer. At heights between 90 and 240 



miles (145 to 380 km) above the earth's surface is 
another region of ionization, known as the F layer. 
Ionization exists at all hours, usually with two 
well defined layers during the daytime and one 
during the night. In this region, at night, the 
single Flayer lies at a height of about 170 miles, 
or 270 km. The atmosphere is so rare at these 
heights that recombination of ions takes place at a 
very slow rate, and sufficient ions remain through- 
out the night to refract high-frequency waves back 
to earth. 

d. Fl and F2 Layers. During the daylight hours, 
especially when the sun is high, as in tropical 
latitudes and during summer months, the F region 
splits up into two distinct layers — the Fl, with a 
lower limit at a height of approximately 90 miles 
(145 km), and the F2, with a lower limit at a 
height of about 160 to 220 miles, depending on 
the seasons and the time of day. The F2 layer is 
the most highly ionized of all the layers and is the 
most useful for long-distance radio communica- 
tion. The degree of ionization for this layer ex- 
hibits an appreciable day-to-day variation in com- 
parison with that of the other layers. The 
intensity of ionization reaches a maximum in the 
afternoon and gradually decreases throughout the 
night. The rise of ion density is very rapid in the 
morning, and the low recombination rate permits 
this high ion density to persist. 

e. Other Layers. In addition to these regions of 
ionization which appear regularly and undergo 
variations daily, seasonally, and from year to year 
in height and ionization, other layers occasionally 
appear, particularly at heights near that of the E 
layers, much as clouds appear in the sky. Fre- 
quently their appearance is in sufficient intensity 
to enable good radio transmission to take place by 
means of reflection from them. At other times, 
especially during disturbed conditions in polar 
regions such as those that cause the aurora boreaiis, 
diffuse ionization may occur over a fairly large 
range of heights and may be detrimental to radio 
transmission because of the excessive absorption 
produced. 

22. Ionosphere Characteristics 

Al though a detailed analysis of what happens to 
electromagnetic waves as they enter the ionosphere 
is beyond the scope of this manual, the principal 
factors and assumptions are described in this 
paragraph in terms of practical effects. 

a. Critical Frequency. In addition to the height, 



28 



the principal ionosphere characteristic which con- 
trols or determines long-distance radio trans- 
mission is -the ionization density of each of the 
layers. The higher the frequency, the greater the 
density of ionization required to reflect waves 
back to earth. In other words, the shorter the 
length of the wat*es, the denser or more closely 
compacted must be the medium to refract them. 
Therefore, the upper layers, which are the most 
highly ionized, reflect the higher frequencies, 
whereas the D layer, which is the least ionized, 
does not reflect frequencies above approximately 
500 kc. Thus, at any given time, jor each layer 
there is a value of highest frequency, called the 
critical frequency, at which waves sent vertically 
upward are reflected directly back to earth. 
Waves of frequencies higher than the critical 
frequency pass on through the ionized layer and 
are not reflected back to earth, unless they are 
reflected from an upper layer. Waves of frequen- 
cies lower than the critical frequency are reflected 
back to earth, unless they are absorbed by, or 
have been reflected from, a lower layer. 

(1) Figure 30 shows waves of different fre- 
quencies radiated vertically into the 



ionosphere. Two of these signals are 
returned directly to the earth and the 
third passes through both layers and is 
not returned. Each of the returned 
waves is at the critical frequency for its 
layer — that is, the highest frequency that 
is returned. All frequencies below the 
critical frequency are returned in the 
same manner. The unreturned signal 
is at a frequency above the critical 
frequency for either layer and therefore 
passes through to outer space. 
(2) This phenomenon may be understood in 
terms of the combined refractive and 
reflective effects of ionization on an 
electromagnetic wave. When a ray, or 
tram of waves, enters an ionospheric 
layer, it is slowed down as soon as it 
starts to penetrate the layer. This 
process of refraction is similar to that of 
the refraction of light passing from air to 
water, as explained in paragraph 9. 
When the signal enters the ionosphere at 
a 90° angle, there is no bending of the 
wave— the whole wave front is slowed 




VftVtS AT 
CHITtCAU 

mawwicies 



I i 



I 
i 

i ■ 
i 

i i 



WAVE ABOVE 

CRITICAL 
..FREQUENCY 



EARTH 



Figure SO. Critical frequencies. 



29 



down uniformly. The higher the fre- 
quency of the signal, the deeper it must 
penetrate the layer before it surrenders 
all of its energy. It should be remem- 
bered, however, that an ionization layer 
is most dense near its center, and that 
the wave will pass on through if this 
center density is insufficient to absorb all 
of the energy. The surrendered energy 
is reradiated by the layer, directly 
downward to the area of transmission. 
By analogy, this effect is similar to that 
obtained in tossing tennis balls vertically 
upward to a wire screen. If the openings 
in the screen are smaller than the diam- 
eter of the ball, all of those thrown are 
reflected back almost as effectively as 
though the screen were a solid piece of 
metal. But if golf balls were thrown at 
the same screen, most of them would 
pass through the screen and not be 
reflected. Thus, it may be concluded 
that (like frequency in respect to ioniza- 
tion) there is for a given screen a critical 
diameter of ball which will be reflected 
back; any smaller ball will pass on 
through. 

b. Critical Angle. The determination of a criti- 
cal frequency by vertical propagation is useful be- 
cause it marks a boundary condition. Electro- 
magnetic waves used in radio communications, 
however, are generally incident at some oblique 
angle to the ionosphere. These waves are re- 
fracted by the ionosphere and may or may not be 
returned to the earth. Obviously, any frequency 
at or below the critical frequency will be returned 
to the earth, but frequencies above the critical fre- 
quency also will be returned if propagated at cer- 
tain angles of incidence. This effect can be under- 
stood by considering for a moment our former 
analogy of the screen and the tennis balls. If, for 
instance, the critical diameter for a given screen 
is the diameter of a tennis ball, then golf balls 
thrown obliquely at the screen will hit the wire 
mesh at an angle and be reflected downward, even 
though they would pass easily through the screen 
if thrown vertically upward. Thus, for this screen 
there is a certain angle of incidence at which most 
of the golf balls would be reflected downward. In 
electromagnetic-wave propagation the same con- 
ditions prevail. At angles of incidence near the 
vertical, a given frequency passes on through the 
ionosphere. But as the angle lessens, a point is 



reached at which the wave is reflected back to 
earth. This angle is called the critical angle. 
The point at which the wave returns is a minimum 
distance, called the skip distance; at smaller angles 
of incidence, the wave returns at greater and 
greater distances. 

(1) The concept of critical angle may be un- 
derstood by consideration of a similar 
optical phenomenon (fig. 31). If a beam 




Figure SI. Critical angle of light beam. 

of light passes from a dense medium 
(water) to one of less density (air) at right 
angles to the boundary between them, it 
passes through with no change in direc- 
tion. As the angle of incidence becomes 
smaller than 90°, the beam is only slightly 
reflected back into the water, most of the 
light being refracted or bent in the air. 
The amount of bending increases as the 
angle grows smaller, but at a given angle, 
which is the critical angle, no light is re- 



30 



fracted by the air, all of it being reflected 
back into the water. At angles smaller 
than the critical angle, the light beam is 
reflected back at greater and greater dis- 
tances from the source. At this point, 
it should be noted that this optical phe- 
nomenon cannot be applied strictly to 
radio waves, since the boundaries be- 
tween the dense ionization of the layer 
and the air above it are not sharp bound - 
ries. The wave is both bent and re- 
flected, and therefore, in propagation 
work the terms refraction and reflection 
tend to be used interchangeably. 
(2) The popular explanation of the return of 
radio waves as a phenomenon in refrac- 
tion alone is illustrated in figure 32. 




Figure 82. Idealized refraction of radio wave. 

Suppose a train of wave fronts to be 
propagated from A so that it enters the 
ionosphere at an angle, 0. As each wave 
front enters the ionosphere, the upper 
part of the wave front feels the effect of 
lowered index of refraction first. There- 
fore, the upper part of each wave front 
has an increased phase velocity, so that 
the entire wave front as it enters the 
ionosphere wheels about like a column of 
soldiers obeying the command, "Column 
right." Since the central parts of the 
ionosphere have a greater ion density, the 
bending effect on the upper part of the 
wave front is greatest, so that the wheel- 



ing process continues and the waves are 
directed back toward the earth at 
point B. 

c. Virtual Height. The oversimplified curved 
path shown in the figure also helps to make clear 
the notion of virtual height of an ionospheric 
layer. In following the curved path, as illustrated 
in the schematic drawing of figure 33, the time of 



A VIRTUAL HEIGHT 




TMeee-22 



Figure S3. Virtual height of ionospheric layer. 

transmission of the radio wave along the actual 
path BCD in the ionized layer is considered to be 
the same as would be required for transmission 
along path BAD if there were no ionized particles 
present, and a perfect reflecting surface at A in- 
stead. The height, H, from the earth to the inter- 
section of the two projected straight parts of the 
path is called the virtual height of the layer. 
Note that this virtual height is considerably 
greater than the actual layer height. However, 
it is a convenient and an important quantity in 
measurements and applications involving iono- 
spheric reflections. 

23. Regular Variations of Ionosphere 

a. General. Since the existence of the iono- 
sphere is dependent on radiations from the sun, it 
is obvious that the movements of the earth about 
the sun, or changes in the sun's state of activity 
which might serve to cause an increase or decrease in 
the amount of its radiation, will result in variations 
in the conformation of the ionosphere. These varia- 
tions include those which are more or less regular 
in their nature and, therefore, can be predicted in 
advance, and the irregular variations resulting 
from the abnormal behavior of the sun. For pur- 



31 



poses of discussion, the regular variations may be 
divided into four classes: the diurnal or daily 
variation, the seasonal, the 11-year, and the 27- 
day. For convenience, table IV lists the regular 



variations together with the resulting effects upon 
the ionosphere and on radio communications, and 
also gives suggestions that may be followed in 
compensating for these effects. 



Table IV. Regular Variations of Ionosphere 



Type of variation 



Effect on ionosphere 



Effect on communications 



Method of compensation 



Diurnal (variation 
with hour of day) . 



Seasonal. 



11-year sunspot cycle. 



27-day (sunspot). 



F layer: Height and density de- 
crease at night, increase after 
dawn. During day, layer 
splits into — (1) Fl layer: 
Density follows vertical angle 
of sun; (2) F2 layer: Height 
increases until midday, den- 
sity increases until later in 
day. 

E layer: Height approximately 
constant, density follows verti- 
cal angle of sun. Practically 
nonexistent at night. 

D layer: Appears after dawn, 
density follows vertical angle 
of sun, disappears at night. 

FZ layer: Virtual heights increase 
greatly in summer, decrease in 
winter. Ionization density 
peaks earlier and reaches 
higher value in winter. Mini- 
mum predawn density reaches 
lower value in winter. 



Fl, E, and D layers: Reach 
lower maximum densities in 
winter months. 

Layer density increases and de- 
creases in accord with sunspot 
activity (maximum, 1947- 
1948 and 1958-1959; mini- 
mum 1944 and 1955). 



Recurrence of SID's (sudden 
ionospheric disturbances) and 
ionospheric storms at 27-day 
intervals. Disturbed condi- 
tions frequently may be identi- 
fied with particularly active 
sunspots whose radiations are 
directed toward the earth 
every 27 days as the sun 
rotates. 



Skip distance varies in 1- to 
30-mc range. Absorption 
increases during day. 



MUF's (maximum usable fre- 
quencies) (par. 28), gen- 
erally reach higher midday 
values in winter but main- 
tain high values later into 
afternoon in summer. Pre- 
dawn dip in MUF's reaches 
lower value in winter. Less 
absorption encountered in 
winter. 



Higher critical frequencies 
during years of maximum 
sunspot activity. MUF 
variation: Sunspot max: 
8-42 mc; sunspot min: 
4-22 mc. 



See effects of SID's and iono- 
spheric storms in table V. 



Use higher frequencies dur- 
ing day, lower frequencies 
at night. 



Provide greater spread be- 
tween nighttime and day- 
time operating frequen- 
cies in winter than in 
summer. 



Provide for higher operating 
frequencies to be used 
during periods of sunspot 
maximum and lower fre- 
quencies for use during 
minimum. (Consult TB 
11-499 to determine 
MUF.) 

See compensation for SID's 
and ionospheric storms in 
table V. 



32 



b. Diurnal. For the most part, the diurnal 
variations and their effects upon the ionosphere 
layers have been discussed in the description of 
these layers (par. 21). Note, in table IV, that 
to compensate for the resulting variations in the 
skip distance, it is suggested that higher medium 
frequencies be used during the daytime, and 
lower medium frequencies at night. The reason 
for this appears in the fact that the ion density 
of the F2 layer is greater during the daytime, and 
will reflect radio waves of higher frequency than 
the F layer will reflect during the night. The 
higher frequency waves suffer less absorption in 
passing through the D region, whereas at night 
the disappearance of the D region permits the use 
of lower frequencies. 



F2 


F2 


E 


— . _. _ E 


D 


D 









F2 












E 










/ , SUMMER \ 


WINTER \^ 



00 06 12 18 00 06 12 16 

TM 666-23 

Figure 34- Daily and seasonal variations in ion density. 



c. Seasonal. As the apparent position of the 
sun moves from one hemisphere to the other with 
corresponding changes in season, the maximum 
ion density in the D, E, and Fl layers shifts accord- 
ingly, each being relatively greater during the 
summer. The F2 layer, however, does not follow 
this pattern in seasonal shift. In most localities, 
the F2 ion density is greatest in winter and least 
in summer, which is quite the reverse of what 
might be expected from simple theory. Figure 
34 shows graphs of the relative ion densities of all 
layers, and also the relative heights of these 
layers above the surface of the earth, as the 



seasonal shifts occur. Note that, in winter the 
ion density of the F2 layer rises to a sharp peak 
at about noon, and assumes a much higher 
density than in summer. Also, note that the 
separation of the Fl and F2 layers is not so well 
defined in summer, since the height of the F2 
layer is relatively less during that season. 

d. Eleven-year. That sunspot activity varies 
according to an 11 -year cycle has been known 
since 1851. Shortly after the discovery of this 
phenomenon, a method was devised for measuring 
the relative intensity of sunspot activity, and, 
by means of this method, the alternations from 
maximum to minimum have been followed closely 
throughout the years. Briefly, the method entails 
the use of the so-called Wolf sunspot number, a 
number obtained for each day by multiplying 
by 10 the number of distinct visible sunspot 
groups and adding thereto the number of individ- 
ual spots observable in the groups. For months 
at a time, the visible surface of the sun may be 
devoid of spots and so that sunspot number 
zero (comparing views shown in fig. 28). This 
frequently occurs during times of sunspot minima. 
At other times, even the mean annual sunspot 
number has been known to rise as high as 140, 
with daily values running into the hundreds 
These conditions occur at the maximum of the 
sunspot cycle. Although the time from minimum 
to minimum has been found to be variable, it 
averages around 11 years. Also, the height of 
the maximum and the depth of the minimum vary 
from cycle to cycle. The increased activity at 
times of sunspot maxima is reflected in an increase 
in ion density of all the ionosphere layers, 
resulting in higher critical frequencies for the 
E, Fl , and F2 layers, and higher absorption in the 
D region. This permits the use of higher fre- 
quencies for communication over long distances 
at times of sunspot maxima than would be usable 
at times of sunspot minima. The increased 
absorption in the D region, which has the greatest 
effect on the lower frequencies, requires that higher 
frequencies be used, but the over-all effect is an 
improvement in propagation conditions during 
sunspot maxima as the critical frequencies are 
raised more than the absorption limits. 

e. Twenty-Seven Day. Another cycle that is 
due to sunspot activity is the 27-day variation re- 
sulting from the rotation of the sun on its axis. 
As the number of sunspots changes from day to 
day with solar rotation or the formation of new 
spots or the disappearance of old ones on the 



33 



visible part of the sun, absorption by the D region 
also changes. Similar changes are observed in the 
E layer critical frequency. These variations ex- 
hibit wide geographic range; they are not effects 
that are observed at one station and not observed 
at others. Although fluctuations in F2 layer criti- 
cal frequencies from day to day are greater than 
for any other layer, these fluctuations are not gen- 
erally of a world-wide character. Because of the 
variability of the F2 layer, precise predictions of 
its critical frequencies cannot be made for indi- 
vidual days, although seasonal and long-term 
trends and geographic distribution may be out- 
lined accurately in advance. It is necessary in 
selecting frequencies for long-distance communica- 
tion to allow for these fluctuations. 

24. Irregular Variations of Ionosphere 

In addition to the more or less regular varia- 
tions in the characteristics of the ionosphere, a 
number of singular, transient effects, though un- 
predictable, have important bearing on propaga- 
tion phenomena. Some of the more prevalent 
of these effects are: sporadic E; sudden ionospheric 
disturbance (Dellinger fade); ionospheric storms; 
and scattered reflections. These variations have 
been listen for convenience in table V where their 
effects on the ionosphere and on radio communica- 
tion, together with suggestions for compensating 
for them, are given. 

a. Sporadic E. The sporadic E, also known as 
the E, layer, is an ionized cloud that appears at 
indefinite intervals, and at a slightly greater height 
than the normal E layer. The nature and cause 
of this abnormal layer are as yet unknown. Some- 
times the sporadic E consists of an extremely 
efficient radiating surface that is capable of re- 
flecting so much of the energy radiated from the 
transmitting antenna, even at frequencies of 10 
to 15 mc, that reflections from the other layers of 
the ionosphere are blanked out completely. At 
other times, the sporadic E may be so thin that, 
although its presence can be verified by sounding, 
reflections from the upper layers easily can be re- 
ceived through it. The sporadic E layer may 
occur during the day or night. Its occurrence is 
frequent, and thus, from 25 to 50 percent of the 
time, long-distance propagation at frequencies up 
to 15 mc is rendered possible by its means in 
middle latitudes. Occurrence of sporadic E is 
not usually simultaneous at all stations. In gen- 



eral, tropical stations exhibit less sporadic E than 
stations in higher latitudes. 

b. Sudden Ionospheric Disturbance or Dellinger 
Fade. The most startling of all the irregularities of 
the ionosphere and of radio wave transmission is 
the sudden type of distrubance manifested by a 
radio fadeout. This disturbance, abbreviated 
SID, and sometimes called the Dellinger fade, 
comes without warning and may prevail for a 
length of time from a few minutes to several hours. 
All stations lying wholly or in part on the sunward 
side of the earth are affected, and, at the onset of 
SID, receiving station operators are inclined to 
believe that their radio sets have suddenly gone 
dead. Examination of the sun at the times of 
occurrence of these effects, however, has revealed 
that in all cases where reliable solar data were 
available the appearance of this ionospheric dis- 
turbance was coincidental with the onset of a 
bright solar eruption (fig. 35), and its duration was 
the same as that of the eruption. Such an eruption 
causes a sudden abnormal increase in the ionization 
of the D region, frequently with simultaneous dis- 
turbances in terrestrial magnetism and earth cur- 
rents. Such increases in D region ionization 
usually result in total absorption, in this region, of 
all frequencies above 1,000 kc. 

c. Ionosphere Storms. An ionosphere storm is a 
period of disturbance in the ionosphere, during 
which there are large variations from normal, of 
critical frequencies, layer heights, and absorption. 
These storms may last for periods of varying in- 
tensity (from several hours to several days), and 
usually extend over the entire earth. High- 
frequency sky-wave transmission above approxi- 
mately 1,500 kc then shows low intensity and is 
subject to flutter fading. During the first few 
hours of severe ionosphere storms, the ionosphere 
is turbulent, stratification is destroyed, and radio- 
wave propagation is erratic. During the later 
stages of severe storms and during the whole 
period of more moderate storms, the upper part of 
the ionosphere is expanded and diffused. The 
critical frequencies are much lower than normal 
because of a decrease in ion density, and the virtual 
heights of the layers much greater, so that the 
maximum usable frequencies are much lower than 
normal. It is often necessary to lower the working 
frequency to maintain communication during one 
of these storms. There is also increased absorption 
of radio waves during the storm. Ionosphere 
storms are most severe at the higher latitudes and 
decrease in intensity toward the equator. These 



34 



Table V. Irregular Variations of Ionosphere 



Type of variation 



Sporadic E layer- 



Sudden ionospheric 
disturbance (SID). 



Ionospheric storm. 



Scattered reflections. 



Effect on ionosphere 



Clouds of abnormal ionization 
occurring in the E layer or 
slightly above for a large por- 
tion of time each month result 
in abnormally high critical 
frequencies. Usually spotty 
in geographic extent and time. 

Unusual amount of ultraviolet 
radiation from solar flare re- 
sults in abnormally high ion- 
ization in all layers. Ioniza- 
tion increase occurs with great 
suddenness throughout day- 
light portion of earth. 



Effect on communications 



Usually accompanies magnetic 
disturbance occurring about 
18 hours after SID's. Prob- 
ably both are due to abnormal 
particle radiation. Upper 
ionosphere expands and dif- 
fuses, critical frequencies be- 
low normal, virtual heights 
above normal. 

Severest effects toward geo- 
magnetic poles, decreasing to- 
ward equator. Few minutes 
to several hours in duration; 
effects disappear gradually in 
few days. 
The ionospheric layers are not 
smooth. Irregularities in den- 
sity and in height are normal. 



Excellent transmission within 
normal skip distance. Oc- 
casionally, long-distance 
communications on fre- 
quencies of 60 mc or higher 
are possible. 



Normal frequencies above 1 
or 1.5 mc are rendered use- 
less because of high absorp- 
tion in the abnormally-ion- 
ized D layer. Frequencies 
considerably higher than 
normal will survive this ab- 
sorption for short hops. 
Low frequencies may not 
penetrate the D layer and 
thus may be transmitted for 
long distances. 

Limits number of usable high 
frequencies. 



Because of irregularities in the 
ionosphere, the electric field 
at a receiver consists of 
several fields arriving from 
slightly different directions 
with varying phase relation- 
ships. The result is fading 
of the signal resulting from 
cancelation and reinforce- 
ment. 



Methods of compensation 

Frequency may have to be 
lowered to maintain short- 
skip communications. At 
times, long-distance com- 
munications on abnor- 
mally high frequencies are 
possible. (See TB 11- 
499.) 

Raise working frequency 
above normal for short- 
hop transmission. Lower 
frequency below normal 
for long-hop transmission. 



Use frequencies lower than 
normal, particularly in 
high latitude circuits. 



Fading of short duration. 
No compensation required. 



storms probably are caused by abnormal particle 
radiation from the sun, and are likely to occur 
during periods of great solar activity. The storms 
are most likely to start about 2 days after an 
active sunspot group crosses the center of the sun's 
disk. 

d. Scattered Reflections. An irregular type" of 
reflection from the ionosphere occurs at all seasons 
and is prevalent both day and night. The iono- 



sphere layers are irregular, and the presence of 
ionized clouds or scattering patches at E layer 
heights has been mentioned previously. Irregular 
reflections are obtained from these because of the 
rapid change of ionization with height. A radio 
wave can reflect from either the top or bottom of 
one of these scattering clouds, and these reflections 
make possible the reception of signals within the 
normal skip zones and at frequencies much higher 



35 




TM666-24 



Figure 35. Bright solar eruption. 

than those well receivable from the regular layers. 
The reflections may cause signal distortion and 
contribute to so-called flutter fading. Signals 
received from such reflectipns either may arrive 
from all directions or, if the transmitter operates 
with a highly directional antenna, may appear to 
come from the direction in which the antenna is 



pointed. The field intensity at the receiving sta- 
tion may be the sum of the components of several 
contributing radio waves of varying phase rela- 
tions. Figure 36 shows the effect of just two of 
these scattered signal components arriving at a 
receiving station by different paths, the one by 
reflection from the lower surface of the scattering 
cloud of ionization, the other after re-reflection 
from the top of this same cloud. It is obvious that, 
with respect to the latter signal component, a 
time lag will occur which either will serve to cancel 
a portion of the signal reflected directly from the 
bottom surface of the scattering cloud, and thus 
cause fading, or will augment this signal, depend- 
ing upon the phase relations of the two components 
at the receiving station. 

25. Ionosphere Predictions 

By sounding the ionosphere it is possible to pre- 
dict for several months in advance the various 
important characteristics of the ionosphere above 
any point on the surface of the earth. Such 
predictions are useful in the selection of optimum 
frequencies for radio communication over a defi- 
nite path at particular times. The average vari- 
ations of the critical frequency and maximum 
usable frequency factors are sufficiently well 
known to permit long-range predictions to be made 
for average conditions on ionospherically quiet 




36 



Figure 86. Scattering of signal components of radio wave. 



days (days without ionosphere disturbances). One 
of the basic principles employed in all ionosphere 
predictions is the relation of ionospheric charac- 
teristics to the sunspot cycle. The actual predic- 
tion consists essentially of first predicting the 
solar activity and then deducing, from the mass of 
data available, the corresponding trends of sea- 
sonal, diurnal, and geographic variations of the 
ionosphere characteristics. No attempt is made 
at present in these long-range predictions to 
estimate the detailed day-to-day variations in the 



ionosphere, these being rather localized and de*- 
pending on conditions of solar radiation and 
terrestrial effects of the particular days. Also, 
no account is taken of ionospheric disturbances', 
either of the ionosphere-storm type or of the 
sudden ionosphere-disturbance type. These abf 
normalities constitute a different forecasting 
problem. Regular predictions now are published 
monthly, for a period of 3 months in advance, as 
TB ll--499-( ), Basic Radio Propagation Predic- 
tions. 



Section III. SKY-WAVE PROPAGATION 



26. General 

Sky-wave propagation refers to those types of 
radio transmission that make use of ionospheric 
reflections to provide signal paths between trans- 
mitters and receivers. Sky-wave transmission, 
being by far the most important method for long- 
distance radio communication, presents many 
problems that can be solved adequately only 
through a complete understanding of the prin- 
ciples involved. A typical question in sky-wave 
propagation is whether the ionosphere will support 
(reflect) a radio wave of a particular frequency 
and whether the received signal will be strong 
enough at the receiver to be heard above the noise 
level present at the receiver. The answer to this 
question can be given only after considering the 
many factors involved — what particular path the 
radio wave will take in traveling from transmitter 
to receiver, whether the frequency of the radio 
wave lies between the limits determined by the 
maximum usable frequency and the lowest useful 
high frequency for the particular signal path; and 
the field strength that may be expected of the 
signal upon its arrival at the receiver (received 
signal strength). 

27. Sky- Wave Transmission Paths 

Figure 37 illustrates some of the many possible 
paths of radio waves from a transmitter to a re- 
ceiver as transmitted by reflection from an elec- 
trically conducting layer of the ionosphere. Note 
that some of the components of the entire wave 
front, which in this case are assumed to be of too 
high a frequency for reflection by the ionized layer, 
pass on through and are lost in outside space, 
unless they happen to be reflected from some 
higher layer having a greater degree of ion density. 



Other components of the wave, which are assumed 
to be of the correct frequency for reflection from 
the ionosphere layer, are returned to earth, and 
it is these components of the wave that provide 
communications. Note also that the skip distance 
is that distance from the transmitter at which the 
ion density of the layer will just support reflection. 
The skip zone and its relation to the ground wave 
are shown in figure 38. When the skip distance 
becomes less than the inner limit of the skip zone, 
both the sky wave and the ground wave may 
have nearly the same field intensity but a random 
relative phase. When this occurs, the field of the 
sky wave successively reinforces and cancels that 
of the ground wave, causing severe fading of the 
signal. Note the distinction between the terms 
■skip distance and skip zone. For each frequency 
(greater than the critical frequency) at which 
reflection from an ionosphere layer takes place, 
there is a skip distance that depends only on the 
frequency and the state of ionization. The skip 
zone, on the other hand, depends on the extent of 
the ground-wave range and disappears entirely if 
the ground-wave range equals or exceeds the skip 
distance. 

a. Sky-Wave Modes. The distance at which 
the wave returns to the earth depends on the 
height of the ionized layer and the amount of 
bending of the path while traversing the layer, 
the latter depending on the frequency of the wave 
as compared to the ion density of the layer 
required to refract or bend the wave. Upon re- 
turn to the earth's surface, part of the energy 
enters the earth, to be rapidly dissipated, but 
part is reflected back into the ionosphere again, 
where it may be reflected downward again at a 
still greater distance from the transmitter. This 
means of travel in hops, by alternate reflections 
from the ionosphere and from the surface of the 



37 




38 




Figure 38. 

earth, may continue, and enables transmission 
to be received at long distances from the trans- 
mitter. Figure 39 illustrates this means of travel 
for paths involving one and two reflections from 
the ionosphere (single- and double-hop modes). 
Figure 40 further illustrates this means of travel 
and reflection from different layers, with the 
layers represented by lines for simplicity. Figure 
40 also relates the heights of the various ionized 
layers to actual distances along the earth's 
surface. 

b. Qreat-CvrcU Path. The paths which the radio 
waves normally traverse in traveling from the 
transmitter to the receiver lie in the plane passing 
through the center of the earth and the trans- 
mission and reception points. The intersection 
of this plane with the surface of the earth is the 
great-circle path between the transmission and 
reception points. Radio-wave transmission paths 
which lie in this plane generally are called, for 
brevity, great-circle paths. Frequently, however, 
waves do not follow paths confined to this plane, 
and this deviation is called non-great-circle trans- 
mission. The part of the ionosphere which 
controls sky-wave propagation is the portion 
directly above the great-circle path. For single- 
hop transmission, this portion is a region centered 
above the midpoint of the great-circle path. 
The situation becomes more complicated when the 
transmission path is too long for a single hop, 
and it is necessary to consider ionsphere con- 
ditions at more than one point along the path 
between the transmitter and the receiver. Waves 
can follow either the major arc or the minor arc 
of the great-circle path. For instance, radio 



tts««a-ae 

Skip tone. 

waves emanated at New York City might travel 
cross-country, or westward, to reach San' Fran- 
cisco, which would be along the minor arc of the 
great-circle path between these cities, or these 
waves might travel eastward, almost around the 
world to the same destination, which would be 
along the major arc. The two types of trans- 
mission are called short-path and long-path trans- 
mission, respectively. 

c. Frequency. As noted previously in the dis- 
cussion of the ionosphere, the higher the fre- 
quency of a wave, the less it is refracted by a given 
ion density. Thus, if the angle of incidence of 
the wave with the ionosphere is fixed and the 
frequency increased, the minimum distance be- 
tween the transmitter and the point of return of 
the wave to the earth increases slightly. Figure 41 
shows three separate waves, of different fre- 
quencies entering at the same angle an iono- 
spheric layer of a given density. Here, the 100- 
mc wave is not refracted sufficiently by the 
ionosphere and is not returned; the 5-mc and 
the 20-mc waves are returned, but the 20 mc 
wave, being refracted less than the 5-mc wave, 
returns at a greater distance. 

d. Incident Angles. For a radio wave of a 
particular frequency and for an ionized layer of a 
particular density of ionization, there is an angle 
of incidence of the wave, called the critical angle, 
at which the wave is reflected and returns to earth 
near its minimum or skip distance. This phe- 
nomenon is explained in detail in paragraph 22, 
but it should be noted that the critical angle of a 
given wave sometimes is denned as the angle at 
which the wave is propagated horizontally within 



99 




Figure 41. Frequency versus distance for returned waves. 



the ionospheric layer and, therefore, does not 
return to earth. On consideration, it will be seen 
that these two definitions are the same, since the 
angle at which the wave first returns and the angle 
at which it just does not return are the same 
angle. Also, the critical angle is measured (for 
purposes of calculation) between the wave path at 
incidence with the ionosphere and a line extended 



from the ionosphere to the center of the earth; 
however, for ease of explanation, the critical angle 
and all other angles of incidence are taken as 
angles made by the wave with either the earth or 
the ionosphere considered as horizontal plane 
surfaces. 

(1) Figure 42 shows a given wave at various 
angles of incidence with the ionosphere 



41 



Figure 4t. Incident wave paths for a plane earth and a plane ionosphere. 



— 



»MC* 



LOWEST ANOLt Of tttClDtNCC 



M feet 



Figure 43. High-frequency wave at 

and its resultant variation in refraction 
or reflection. Note that at angles of 
incidence larger than the critical angle, 
the wave is not sufficiently refracted in 
the ionosphere and escapes into space. 
As the angle of incidence decreases below 

42 



angles of incidence. 

the critical angle, the wave returns to 
earth at decreasing distances from the 
transmitter until a point of minimum 
distance, the skip distance, is reached. 
Then, as the angle of incidence continues 
to decrease, the distance between the 




TMeee-28 



Figure 44- Relation of angle of incidence to usable frequency. 



transmitter and the point at which the 
wave returns increases and continues to 
increase for smaller and smaller angles of 
incidence. Also, any high-angle wave 
which returns beyond the skip distance is 
attenuated greatly, and the skip distance 
remains as the point at which the wave 
first is returned in strength to the earth. 

(2) This irregular variation of the return 
distance with regular variation of the 
incident angle results from the fact that 
the ionosphere acts principally as a 
refracting medium for the larger angles 
of incidence. If the angle of radiation of 
the transmitted wave can be controlled, 
the smaller angles of radiation result in 
greater distance of communication. 
Figure 43 shows sky waves of a fixed 
frequency propagated at the critical 
angle and at various smaller angles. 
Note that the smaller the angle the 
greater the distance at which the wave is 
returned to earth. 

(3) The critical angle for a given frequency is 
not to be confused with the critical 
frequency for a given layer of the iono- 
sphere. The critical frequency, as ex- 
plained in paragraph 22, is the highest 
frequency a given density of ionization 
will return directly to the earth when 
propagated at a vertical angle (incident 



at 90° to the ionosphere). Although a 
vertically propagated frequency higher 
than . the critical frequency does not 
return to the earth, it is possible that this 
same frequency propagated at a different 
angle will return. In other words, fre- 
quencies higher than the critical fre- 
quency may be used if the angle of 
incidence is less than 90°. Figure 44 
illustrates this relationship between the 
angle of incidence and the use of fre- 
quencies higher than the critical fre- 
quency to obtain communication. 

28. Maximum Usable Frequency (MUF) 

a. For any given ionized layer of fixed height 
and ion density, and for a transmitting antenna 
with a fixed angle of radiation, there is a frequency 
(higher than any other) that will return to the 
earth at a given distance. This frequency is the 
maximum viable frequency for that distance; more- 
over, it is always a frequency higher than the 
critical frequency because the angle of incidence is 
less than 90°. Thus, for any given great-circle dis- 
tance along the earth, there is a maximum usable 
frequency which is the highest frequency that will 
be reflected from a given layer of the ionosphere 
and that will return to the earth at the great circle 
distance. If the distance between transmitter 
and receiver increases, the maximum usable fre- 



43 




Figure 45. Relationship of distance, time, angle, and frequency. 



quency increases. In other words, the greater the 
transmission distance, the higher the maximum 
usable frequency. 

b. In selecting the proper operating frequency 
for sky waves which travel «long a fixed radio path, 
the maximum usable frequency is perhaps the most 
important factor to be considered. If the oper- 
ating frequency is above the maximum usable fre- 
quency, the wave is said to escape, since it then 
will not be reflected by the ionosphere layer but 
will pass on through. On the other hand, if the 
operating frequency is decreased below the maxi- 
mum usable frequency in the daytime, the wave 
becomes increasingly attenuated, since, in the high- 
frequency range, the lower the frequency, the 
more wave energy is lost through ionospheric ab- 
sorption. Hence, it is usually desirable for trans- 
mission to occur on a frequency as near to the 
maximum usable frequency as possible. A direct 
relationship exists between the maximum usable 
frequency, the condition of the ionosphere, time, 
and the angle of radiation, as shown in figure 45. 
Thus, it is possible to predict mean values of maxi- 
mum usable frequency for propagation over any 



path for any time in any future month. Since the 
method of problem solution entails the use of 
world-contour charts and the use of complicated 
procedures, it is beyond the scope of this manual, 
but, as mentioned previously, this information 
may be obtained by consulting TM 11-499, Radio 
Propagation, and TB ll-499-( ), the monthly 
supplement thereto. 

c. If the density of the ionosphere is such that 
the maximum usable frequency is at a frequency 
near the critical frequency, the wave is exces- 
sively retarded in the ionized layer, and, because 
of the effects of the earth's magnetic field, splits 
into two components known as the ordinary wave 
and the extraordinary wave. These components, 
shown in figure 46, are usually of different polari- 
zation and phase. The critical frequency for 
the extraordinary wave is higher than that for 
the ordinary wave, the difference varying with 
the intensity of the earth's magnetic field, which 
changes with geographic position. Since the 
critical frequency for the ordinary wave is lower 
than that for the extraordinary wave, a layer of 
given ion density bends the ordinary wave less 



44 



HEIGHT OF MAXIMUM ION DENSITY 




TM 666-29 

Figure 46. Ordinary and extraordinary waves. 



than it does the extraordinary wave. From 
another point of view, the ordinary wave, which 
obeys the laws of simple refraction, must penetrate 
a greater distance into the layer than the extraor- 
dinary wave, which responds to both refraction 
and reflection. Figure 46 shows these two waves 
in conjunction with the MUF. However, it 
should be noted that this effect is important only 
for flayer transmission, causing interference fad- 
ing. The extraordinary wave reflected from the 
E layer is so weak that it does not affect radio re- 
ception. 

29. Lowest Useful Frequency (LUF) 

a. Absorption. The presence of ions in the 
upper atmosphere not only causes bending and 
the return to earth of a radio wave of sufficiently 
low frequency, but also causes part of the wave 
energy to be absorbed. The collisions of electrons 



with neighboring molecules of air reduce the 
intensity of the radio wave below that result- 
ing from the normal spreading of the wave 
front as it travels out from the transmitter. This 
absorption process is also of great importance in 
the practical use of ionospheric radio transmission. 
During the day, absorption takes place mainly in 
the D region of the ionosphere. Electron densi- 
ties in this region are considerably less than in the 
higher regions, but the increased density of the 
atmosphere itself results in an increase in the 
number of collisions between electrons and mole- 
cules of air, and more than compensates for the 
scarcity of electrons. During the night, ioniza- 
tion and absorption in the D region become negli- 
gible. However, there is some absorption for 
frequencies near the MUF of the F2 layer because 
waves at such frequencies are retarded to such an 
extent that there is sufficient time for appreciable 
energy loss to take place in spite of the relatively 



45 



small number of collisions. Such absorption is 
called deviative absorption, because it occurs in 
conjunction with retardation, which causes bend- 
ing of the waves. Absorption which takes place 
even though the wave is not appreciably retarded 
is called nondeviative absorption. The absorption 
in the D region is largely nondeviative. 

b. Lowest Useful High Frequency (LUHF). At 
certain frequencies of transmission, radio waves 
penetrating into the ionosphere, primarily in the 
D region and in the lower portion of the E region, 
lose some of their energy by absorption. Generally 
speaking, the higher the frequency used, up to the 
limit of the maximum usable frequency, the less 
will be the total absorption and the more satisfac- 
tory will be the level of communication. Absorp- 
tion is at a maximum for frequencies of about 500 
kc to 2 mc in the daytime, and decreases for both 
higher and lower frequencies at night. Thus, for 
frequencies above about 1 mc, the strength of the 
received sky waves will, in the daytime, increase 
with frequency (corresponding to decreasing ab- 
sorption). Finally, a frequency will be reached 
for any given sky-wave path where the strength 
of the received signal just overrides the noise level. 
This frequency is called the LUF. Frequencies 
lower than the LUF are absorbed to such an extent 
as to render them too weak for useful communica- 
tion. It should be noted, however, that the LUF 
depends on the power of the transmitter as well 
as on the distance concerned. At night, the noise 
level increases with decreasing frequency so that, 
as the frequency is lowered, the signals become 
weaker with respect to the noise and the LUF 
eventually is reached. Thus, the term lowest 
useful frequency may apply to either day or night 
transmission. 

c. Summary for Variable Frequency. Assuming 
constant ionospheric conditions, a constant dis- 
tance, and single-hop transmission, it can be said 
that — 

(1) Frequencies considerably below the MUF 
will be attenuated greatly by nondevia- 
tive absorption. 

(2) Frequencies somewhat below the MUF 
will be reflected as ordinary and extraor- 
dinary waves, either or both of which 
may be attenuated greatly by deviative 
absorption. 

(3) Frequencies near the MUF will be re- 
flected as ordinary and extraordinary 
waves, both of fair strength. 

(4) Frequencies at the MUF will fee received 



in the greatest possible strength as one 
wave. 

(5) Frequencies above the MUF will escape 
and not be received, except as scattered 
waves (par. 24). 
d. Summary for Variable Distance. Assuming 
constant ionospheric conditions, a fixed frequency, 
and single-hop transmission, it can be said that — 

(1) At short distances, the wave will skip and 
not be received, except of course as a 
ground wave. 

(2) At just a certain distance, called the skip 
distance, the wave will be received as one 
wave and at its greatest strength. 

(3) At a greater distance, the wave still will 
be received but as an ordinary wave and 
extraordinary waves, with resultant fad- 
ing because of random polarization. 

(4) At still greater distances, the ordinary and 
extraordinary waves will be received, but 
either or both will be attenuated consider- 
ably by deviative absorption. 

(5) At even greater distances, the wave will 
be attenuated greatly by both deviative 
and nondeviative absorption. 

Note. An important fact to be borne in mind 
is that radio waves of fixed radiation angle are 
receivable at distances greater than the skip 
distance, but that as this distance is increased 
appreciably, increased attenuation results. 

30. Optimum Working Frequency (FOT) 

Variations in the ion density of the ionosphere 
layers occur from day to day, and from hour to 
hour. Predictions on which the MUF's are based 
are made by averaging long-range observations, 
and do not take into account these day-by-day 
fluctuations. Therefore, the actual upper limiting 
frequency must be selected at a value which will 
insure against the probability of the operating 
frequency becoming greater than the MUF for 
any particular day. For the F2 layer, the optimum 
working frequency thus is selected at approxi- 
mately 85 percent of the MUF for that particular 
transmission path. The optimum working fre- 
quency for the combined E~F\ layer may be 
taken as the MUF, since the day-by-day variations 
in E layer ionization are small. Of course, if the 
LUF is nearly equal to the MUF for a given 
transmission path, the optimum working fre- 
quency must be selected at a value consistent 
with both. During moderate ionospheric storms, 
communication often can be assured by operating 



46 



at frequencies slightly lower than normal, since 
critical frequencies are usually lower than normal 
during these periods. 

31 . Received Signal Strength 

a. Factors. Whether the ionosphere will support 
transmission of sky waves over a given signal path 
at a certain time may be determined by finding 
the MUF and LUF for this path. If a consistent 
optimum working frequency can be derived from 
these factors, radio communication over this 
signal path is known to be possible. In the 
downcoming sky wave, we are not dealing with a 
steady wave of constant amplitude and phase, but 
one which may fade suddenly and greatly, whose 
polarization may be changing constantly, which 
may be composed of not one but many component 
waves, which is affected by reflection at the 
ground near the receiver, and which is subject to 
the variations in height and energy absorption in 
the ionosphere, and to focusing by the ionosphere. 
These difficulties may be minimized by due 
regard to certain factors upon which the received 
signal strength depends, such as transmitter 
power, antenna gain, transmission-path distance, 
absorption function of the signal path, and 
interference losses. It is obvious that the trans- 
mitter must supply the amount of power required 
to provide a field of sufficient strength at the 
receiver. 

b. Gain of Antenna. The gain of an antenna 
depends primarily on its design. The various 
types of antennas and their characteristics will be 
discussed in chapter 3, but it may be said here 
that transmitting antennas are designed for high 
efficiency in radiating energy, and receiving 
antennas are designed for the efficient pickup of 
energy. On many radio circuits, transmission is 
required between a transmitter and only one 
receiving station. In such cases, it is desirable to 
radiate as much energy as possible in the proper 
direction since radiated energy is useful only in 
that direction. Directional characteristics in a 
receiving antenna increase the energy pickup or 
gain in the favored direction and reduce the 
reception of unwanted noise and signals from 
other directions. The general requirements for 
receiving and transmitting antennas are that 
they have small energy losses and that they be 
efficient as receptors and radiators. 

c. Field Intensity. In traversing a nonionized 
region of the atmosphere, practically no energy 



is lost from the wave, and the only decrease in 
field intensity is that caused by the spreading out 
of the wave front, the inverse distance attenuation. 
The field intensity along a path, encountering no 
obstacles (neither large masses nor ions) and 
no interfering wave trains, varies inversely as the 
distance from the emitting source; the energy 
density in the waves, which is proportional to the 
square of the field intensity, varies inversely as 
the square of the distance (the familiar inverse- 
square law). As mentioned previously, the field 
intensity usually is measured in microvolts 
per meter. 

d. Absorption. The presence of ions in the 
upper atmosphere not only causes bending and the 
return to earth of a radio wave of sufficiently low 
frequency, but also causes part of the wave energy 
to be dissipated because of the collisions of the 
electrons with neighboring molecules of air. This 
reduces the intensity of the radio wave below that 
resulting from the normal spreading of the wave 
front as it travels out from the transmitter. This 
absorption process is of great importance in the 
practical use of ionospheric radio transmission. 
During the day, absorption takes place mainly 
in the D region of the ionosphere. Here, electron 
densities are considerably less than in higher 
regions, but the increased density of air molecules 
results in an increase in the number of collisions 
which more than compensates for the scarcity of 
electrons. During the night, ionization and ab- 
sorption in the D region become negligible. How- 
ever, there is still some absorption for frequencies 
near the MUF of the F 2 layer because waves at 
such frequencies are retarded, and there is suffi- 
cient time for appreciable energy loss to take place 
in spite of the relatively small number of collisions. 
As stated before, such absorption is called deviative 
absorption because it occurs in conjunction with 
retardation, which also causes bending of the 
waves. Absorption which takes place even when 
the wave is not appreciably retarded is called 
nondeviative absorption. Absorption in the D 
region is largely nondeviative. 

e. Antenna Height. The received signal field is 
usually a combination of the direct field resulting 
from the downcoming sky wave, together with 
that caused by the wave reflected from the ground. 
The resultant electric vector at the antenna, there- 
fore, is dependent on variations of the ground- 
reflection coefficient as well as on the instantaneous 
changes in both the amplitude and direction of 
the downcoming sky wave. The height of the 



47 



receiving antenna and the angle at which the sky 
wave approaches it may thus be contributing 
factors to the received signal strength, since 
polarization and phase of the ground-reflected 
component may serve either to cancel out, or to 
contribute to, the resultant field strength at the 
antenna. 

32. Fadins 

Because of fluctuations in iondspheric condi- 
tions, the received intensity of the sky wave is not 
constant, but varies with time. The term Jading 
refers to relatively rapid variations which occur 
during a space of minutes, seconds, or even frac- 
tions of a second. In general, fading is more 
sudden on high than on low frequencies. A type 
of fading known as selective jading also can cause 
distortion in radiotelephone signals. In such 
cases, the fading affects certain frequencies more 
than others and, therefore, may affect the side 
bands and the carrier wave differently. Fading, 
which is usually a nuisance, may be reduced by 
several methods, such as automatic volume con- 
trol, suppressed carrier transmission, and diversity 
reception. Discussion of these methods is beyond 
the scope of this manual. 

a. Types of Fading. The many types of fading 
fall into four principal classes — (1) interference 
fading, (2) polarization fading, (3) absorption 
fading, and (4) skip fading. Most of the rapid 
fading in the input to a receiver is a combination 
of the first two types; the other two are responsible 
for slower changes. 

b. Interference Fading. Interference fading is 
caused by phase interference of two or more waves 
from the same source arriving at the receiver 
over slightly different paths. If the paths are of 
different lengths, and their relative lengths vary 
for some reason, such as fluctuations in the height 
of the ionosphere layers, the relative phases of the 
waves arriving over the different paths vary with 
time, causing alternate reinforcement and cancel- 
ation of the field intensity. Because of irregu- 
larities in the ionosphere, one downcoming sky 
wave is really the summation of a great number of 
waves of small intensity and of random relative 
phases, and thus the resultant field intensity can 
vary over wide limits. The rms (root mean 
square) value of the fading intensity is equal to the 
homogeneous field, or the steady value of the field 
that would have existed had the ionosphere not 
broken the wave up into many components. 



c. rotanzation Fading. Additional variation in 
the field intensity affecting the receiving antenna 
occurs as a result of changes in the state of polari- 
zation of the downcoming wave relative to the 
orientation of the antenna. This variation is 
called polarization fading. In general, the state 
of polarization of the downcoming sky wave is 
changing constantly. This is due mainly to the 
combination, at random amplitudes and phases, 
of the two oppositely polarized components, the 
ordinary and the extraordinary wave. The polar- 
ization of the downcoming sky wave is generally 
elliptical. By elliptical polarization is meant that, 
as the wave travels along the signal path, the 
electric and magnetic fields remain at right angles 
to each other and to the direction of propagation, 
but rotate about the signal path in more or less 
corkscrew fashion instead of remaining constantly 
in either a vertical or a horizontal plane with 
respect to the path, as does the plane polarized 
wave. This results in random and constantly 
changing values of the amplitude and orientation 
of the electric field with respect to the receiving 
antenna. The state of polarization bf sky waves 
varies more rapidly the higher the frequency, 
which accounts in part for the rapid fading on the 
higher frequencies. 

d. Absorption Fading. Absorption fading is 
caused by short-time variations in the amount of 
energy lost from the wave because of absorption 
in the ionosphere. In general, the period of this 
type of fading is much longer than for the other 
two types, since the ionospheric absorption usually 
changes slowly. The sudder^ ionospheric disturb- 
ance is an extreme case of this type of fading, 
although usually it is classified as an irregular dis- 
turbance rather than as fading. Somewhat simi- 
lar to this type of fading, although not caused in 
the ionosphere but by reflections and absorption 
in objects close to the receiver, is the type of fading 
experienced in receiving a signal while moving 
along in an automobile. The fading out of the 
signal when the automobile is passing under a 
bridge or near a heavy steel structure is caused b} 7 
absorption of the wave's energy by the structure. 
Effects of this sort are involved in so-called dead 
spots or places where radio reception is particu- 
larly difficult. Also, radiation from wires, fences, 
and steel structures can cause an interference pat- 
tern that is relatively fixed in space, and can be 
noticed on moving the receiving equipment 
around. Where there are nearby structures which 



48 



can cause these effects, care must be exercised in 
the selection of the receiving site. 

e. Skip Fading. Skip fading is observed at 
places near the limit of the skip distance, and is 
caused by the changing angle of refraction. Near 
sunrise and sunset, when the ionization density of 
the ionosphere is changing, it may happen that the 
MUF for a given transmission path fluctuates 
about the actual operating frequency. When the 
skip distance moves out past the receiving station 
(sometimes called going into the skip) the received 
intensity abruptly drops by a factor of 100 or more, 
and just as abruptly increases again when the skip 
distance moves in again. This may take place 
many times before steady conditions for trans- 
mission are established. 

33. Radio Noise and Required Sisnal Strength 

a. Required Signal Strength. The minimum 
radio field intensity necessary to allow the satis- 
factory reception of an intelligible signal of a 
particular type in the presence of radio noise at 
the receiving station is called the required signal 
strength for this type of service. As a propaga- 
tion factor, the required signal strength is subject 
to wide variation. It depends on the receiving 
set; the local noise or static; the type of modulation 
of the radio wave, or, in other words, the type of 
service; and the grade of service desired — e. g., 
barely intelligible, high fidelity, and so on. It 
also varies, with the radio noise, according to the 
time of day and season. 

b. Types of Radio Noise. Radio noise may be 
defined as interference, the energy of which is not 
confined to a narrow band of frequencies. Two 
general types of radio noise may be distinguished — 
(1) impulse noise, which is interference resulting 
from a single elementary disturbance, or from an 
aggregate of elementary disturbances with sys- 
tematic relative phase; and (2) random, or fluctua- 
tion noise, which is the aggregate of a large number 
of elementary disturbances with random relative 
phases. A distinction between impulse and ran- 
dom noise is not always easy to make. However, 
electrical, or man-made, noise caused by the opera- 
tion of electrical equipment is usually of the 
impulse type, whereas atmospheric noise, originat- 
ing in thunderstorms or caused by other atmos- 
pheric conditions, ordinarily may be considered to 
have the bandwidth characteristics of random 
noise. The best example of random noise is the 
fluctuation noise originating in the resistance com- 



ponents of impedance elements in the receiver or 
brought about by the fluctuations of electrons 
within vacuum tubes. Another example is the 
noise generated by cosmic rays, which are suffi- 
cently high in frequency to penetrate the atmos- 
phere of the earth. This cosmic noise is noticeable 
only in receivers capable of detecting these 
frequencies. 

(1) Atmospheric noise. At the frequencies 
under consideration in this manual, at- 
mospheric noise and precipitation noise 
are the most important types to be 
considered. Radio noise from electrical 
apparatus, such as the ignition systems 
of automobiles, may be very serious, but 
is, more or less, under the control of the 
observer, and can be largely eliminated 
if necessary. Atmospheric or precipita- 
tion noise, on the other hand, since it 
originates in thunderstorms, or in rain, 
snow, or dust storms, usually cannot be 
eliminated and thus sets the limit for 
radio reception. Most atmospheric noise 
is considered to originate in the lightning 
flashes associated with thunderstorms. 
Generally, thunderstorms occur much 
more frequently over the land than over 
the sea and are more common at low 
than at high altitudes. 

(2) Cosmic and solar radio noise. Between 
frequencies of about 10 and 100 mc, 
cosmic radio noise originating in inter- 
stellar space is known to be the principal 
source of interference to reception under 
many circumstances. As stated above, 
cosmic noise has about the same char- 
acteristics as the fluctuation noise orig- 
inating in components of a receiving set. 
The sources of cosmic noise are not dis- 
tributed evenly over the sky but tend to 
be concentrated in several regions of the 
celestial sphere, the principal of these 
regions being near the center of the 
Milky Way. Consequently, when re- 
ceived on a directional antenna, the 
noise varies in characteristic manner 
from hour to hour and from day to day. 
The reason for the existence of cosmic 
noise is not well known. Some investiga- 
tors believe it to be radio-frequency 
radiation from eruptions, similar to the 
spot eruptions on our sun, occurring on 
all the stars in the galaxy; others have 



49 



considered it as originating in electron 
activity in the space between the stars. 
Recently, it has been found that the sun 
also acts as a radiator of radio noise at 
frequencies from about 200 mc up. 
Except at the time of large sunspot 
eruptions, solar noise is important only 
on very high frequencies and when highly 
directional antennas actually are pointed 
at the sun; therefore, it need not be con- 
sidered in relation to practical problems 
of propagation. 
(3) Receiving set noise. Noise generated in- 
ternally in a receiving set is caused by 
the random motion of electrons in resist- 
ance components of impedance elements 
and in the fluctuations of the electrons in 
vacuum tubes. In the absence of all 
external noise, signals, to be intelligible, 
must be strong enough to override this 
internal noise. With only internal noise 
present, the ability of a receiver to re- 
ceive a signal usually is expressed as the 
noise figure of the receiver. Experi- 
mental determination of the receiver 
input terminal voltage required to over- 
ride the internal noise in typical Army 
communications receivers shows a value 
of approximately 2 microvolts for 90 per- 
cent intelligibility of 100 percent modu- 
lated radiotelephony. Though this value 
is somewhat dependent on frequency, it 
is considered sufficiently accurate for all 
frequencies between 1.5 and 20 mc. 
c. Noise Figure. For many years, radio engi- 
neers were faced with the problem of devising a 
system for rating a receiver or an amplifier on its 
merits from the standpoint of low noise. The 
problem was complicated by the fact that in ad- 
dition to the useful output voltage of a generator 
(the generator, under operating conditions, being 
an antenna and the useful output voltage being 
the desired signal voltage) a certain noise voltage 
is always present. In an antenna, this noise 
voltage would include that caused by thermal re- 
sistor noise, and atmospheric and cosmic noise; in 
a standard voltage generator, this voltage would 
include only that resulting from thermal resistor 
noise. Because of the fluctuations of atmospheric- 
and cosmic-noise voltages with time, location, and 
construction and orientation of the antenna, these 
noise voltages do not offer a constant standard for 
rating a receiver or an amplifier. However, ther- 



mal noise, presenting a readily computed voltage 
offers a satisfactory standard against which the 
noise introduced by a receiver or an amplifier can 
be rated. Based on this principle, a system of 
rating a receiver in terms of its noise figure has 
been devised for this purpose. 

(1) In a receiving system, the total noise is 
the sum of the tube noise, the thermal 
noise in the input circuit, the thermal 
noise in the output circuit, and the 
antenna noise. Antenna noise is the 
induced atmospheric and cosmic noise 
appearing at the receiver input. 

(2) The signal-to-noise ratio of an ideal re- 
ceiving system can be expressed as 

available signal 
Signal-to-noise power _ power 
ratio of ideal system - ideal available 

noise power 

where the ideal available noise power is 
the power developed across the antenna 
resistance by the thermal noise voltage. 
The available signal power at the re- 
ceiver input is the power that the signal 
will develop across an input resistance 
equivalent to the antenna resistance. 
Noise figures usually are expressed in 
terms of power ratios or in db. 

(3) The noise figure of an actual receiver is 
obtained from the following ratio: 

signal-to-noise power ratio 

XT . for an ideal receiver 
Noise figure = -. rr : re- 
sign al-to-noise power ratio 

of an actual receiver 

(4) The required signal power at the input of 
an actual receiver is the required signal 
power for an ideal receiver multiplied by 
the receiver noise figure for the same 
signal-to-noise ratio. 

d. Types oj Modulation and Sermce Gain. Other 
factors upon which the required signal strength of 
a receiving system depends are known as type oj 
modulation and type oj service gain. Higher signal- 
to-noise ratios are required in commercial high- 
quality broadcast work than in many other types 
of service. On the other hand, in general code 
systems, such as automatic high-speed telegraphy 
or teletypewriter systems, the output signal-to- 
noise ratio need not be large, since the mechanism 
operates when the signal exceeds the noise by only 



■50 



a small margin. The gain required for a certain 
type of service is the relative signal strength re- 
quired for that type of communication as com- 
pared with the signal required for a reference type. 



This reference type of service corresponds to 90 
percent intelligibility of speech and is comparable 
to the grade of service known as order wire in 
telephonic communications. 



Section IV. SUMMARY AND REVIEW QUESTIONS 



34. Summary 

a. Ground-wave propagation refers to those 
types of radio transmission which do not make use 
of ionospheric reflections. 

b. The direct-wave component travels directly 
from the transmitting to the receiving antenna. 

c. The ground-reflected component undergoes a 
phase reversal of 180° upon reflection from the 
ground. 

d. This phase reversal may cause serious signal- 
voltage cancelation between the ground-reflected 
and the direct-wave components. 

e. The surface-wave component is affected pri- 
marily by the conductivity and dielectric constant 
of the earth. 

j. The surface-wave component is essentially 
vertically polarized at appreciable distances from 
the antenna. 

g. The tropospheric-wave component is re- 
fracted in the lower atmosphere by sharp changes 
in density and humidity of the air. 

k. One of the common causes of tropospheric 
refraction is temperature inversion. 

i. Trapped waves may follow the curvature of 
the earth for distances far beyond the optical 
horizon of the transmitter. 

j. The frequency characteristics of the ground 
wave determine what particular component will 
prevail along any given signal path. 

k. For frequencies above 30 mc, the distance 
range of the ground wave can be increased by 
increasing the antenna height as well as by 
increasing the radiation power. 

/. The ionosphere is composed of one or more 
electrically conducting layers which bend radio 
waves back toward the earth. 

m. The ionosphere layers are formed by ioniza- 
tion of the gas molecules composing them. 

n. Recombination goes on constantly, so that 
an ionized layer does not necessarily last indef- 
initely. 

o. The chief cause of ionization of the ionosphere 
is ultraviolet radiation from the sun. 

p. Sunspots have the effect of increasing the 
ionization of ionized layers. 



q. Ionization occurs in different layers, depend- 
ing on the frequency of the ultraviolet radiation 
causing it, and on the critical density of the 
atmosphere. 

r. Although the number of layers is subject to 
variation from time to time, there are usually four 
distinct layers during the daytime. 

s. During the nighttime, only one ionized 
layer — the F layer — usually exists. 

t. The D region is the lowest layer, and it is 
chiefly important for its absorption effects. 

u. The E layer is important for reflection of 
radio waves up to about 20 mc. 

v. For transmission above 20 mc, the F, Fl, and 
F2 layers are most important. 

w. The virtual height, or apparent height, of an 
ionized layer is considerably greater than the 
actual layer height. 

x. The chief factor that controls long-distance 
communication is the ionization density of the 
ionized layer. 

y. The higher the frequency of transmission, the 
greater must be the density of ionization to reflect 
waves back to earth. 

z. The critical frequency is the highest frequency 
at which waves sent vertically upward are re- 
flected directly back to earth. 

aa. The upper layers are the most highly ionized 
and, therefore, they reflect the higher frequencies. 

ab. Waves of all frequencies higher than the 
critical frequency are not reflected back to earth, 
but are said to escape. 

ac. Changes in the sun's state of activity which 
cause variations in the amount of its radiation will 
result in variations in the conformation of the 
ionosphere. 

ad. Regular variations can be predicted, and 
fall into four classes: diurnal, seasonal, 11-year, 
and 27-day. 

ac. Diurnal variation is caused by the rotation 
of the earth and results in higher intensities of 
ionization during the daytime. 

aj. Seasonal variation causes shifts in the maxi- 
mum ion density in the D, E, and F layers, being 
greater in summer than in winter. 

51 



ag. The ion density of the F2 layer, however, is 
much greater in winter than in summer. 

ah. The 11-year variation is caused by the cycle 
of sunspot activity which rises to a maximum 
approximately every 11 years and decreases to a 
minimum in the intervening years. 

ai. At times of sunspot maxima, higher fre- 
quencies may be used generally for communica- 
tions over long distances. 

aj. The 27-day variation results from the rota- 
tion of the sun about its axis. 

ak. The sporadic E is an ionized cloud which 
appears at indefinite intervals and at a slightly 
higher level than the normal E layer. 

al. Sudden ionospheric disturbances are the 
cause of sudden radio fadeouts. 

am. During ionospheric storms, there are large 
variations from normal of critical frequencies, 
layer heights, and absorption. 

an. Scattered reflections may cause signal dis- 
tortion and so-called flutter Jading. 

ao. Sky-wave transmission is possible because of 
reflections from ionosphere layers. 

ap. The skip distance is the shortest distance 
from the transmitter at which radio waves of a 
given frequency will be reflected back to earth. 

aq. The skip zone depends upon the extent of 
ground-wave range and disappears entirely if the 
ground-wave range equals or exceeds the skip 
distance. 

ar. Signal paths involving one and two reflec- 
tions from the ionosphere are called, respectively, 
single- and double-hop modes of transmission. 

as. Paths which radio waves normally traverse 
in traveling from transmitter to receiver are 
usually directly above great-circle paths. 

at. The MUF is the highest sky-wave frequency 
that is usable for a particular radio circuit at a 
particular time. 

au. The greater the transmission distance, the 
higher may be the MUF. 

av. The chief effect of the extraordinary wave 
on communications is to cause severe interference 
fading. 

aw. The LUF is the lower limiting frequency for 
satisfactory sky-wave communication for a radio 
circuit at a particular time. 

ax. The LUF is determined by the strength of 
the sky-wave signal in relation to that required to 
overcome noise. 

ay. The sky-wave field intensity is equal to the 
required field intensity at the LUF. 

az. For the F2 layer, the optimum working fre- 



quency usually is selected at approximately 85 
percent of the MUF for the particular signal path. 

ba. The received signal strength depends upon 
such factors as transmitter power, antenna gain, 
transmission path distance, absorption function, 
and interference losses. 

bb. The gain of an antenna depends primarily 
upon its design. 

6c. The general requirements for receiving and 
transmitting antennas are that they have small 
energy losses and are efficient receptors or 
radiators. 

bd. The free-space electric field intensity is 
inversely proportional to the distance from the 
transmitter. 

be. Absorption that takes place even when the 
wave is not appreciably retarded in the ionosphere 
is called nondevlatioe absorption. 

bf. Deviative absorption occurs in conjunction 
with retardation which also causes bending of the 
waves. 

bg. The height of the antenna and the angle 
at which the sky wave approaches it may be con- 
tributing factors to the received signal strength. 

bh. Phase interference of two or more waves 
from the same source arriving at the receiver by 
different paths is called interference fading. 

bi. Changes in the state of polarization of down- 
coming sky waves relative to the orientation of the 
antenna is called polarization fading. 

bj. Absorption fading is caused by short-time 
variations is the amount of energy lost from the 
wave absorption in the ionosphere. 

bk. Skip fading is caused by waves alternately 
escaping and returning to earth. 

bl. The required field strength is the minimum 
radio field intensity necessary to the satisfactory 
reception of an intelligible signal. 

bm. Radio noise may be classified as impulse 
noise, and random, or fluctuation, noise. 

bn. Most atmospheric noise is considered to 
originate in lightning flashes associated with 
thunderstorms. 

bo. Cosmic noise originates in outer space and 
usually affects reception at frequencies of from 
10 to 100 mc. 

bp. Receiving set noise is caused by the random 
fluctuations of electrons in resistance components 
and in vacuum tubes. 

bq. Radio receivers may be rated according to 
their noise figure. 

br. The noise figure is equal to the ratio between 



52 



the signal-to-noise ratio for an ideal receiver and 
that for an actual receiver. 

6s. Different types of service require different 
values of signal-to-noise ratio for satisfactory 
operation. 

35. Review Questions 

a. What are the factors affecting ground-wave 
propagation? 

6. What are the separate components of the 
ground wave? 

c. What happens to the ground-reflected com- 
ponent upon reflection from the earth's surface? 

d. What may be done to reduce the signal- 
voltage cancelation caused by the ground-re- 
flected component arriving at the receiver out of 
phase with the direct-wave component? 

e. What affects the surface-wave component? 
/. What is meant by temperature inversion? 

g. What is the normal range of frequency for 
ground waves? 

h. Describe trapped waves. 

i. How are the ionosphere layers formed? 

j. What is meant by ionization? By recombi- 
nation? 

k. What is the principal cause of ionization? 
I. What effects do sunspots have on the ion 
layers? 

m. What is meant by Bellinger fade? 
n. Name the ionosphere layers and their relative 
heights. 

o. What general effect does the D region have 
on hf waves? 

p. What happens to the various layers during 
the night? 

q. Are the ionosphere layers limited to any 
given number? 

r. How does refraction take place in the iono- 
sphere? 

s. Describe virtual height. 

t. Define critical frequency. 



u. Briefly describe the regular variations of the 
ionosphere. 

v. What is sporadic Ef 

w. What usually happens to radio communica- 
tions during severe ionospheric storms? 

x. What are scattered reflections? 

y. Give the factors affecting sky-wave propa- 
gation. 

z. What is the skip zone? 

aa. What is the difference between the skip 
zone and the skip distance? 

ab. Describe single- and double-hop radio paths. 

ac. What is meant by short-path and long-path 
transmission? 

ad. Define maximum usable frequency. 

ae. How does variation in the oblique angle of 
incidence affect the MUF? 

of. How does the extraordinary wave affect 
communications? 

ag. What happens to waves of frequencies 
greater than the MUF? 

ah. What is meant by the lowest usable high 
frequency? 

ai. What happens to waves of frequencies lower 
than the LU"^ 

aj. What jmum working frequency usually 
is selected for F2 layer propagation? 

ak. What is received signal strength? 

al. What are the general requirements for 
receiving and transmitting antennas? 

am. What is the cause of absorption? 

an. How does the height of the receiving anten- 
na affect reception? 

ao. What is meant by interference fading? By 
polarization fading? 

ap. When does skip fading usually occur? 

aq. Upon what factors does reguired signal 
strength depend? 

ar. What is the cause of atmospheric noise? 
Cosmic noise? Receiving set noise? 

as. What is the noise figure of a radio set? 

at. How does the type of service affect the 
required signal-to-noise ratio? 



53 



CHAPTER 3 

HALF-WAVE AND QUARTER-WAVE ANTENNAS 



Section I. BASIC THEORY 



36. Introduction 

The electric and magnetic fields radiated from 
an antenna form the electromagnetic field, and 
this field is responsible for the transmission and 
reception of electromagnetic energy through free 
space. An antenna, however, is also part of the 
electrical circuit of a transmitter or a receiver and, 
because of its distributed constants, it acts as a 
circuit containing inductance, capacitance, and 
resistance. Therefore, it can be expected to dis- 
play definite voltage and current relationships in 
respect to a given input. A current through it 
produces a magnetic field, and a charge on it pro- 
duces an electric field. These two fields taken 
together form the induction field. To gain a 
better understanding of antenna, theory, a review 
of the basic electrical concepts of voltage and 
electric field, and current and magnetic field is 
necessary. 

37. Voltage and Electric Field 

a. Electric Field. 

(1) When a capacitor is connected across a 
source of voltage, such as a battery (fig. 
47), it is charged some amount, depend- 
ing on the voltage and the value of 
capacitance. Because of the emf (elec- 
tromotive force) of the battery, negative 
charges flow to the lower plate, leaving 
the upper plate positively charged. Ac- 
companying the accumulation of charge 
is the building up of the electric field. 
The flux lines are directed from the 
positive to the negative charges and at 
right angles to the plates. 

(2) If the two plates of the capacitor are 
spread farther apart, the electric field 
must curve to meet the plates at right 
angles (fig. 48). The straight lines in A 
become arcs in B, and approximately 
semicircles in C, where the plates are in a 



CAPACITOR 







C\ /T\ /T\ /T\ /TwrTI 


L 

r , 


d 






r i 


y 


r i 


r ^ 


r i 


r 




i 





TM 666-52 

Figure 47. Charges on plates of a capacitor. 

straight line. Instead of flat metal 
plates, as in the capacitor, the two ele- 
ments can take the form of metal rods or 
wires. The three-dimensional view in 
figure 49 depicts the electric field more 
accurately. In A of figure 49 the wires 
are approximately 30° apart, and the 
flux lines are projected radially from the 
positively charged wire to the negatively 
charged wire. In B of figure 49 the two 
wires lie in a straight line, and the flux 




TM 666-33 



Figure 48- Electric field between plates at various angles. 



54 



A B 

tm «ee-S4 

Figure 49. Electric field between wires at various angles. 



lines form a pattern similar to the lines of 
longitude around the earth. To bring 
out the picture more clearly, only the 
lines in one plane are given. 
b. Voltage. 

(1) Assume that the sphere marked E in B 
of figure 49, is a transmitter supplying 
r-f energy. The two wires then can 
serve as the antenna for the transmitter. 
R-f energy is radiated from the antenna 
and charges move back and forth along 
the wires, alternately compressing and 
expanding the flux lines of the electric 
field. The reversals in polarity of the 
transmitter signal also reverse the direc- 
tion of the electric field. 

(2) When a charge is put on the plates of a 
capacitor by means of a battery, an 
electric field is set up between its plates. 
The flow of charge from source to capaci- 
tor ceases when the capacitor is fully 
charged, and the capacitor is said to be 
charged to a voltage equal and opposite to 
that of the source. The charged capaci- 
tor can be used as a source of emf since 
it stores energy in the form of an electric 
field. This is the same as saying that 
an electric field indicates voltage. The 
presence of an electric field about an 



antenna also indicates voltage. Since 
the polarity and the amount of charge 
depend on the nature of the transmitter 
output, the antenna voltage also depends 
on the energy source. For example, if a 
battery constitutes the source, the an- 
tenna charges to a voltage equal and 
opposite to that of the battery. If r-f 
energy is supplied to a half -wave antenna, 
the voltage across the antenna lags the 
current by 90°. The half -wave antenna 
acts as if it were a capacitor, and it can 
be described as capacitive. 

38. Current and Magnetic Field 

a. Current. A moving charge along a conductor 
constitutes a current and produces a magnetic 
field around the conductor. Therefore, the flow 
of charge along an antenna also will be accom- 
panied by a magnetic field. The intensity of this 
field is directly proportional to the flow of charge. 
When the antenna is uncharged, the current flow 
is maximum, since there is no opposing electric 
field. Because of this current flow, a charge 
accumulates on the antenna, and an electric field 
builds up in increasing opposition to the emf of 
the source. The current flow decreases and when 



55 



DIRECTION OF 
.CURRENT FLOW 




DIRECTION OF MAGNETIC 
FIELD 



LEFT HAND 



TM 666-5S 

Figure 50. Magnetic field about a half-wave antenna. 

the antenna is fully charged, the current no longer 
flows. 

6. Magnetic Field. The magnetic field in the 
space about a current-carrying device has a specific 



configuration, with the magnetic flux lines drawn 
according to a definite rule (fig. 50). Whereas, 
in the electric field, the electric lines are drawn 
from a positive charge to a negative charge, in the 
magnetic field, the flux lines are drawn according 
to the left-hand rule. The direction of current 
.flow is upward along both halves of the antenna. 
The lines of magnetic flux form concentric loops 
which are perpendicular to the direction of current 
flow. The arrowheads on the loops indicate the 
direction of the field. If the thumb of the left 
hand is extended in the direction of current flow 
and the fingers clenched, then the rough circles 
formed by the fingers indicate the direction of the 
magnetic field. This is the left-hand rule, or 
convention, which is used to determine the direc- 
tion of the magnetic field. 

39. Combined Electric and Magnetic Fields 

a. When r-f energy from a transmitter is 
supplied to an antenna, the effects of charge, 
voltage, current, and the electric and magnetic 
fields are taking place simultaneously. These 
effects (fig. 51) have definite time and space 
relationships to each other. If a half-wave 



oz 

— Ul 
o ± 

UJ 



a 



2af 

I 




COUNTER" 
h-CLOCKWISE 
FLUX 
LINES 



56 



TM 666-S6 

Figure 51. Electric and magnetic fields 90° out of -phase. 



antenna is used, the relations between charge and 
current flow can be predicted because of the 
capacitive nature of this antenna. The voltage 
will lag the current by 90° and the electric and 
magnetic fields will be 90° out of phase. With 
no electric field present (no charge), the current 
flow is unimpeded, and the magnetic field is 
maximum. As charge accumulates on the an- 
tenna, the electric field builds up in opposition 
to current flow and the magnetic field decreases 
in intensity. When the electric field reaches its 
maximum strength, the magnetic field has decayed 
to zero. 

b. A reversal in polarity of the source reverses 
the direction of current flow as well as the polarity 
of the magnetic field, and the electric field aids 
the flow of current by discharging. The magnetic 
field builds up to a maximum, and the electric 
field disappears as the charge is dissipated. The 
following half-cycle is a repetition of the first 
half-cycle, but in the reverse direction. This 
process continues as long as energy is supplied 
to the antenna. The fluctuating electric and 
magnetic fields combine to form the induction 



field, in which the electric and magnetic flux 
maximum intensities occur 90° apart in time, or 
in time quadrature. Physically, they occur at 
right angles to each other, or in space quadrature. 
To sum up, the electric and magnetic fields about 
the antenna are in space and time quadrature. 

40. Standing Waves 

a. The Infinitely Long Conductor. Assume that 
it is possible to have a wire conductor with one 
end extending infinitely, with an r-f transmitter 
connected to this wire. When the transmitter 
is turned on, an r-f current in the form of sine 
waves of r-f energy moves down the wire. These 
waves of energy are called traveling waves. The 
resistance of the conductor gradually diminishes 
the amplitude of the waves, but they continue 
to travel so long as the line does not come to an 
end. 

6. The Finite Conductor {Antenna). The an- 
tenna, however, has some finite length. There- 
fore, the traveling waves are halted when they 
reach the end . of the conductor. Assume that 




c 

TM 666-S7 

Figure 52. Traveling waves on an antenna and typical resultant wave. 



57 



the r-f transmitter is turned on just long enough 
to allow one sine wave of energy to get on the 
line (A of fig. 52) . This traveling wave is moving 
down the antenna toward the end. When this 
wave reaches the end of the conductor, the 
current path is broken abruptly. With the 
stoppage of current flow, the magnetic field 
collapses. A voltage is induced at the end of the 



RESULTANT 



conductor that causes current to flow back toward 
the source, as in B of figure 52. The wave is 
reflected back to the source, and, if a continual 
succession of waves is sent down the line, they will 
be reflected in the same continual pattern. The 
wave moving from the transmitter toward the 
end is called the incident wave, and its reflection 
is called the reflected wave. 




RESULTANT 




RESULTANT 




RESULTANT 




RESULTANT 




TM 666- 58 

Figure 53. Development of standing wave from traveling waves. 



58 



c. Standing Waves. 

(1) A continuous flow of incident waves 
results in a continuous flow of reflected 
waves. Since there is only one con- 
ductor, the two waves must pass each 
other. Electrically, the only current 
that actually flows is the resultant of 
both of these waves. The waves can rein- 
force or cancel each other as they move. 
When they reinforce, the resultant wave 
is maximum; when they cancel, the 
resultant wave is minimum. In a con- 
ductor which has a finite length, such as 
an antenna, the points at which the 
maxima and minima of the resultant 
wave occur (C of fig. 52) are stationary. 
In other words, the maximum and 
minimum points stand still, although both 
the incident and reflected waves are 
moving. The resultant wave stands still 
on the line, only its amplitude being 
subject to change. Because of this 
effect, the resultant is referred to as a 
standing wave. 

(2) The development of the standing wave 
on an antenna by actual addition of the 
traveling waves is illustrated in figure 53. 
At the instant pictured in A, the incident 
and reflected waves just coincide. The 
result is a standing wave having twice 
the amplitude of either traveling wave. 
In B, the waves move apart in opposite 
directions, and the amplitude of the 
resultant decreases, but the points of 
maximum and minimum do not move. 
When the traveling waves have moved 
to a position of 180° phase difference, the 
resultant is zero along the entire length 
of the antenna, as shown in C. At this 
instant there can be no current flow in 
the antenna. The continuing move- 
ment of the traveling waves, shown in 
D, builds up a resultant in a direction 
opposite to that in A. The in-phase 
condition of the traveling waves results 
in a standing wave, in E, equal in ampli- 
tude, but 180° out of phase with the 
standing wave in A. 

(3) If the progressive pictures of the standing 
wave are assembled on one set of axes, 
the result is as in figure 54. The net 
effect of the incident and reflected waves 
is apparent. The curves are lettered 




NODE 



TM 666-89 

Figure 64- Standing waves. 

with reference to figure 53 . As the travel- 
ing waves move past each other, the 
standing wave changes only its ampli- 
tude. The fixed minimum points are 
called nodes, and the curves representing 
the amplitude are called loops. 
(4) The concept of the standing wave can be 
applied to the half-wave antenna with 
reference to either current or voltage 
distribution at any instant. This appli- 
cation is possible because there are 
traveling waves of both voltage and 
current. Since voltage and current are 
out of phase on the half-wave antenna, 
the standing waves also are found to be 
out of phase. 

41 . Voltage and Current Distribution on Half- 
Wave Antenna 

a. Instantaneous Voltage and Current. 

(1) When an r-f transmitter is feeding a 
half-wave antenna, positive and negative 
charges move back and forth along the 
antenna (figs. 55 and 56). The first 
picture shows the position of the charges 
at some arbitrary time, TO. The r-f 
charges being observed are at the ends 
of the antenna, and there is a maximum 
difference in potential between the ends, 
A and B. The remaining illustrations 
show the instantaneous positions of the 
charges at regular intervals of 22.5° 
throughout a complete cycle. 

(2) To the right of each instantaneous posi- 
tion of the charges are curves representing 

59 



DISTRIBUTION CURVES 
VOLTAGE CURRENT 




TM 666-60A 

Figure 55. Voltage and current distribution in terms of positive and negative charges. 

60 



MOVEMENT OF POINTS OF CHARGE 
(CONTINUED) 



A 

T8 g 



-180- 
-180°- 



T9 



ens: 



5LD 



202.5-*- 



-202.5 



T«0 ( ) {-) 



ft ) 



-225- 



-225- 



T.. £ 



'r) "1 



-247.5- 



247.5- 



T.2 (J 



-270- 



-270- 



TI3 



-ii 



-292.5 



292.5 



TI4 



M 



r) ) 



315° 



315- 



TI5 



(TIT 



337.5" 



-337.5 



TI6 



q: 



3 



360 - 
-360° 



DISTRIBUTION CURVES (CONTINUED) 



VOLTAGE 







CURRENT 






TM 666-60B 



Figure 56. Voltage and current distribution in terms of positive and negative charyes. 



*1 



the current and voltage at that particular 
time for any point on the antenna. For 
example, at time TO, the positive and 
negative charges are at points A and B 
on the antenna. The voltage between 
these points represents a maximum dif- 
ference of potential. The current, being 
90° out of phase in respect to the voltage, 
is everywhere zero. These distribution 
curves are standing waves derived in the 
same manner as those discussed in the 
previous paragraph. 

(3) The next illustration shows the position 
of the charges at time Tl . The standing 
wave of current is a relative maximum 
at the center of the antenna. This 
current loop has nodes which remain at 
the ends of the antenna, and it is, there- 
fore, 90° out of phase with the standing 
wave of voltage. 

(4) At T2 and T3, the charges move closer 
together, and the standing wave of 
voltage slowly decreases in amplitude. 
Conversely, the current loop increases 
in magnitude. When the charges meet 
after 90° of the r-f cycle (T4), the effect 
is that of having the positive and negative 
charges cancel . The voltage loop accord- 
ingly is zero everywhere on the antenna, 
and the current loop rises to its maximum 
value, unimpeded by any charge on the 
antenna. 

(5) At time T5, the charges have passed each 

other, each charge having moved past the 
center point of the antenna. The polarity 
of the voltage loops is reversed, and they 
build up in the opposite direction, keep- 
ing the node always at the center point of 
the antenna. The reversal of polarity is 
shown in the charge positions at T3, T4, 
and T5. The separation of the charges 
also is accompanied by a decrease in the 
amplitude of the current loop. 

(6) From T5 to T8, the charges move out to 

the ends of the antenna. During this 
time, the voltage loops increase and the 
current loops decrease in amplitude. At 
time T8, which occurs 180° after TO in 
the r-f cycle, the charges have moved to 
opposite ends of the antenna. Compare 
the picture in TO to the picture in T8. 
It is seen that the negative charge is now 
at point A and the positive charge at 



point B. Since the positions of the 
charges have been reversed from TO to 
T8, the voltage loops in T8 are 180° out 
of phase compared with the loops in TO. 
(7) From T8 to Tl6 in figure 56, the move- 
ment of the charges is shown in the 
opposite direction, the current loop 
reaching a maximum at T12. When the 
entire r-f cycle is completed at time T16, 
the charges have returned to the posi- 
tions they occupied at TO. The distri- 
bution curves of voltage and current also 
are in their original conditions. The 
entire process then is repeated for each 
r-f cycle. 

6. Standing Waves oj Voltage and Current. 

(1) The distribution curves of the current and 

voltage are standing waves. This means 
that they are the resultants obtained by 
adding two traveling waves. The two 
traveling waves are associated with the 
positive and negative charges. The wave 
caused by the negative charge can be 
called the incident wave, and the wave 
caused by the positive charge the re- 
flected wave. The discussion, however, is 
clearer when the concept of negative and 
positive charges is used. 

(2) The positive charge above, taken at time 

TO in figure 55, produces a traveling wave 
of voltage, shown by the dashed line in A, 
figure 57. The negative charge at the 
opposite end of the antenna produces an 
identical traveling wave (dash-dot curve). 
These two add together to produce the 
TO voltage distribution curve, which is 
the resultant wave of A of figure 55. 
Both of these waveforms are identical, 
being the standing wave of voltage at 
time TO. All the following distribution 
curves of figure 57 are produced in the 
same manner. They are the standing- 
wave resultants caused by the traveling 
waves accompanying the charges. 

(3) In B of figure 57, each of the traveling 
waves has moved 45°, the positive travel- 
ing wave moving to the right and the 
negative traveling wave moving to the 
left. This time corresponds to T2 in 
figure 55. The standing wave produced 
corresponds to the voltage distribution 
curve at T2. The standing waves of cur- 
rent are produced in the same manner. 



AO 



180' 




B AO 



45» 



135' 



180* 



AO 




A 




t80» 



B 



A 



45* 



90* 135* ^080* 

■ i ,> B AO 




J 80* 



VOLTAGE WAVES 

IN A.B.ANDC 

VOLTAGE WAVE DUE TO POINT OF + CHARGE 

VOLTAGE WAVE DUE TO POINT OF - CHARGE 

RESULTANT VOLTAGE WAVE 



CURRENT WAVES 
IN D.E.AND F 

CURRENT WAVE DUE TO POINT 0F+ CHARGE 

CURRENT WAVE DUE TO POINT OF - CHARGE 

RESULTANT CURRENT WAVE 

TM ««»-6l 



Figure 67. Standing waves of voltage and current. 



63 



VOLTAGE DISTRIBUTION 

A 



(9 

2. o. 




J TO Tl T2 T3 T4\T5 T6 T7 T8 T9 TIO Til ,ft\Z TI3 TI4 TI5 TI6 



VOLTAGE AT A 
VOLTAGE AT Y 



B 

TM 666-62 



Figure 58. Standing waves of voltage at a point on the antenna. 



The current curves at D, E, and F of 
figure 57 correspond to times TO, T2, 
and T4 of figure 55. 
c. Standing Waves of Voltage. 

(1) In A of figure 58, voltage standing waves 
occurring at different times are brought 
together on one axis, AB, representing a 
half-wave antenna. Essentially, these 
are the same curves shown progressively 
in figures 55 and 56 as voltage distribu- 
tion curves. They can be used to deter- 
mine the voltage at any point on the 
antenna at any instant of time. For 
example, if it is desired to know the 
variations of voltage occurring at point 
Y on the antenna over the r-f cycle, the 
variations are graphed in respect to 
time, as shown in B of figure 58. At TO 
the voltage at Y is maximum. From 
TO through T3 the voltage decreases, 
passing through zero at T4. The voltage 
builds up to a maximum in the opposite 



direction at T8, returning through zero 
to its original position from T8 to T16. 
(2) Between TO and Tl6, therefore, an entire 
sine-wave cycle, Y, is reproduced. This 
is true also of any other point on the an- 
tenna with the exception of the node at 
X. The peak amplitude of the sine 
wave produced at any point depends on 
its position on the antenna. The nearer 
the point is to either end, the greater its 
peak amplitude. 
d. Standing Waves oj Current. The standing 
waves of current occurring at various times 
through the r<-f cycle are assembled on a single 
axis in figure 59. This axis, AB, represents the 
half-wave antenna. If the current variations at 
point Y from TO to Tl6 are graphed in respect to 
time, the result is the sine wave in B of figure 59. 
This is true for any point along the antenna with 
the exception of the nodes at the ends. The 
current has its greatest swing at X, the center of 
the antenna. Comparison of the voltage varia- 



64 



TI2 TI2 



CURRENT DISTRIBUTION 

A 



R 
R 
E 
N 
T 



TO Tl T2 T3 T4 T5 T6 T7 T8V T9 TIO Til TI2 TI3 TI4 TIS 



r TI6 



CURRENT AT X 

CURRENT AT Y B 

TM 666-63 

Figure 59. Standing waves of current at a point on the antenna. 



tion curve (A of fig. 58) with the current variation 
curve (A of fig. 59) shows the voltage curve lead- 
ing the current curve by 90° at Y. This relation 
can be expected on any half-wave device. 

e. Measurement of Standing Waves. In figure 
60, the standing waves of voltage E, and current 
/, are indicated along the antenna. There are 
current nodes at A and B and a voltage node at X. 
These standing waves are found on any half-wave 
antenna. A meter that indicates the effective 
value (0.707 of peak) of the a-c signal can be used 
to measure the standing waves present on the 
half-wave antenna. 




A X B 



Figure 60. Standing waves as measured with a meter. 



42. Velocity of Propagation and Antenna 
Length 

a. In free space, electromagnetic waves travel 
at a constant velocity of 300,000 kilomete s or 
186,000 miles per second. The r-f energy on an 
antenna, however, moves at a velocity consider- 
ably less than that of the radiated energy in free 
space because the antenna has a dielectric constant 
greater than that of free space. Since the dielec- 
tric constant of free space (air or vacuum) is 
approximately 1, a dielectric constant greater 
than 1 retards electromagnetic-wave travel. 

b. Because of the difference in velocity between 
the wave in free space and the wave on the an- 
tenna, the physical length of the antenna no longer 
corresponds to its electrical length. The antenna 
is a half-wavelength electrically, but somewhat 
shorter than this physically. This is shown in 
the formula for the velocity of electromagnetic 
waves, 

V=fk 



65 



where V is the velocity, / is the frequency, and X 
is the wavelength. Since the frequency of the 
wave remains constant, a decrease in the velocity 
results in a decrease in the wavelength. There- 
fore, the wave traveling in an antenna has a 
shorter wavelength than the same wave traveling 
in free space, and the physical length of the 
antenna can be shorter. 

c. The actual difference between the physical 
length and the electrical length of the antenna 
depends on several factors. A thin wire antenna, 
for example, has less effect on wave velocity than 
an antenna with a large cross section. As the 
circumference of the antenna increases, the wave 
velocity is lowered as compared with its free-space 
velocity. The effect of antenna circumference 
on wave velocity is illustrated in the graph of 
figure 61. 




.001' 1 1 1 1 1 1 1 1 1— LI 1 1 

.2 A .6 .8 10 

WAVE VELOCITY 

FREE- SPACE VELOCITY 

TM 666-69 

Figure 61. Effect of antenna circumference on wave velocity. 

d. Other factors are involved that lower wave 
velocity on the antenna. Stray capacitance, for 
example, increases the dielectric constant and 
lowers wave velocity. This capacitance can be 
caused by the line connecting the antenna to the 
transmitter, the insulators used to give physical 
support to the antenna, or nearby objects made 



of metallic or dielectric materials. The change 
in velocity resulting from stray capacitance is 
called end effect because the ends of the antenna 
are made farther apart electrically than they are 
physically. End effect is counteracted by making 
the physical length about 5 percent shorter than 
the electrical length, as expressed in the formula 

1=0.95(492//) 
=468// 

where L is the physical length in feet and / is the 
frequency in megacycles. This formula is ac- 
curate for all practical purposes in determining 
the physical length of an antenna 1 half-wave- 
length at the operating frequency. 

e. The capacitive end effect also changes 
slightly the standing waves of voltage and current. 
When the standing waves are measured, it is 
found that the nodes have some Value and do not 
reach zero, because some current is necessary to 
charge the stray capacitance. The standing 
waves measured in figure 62 show the results of 
end effect. 

43. Resonance, Resistance, and Impedance 

a. Resonance. The antenna is a circuit element 
having distributed constants of inductance, capac- 
itance, and resistance, which can be made to form 
a resonant circuit. The half-wave antenna is the 
shortest resonant length of antenna. However, 
antennas which are 2 or more half-wavelengths 
also can be resonant. Such antennas are said to 
operate on harmonics. If an antenna is 4 half- 
wavelengths at the transmitter frequency, it is 
being operated at the fourth harmonic of its 
lowest resonant frequency. In other words, this 
antenna is a half-wavelength at one-quarter of the 
frequency of operation. An antenna operating on 
the third harmonic is shown in figure 53. 

b. Resistance. 

(1) A current flowing in the antenna must 
contend with three kinds of resistance. 
With the antenna considered as a radi- 
ator of energy, the power expended in the 
form of radiation can be thought of as an 
PR T loss. R r is called the radiation 
resistance. With the antenna considered 
as a conductor, a certain amount of 
energy is dissipated in the form of heat. 
In this PR C loss, R is the ohmic resist- 
ance. There is also an PR loss because of 
the leakage resistance of dielectric ele- 



66 



ments, such as insulators. This R usu- 
ally is included in the ohmic resistance. 

(2) The purpose of the antenna is to dissipate 
as much energy as possible in the form of 
radiation. The energy dissipated by the 
radiation resistance, therefore, is the use- 
ful part of the total power dissipated. 
Since the actual power loss depends on 
the ohmic resistance, this resistance 
should be kept as low as possible. In 
the half-wave antenna, the radiation re- 
sistance is large compared to the ohmic 
resistance, and most of the available 
energy is radiated . The half -wave anten- 
na is, therefore, a very efficient radiator 
for most purposes. 

(3) For a half-wave antenna fed at the center 
point, the radiation resistance is equal 
to 73 ohms. The reference point is the 
center of the antenna at the time of peak 
current flow. Ohmic resistance is refer- 
red to this point. The total resistance is 
of importance in matching the antenna 
to a transmission line. 

c. Impedance. 

(1) Because the half -wave antenna has differ- 
ent conditions of voltage and current at 
different points and because impedance is 
equal to the voltage across a circuit 
divided by the current through it, the 
impendance will vary along the length of 
the antenna. If E is divided by / at 
each point of the voltage and current 
curves in figure 62, the result is the 
impedance curve, Z. The impedance is 
about 73 ohms at the center point and 
rises to a value of about 2,500 ohms at 
the ends. 

(2) The impedance of the half-wave antenna 
usually is considered to be the impedance 
as seen by the transmitter at the input 
terminals. This impedance consists of 
both resistance and reactance. If the 





Figure 62. Impedance along half-wave antenna. 

antenna is cut to a length of exact 
resonance, the reactance is zero and the 
impedance is purely resistive. However, 
if the antenna is longer or shorter than 
resonance, reactance is present. When 
the antenna .is made shorter, capacitive 
reactance is present; when the antenna 
is made longer, inductive reactance is 
present. 

(3) The impedance at the antenna input 
terminals is important in terms of power 
efficiency. If the transmitter is feeding a 
nonresonant antenna, a power loss is 
caused by the reactive component of the 
antenna impedance. Conversely, if the 
frequency of the transmitter is changed, 
the electrical length of the antenna also 
changes. If the frequency is made some- 
what higher, the electrical length is made 
greater, and inductive reactance is added 
to the impedance. If the frequency is 
lowered, the electrical length is shortened, 
and capacitive reactance is added to the 
impedance. 



Section II. TRANSMISSION LINES 



44. Introduction 

a. A transmission line is a device for guiding 
electrical energy from one point to another. 
Therefore, it can be used to transfer the output 
of a transmitter to an antenna. Although it is 



possible to connect the antenna directly to the 
transmitter, the antenna generally is located some 
distance away. In a vehicular installation, for 
example, the antenna is mounted outside and the 
transmitter inside the vehicle. A transmission 
line, therefore, is necessary as a connecting link. 



b. The transmission line has a single purpose in 
respect to both the transmitter and the antenna. 
This purpose is to transfer the power output of 
the transmitter to the antenna with the least 
possible loss. How well this purpose is accom- 
plished depends on the characteristics of the 
transmission line used. 

45. Transmission-Line Characteristics 

a. Terminology. 

(1) The transmission line used to couple 
energy from the transmitter to the an- 
tenna has an input end and an output 
end. The output circuit of the trans- 
mitter is coupled to the input end, also 
called the generator end or source. The 
antenna is coupled to the output end, 
also called the load end or the sink. 



(2) The ratio of voltage to current at the 
input end is known as the input imped- 
ance. The ratio of voltage to current at 
the output end is known as the output 
impedance. If the line were of infinite 
length, the characteristic impedance would 
be the ratio of voltage to current on this 
infinite line. This value is a constant 
for a given transmission line. 

(3) By comparing its electrical length to the 
wavelength of the energy to be trans- 
ferred, a transmission line can be called 
long or short. It is short when its length 
is short compared with a wavelength, 
and long when its length is long com- 
pared with a wavelength. This becomes 
important when considering the efficiency 
of energy transfer through the line, be- 
cause the line has distributed constants 
the effect of which increases with length. 



TRANSMITTER 



INPUT 
END 



TRANSMISSION 
LINE 



OUTPUT 
ENO 



ANTENNA 



A 



r" 



TRANSMITTER 



ANTENNA 



L I 

2 3 



L 



B 

TM 666-67 



Figure 63. Basic transmission line and equivalent circuit. 



68 



6. Distributed Constants. 

(1) The transmission line is essentially a four- 
terminal device. Two terminals (input 
end) are connected to the transmitter, 
and two terminals (output end) are con- 
nected to the antenna. Between these 
terminals are distributed constants of 
inductance, capacitance, and resistance; 
their values depend on the physical char- 
acteristics of the line. 

(2) A of figure 63, shows a basic system con- 
sisting of a transmitter, two wires, and 
an antenna. The equivalent circuit, B, 
shows that any given section of the line 
has a certain amount of distributed con- 
stants which are divided into three equal 
sections of lumped constants. The num- 
ber of sections depends on the unit of 
length chosen. The amounts of induct- 
ance, capacitance, and resistance depehd 
on the length of the line, the size of the 
conducting wires, the spacing between 
the wires, and the dielectric (air or insu- 
lating medium) between the wires. 

(3) These constants actually cannot be dis- 
tinguished in the manner shown in B of 
figure 63. For example, the resistance is 
distributed uniformly along the entire 
length of the line and usually is measured 
in ohms per unit length. This is repre- 
sented as R s in section 1. There is also 
a certain amount of leakage resistance 
between the wires, represented as R S b- 
This resistance is in shunt with the input 
and output ends and is the result of cur- 
rent leakage between the wires through 
the dielectric. 

(4) The wires forming the transmission line 
also possess distributed inductance. This 
inductance can be seen in the action of 
magnetic fields set up by current flow. 
For example, if current flow attempts to 
drop to zero suddenly, the collapsing 
magnetic fields sustain the current for a 
time . The time during which the current 
is sustained is a measure of the distributed 
inductance. The distributed inductance, 
L, is considered to be in series With 
the line, and is measured in microhenries 
per unit length. 

(5) The two wires substituted for the plates 
of a capacitor in figure 49 show an electric 
field produced between them when they 



were connected to a source of emf. This 
is also true of the wires that constitute 
the transmission line. The intensity of 
this electric field is a measure of the dis- 
tributed capacitancy, which is expressed 
in micromicrofarads per unit length. 
These wires act as a capacitance, C, 
shunted across the line (B of fig. 63). 

c. Characteristic Impedance. In addition to 
having the distributed constants, a transmission 
line has a characteristic impedance. If an infinitely 
long transmission line is assumed, then the charac- 
teristic impedance, Z , determines the current that 
flows when a given voltage is applied. This 
impedance is purely resistive, and it is constant 
for a given transmission line. The characteristic 
impedance is important in determining how well 
energy is transferred from the source to the load. 
For the infinitely long line, all of the energy is sent 
out on the line, and none returns to the source. If 
a finite line is terminated with a purely resistive 
load equal to Z , the source appears to see an infi- 
nitely long line, and all the energy passes into the 
line. If the line is terminated in &ny other load, 
energy is reflected back to the source. 

d. Attenuation and Losses. The ideal trans- 
mission line has no losses. It transfers all the 
energy available at the transmitter to the antenna. 
Actual transmission lines, however, dissipate power 
in three ways. 

(1) Radiation. The transmission line tends 
to act like an antenna. Radiation losses 
can be considerable with certain types of 
lines. 

(2) Heating. The resistance of the conduc- 
tors dissipates a certain amount of power 
in the form of heat {PR loss). At higher 
frequencies, an appreciable amount of 
heat loss can result from skih effect. An 
PR loss also results from leakage between 
the conductors (dielectric loss). Heat 
loss increases with lines having a lower 
characteristic impedance because of the 
higher currents that are permitted to flow. 

(3) Reflection. It has been explained that a 
load other than Z reflects energy back 
along the line. This results in the reflec- 
tion loss explained below. 

e. Reflection of Energy. 

(1) In discussing the infinitely long antenna 
wire, it was pointed out that energy 
injected by the transmitter results in 
traveling waves. The same can be said 



69 



of an infinitely long transmission line. 
Traveling waves of voltage and current 
continue to move down the line so long 
the line has no end. 

(2) Assume a finite line where the two con- 
ductors terminate abruptly, as if they 
were cut. Traveling waves reaching this 
open end are reflected in the same manner 
and this results in the formation of stand- 
ing waves of voltage and current that are 
out of phase, just as on the antenna. 
The reflected waves represent energy that 
is not absorbed by the load but is reflected 
back along the line. This is undesirable 
in a transmission line, where the object is 
to transfer as much energy as possible to 
the load. 

(3) If energy is reflected, standing waves are 
formed, which means a changing ratio of 
voltage to current along the line, and 
therefore a changing line impedance. If 
all energy is reflected from the output 
end and none is absorbed by the load, the 
impedance is purely reactive all along the 
line. If some energy is absorbed and 
some reflected, the line impedance either 
can be resistive (more or less than Z„) or 
can have both resistive and reactive 
components. 

/. Z e and Reflection. 

(1) There can be no reflected waves, and 
hence no standing waves, on an infinitely 
long line. However, an infinitely long line 
has an impedance of Z . When the trans- 
mitter injects energy into a line impedance 
equal to the characteristic impedance, 
there are no standing waves and no 
reflections. The transmitter can appear 
to see an infinitely long line if a resistive 
load equal to the characteristic im- 
pedance is placed across the output end. 
Consequently, a line terminated in this 
manner causes no reflection of energy and 
no standing waves. This results in a 
maximum transfer of energy from trans- 
mitter to antenna. 

(2) Inductance, capacitance, and resistance 
found in a transmission line are distrib- 
uted uniformly along its length. There- 
fore, no reflection of energy takes place 
unless the impedance at some point on 
the line is different from that caused by 
the distributed constants. The imped- 



ance seen by the transmitter can be 
changed by changing the load. The 
traveling waves reaching the load sud- 
denly encounter an impedance different 
from that along the line, resulting in the 
formation of standing waves and reflec- 
tion of energy. Reflections occur so long 
as the load differs in any way from Z„. 
g. Basic Line Terminations. The load which 
terminates the transmission line can vary from 
zero to infinity. It can be either resistive or reac- 
tive, or both. A line terminated in a resistance 
equal to Z„ is said to be properly terminated. In 
this case, the line appears to be infinitely long 
looking from the source toward the load, and no 
reflection occurs. Any other load causes reflection 
and the formation of standing waves. The two 
extreme cases of line terminations are shorted or 
open output terminals. In the first, the short 
represents a load equal to zero. Since a zero load 
differs from the characteristic impedance, standing 
waves are set up on the transmission line. This 
short-circuited line is known as a closed-end line. 
If the output terminals are left open, the load is 
infinite, also resulting in standing waves. This is 
called an open-end line. In either the closed-end 
or the open-end line discussed in (1) and (2) below, 
no power is delivered to the load (antenna). The 
use of either line results in complete reflection of 
energy back to the source (barring line losses). 
(1) The standing waves created on a closed- 
end line are shown in A of figure 64. 
The line used is a half-wavelength at the 
operating frequency. For simplicity, the 
waves are shown using the top wire as 
a zero baseline. Because of the short 
across the output terminals, the current 
loop is maximum at that point, and the 
voltage is zero. A quarter-wavelength 
back along the line, there is a current 
node with a maximum on the voltage 
loop. A half-wavelength back (at the 
input terminals), there is a voltage node 
with a current loop (maximum negative) . 
The voltage and current ratio, and there- 
fore, the impedance, continually change 
along the line. At the load, the im- 
pedance is zero. At the current node, 
the voltage is maximum and the im- 
pedance infinite (there is no current 
flow). At the source, the current again 
is maximum and the impedance zero. 
At the load end, therefore, no power is 



70 



absorbed (P=PZ=P x 0=0), all of it 
being reflected back along the line. At 
the source, the transmitter is working 
into zero impedance, and there is no ex- 
penditure of power. (In the actual case, 
however, some loss is caused by numerous 
factors, such as radiation or heat.) Con- 
sequently, no energy is transferred to 
the antenna if there is a short across the 
output terminals of this line. 
(2) In a half-wavelength section of open-end 
line, the load, Z L , is infinite, and creates 



EI IE 




A B 

TM 6*6-68 

Figure 64- Standing-wave formation on closed- and open-end 
half-wave transmission lines. 

u. X »j 

2 



the standing waves shown in B of figure 
64. Although the standing wave of 
voltage is maximum at the open end, 
current is zero. At a quarter-wavelength 
back, the current is maximum, the voltage 
is zero, and the impedance is zero. At 
the source, the current again is zero, the 
voltage maximum, and the impedance 
infinite. Since the transmitter is work- 
ing into an infinite impedance, no current 
flows and no power is expended (P=PZ= 
x Z=0). 

(3) If a transmission line is terminated in 
either a low or a high resistance (fig. 65), 
as compared with Z , the effect is similar 
to an open or a closed end. The low 
resistance termination causes standing 
waves, which have nodes and loops at 
the same points as with a closed end. 
The amplitudes of the standing waves are 
lower, however, and the nodes no longer 
reach zero. This is because some power 
is absorbed by the low resistance, al- 
though most of it is reflected back to the 

. x . 

2 




A B 

TM 666-69 

Figure 65. Half-wave lines terminated in low and high resistance. 



71 



source. Compare A of figure 65, with 
A of figure 64. 

(4) In B of figure 65, the line is terminated 
in a high resistance. This produces 
standing waves similar to those produced 
by an open end, as in- B of figure 64. 
Again, the difference is only in standing- 
wave amplitude. Figure 66 shows that 
the standing waves decrease in amplitude 
when the characteristic impedance is 
approached from either side. Values of 
resistive load ranging from a closed to an 
open end are noted. 

(5) The effects of purely reactive loads are 
shown in figure 67. In A, a half-wave 
section of line is terminated in a purely 
capacitive load. Compare the standing 
waves in this case to the standing waves 
resulting from an Open-end line. The 
standing waves are shifted an eighth- 
wavelength forward from the source and 
the current leads the voltage, the latter 
condition resulting in an input impedance 
consisting of capacitive reactance. As 
the capacitance is increased, the voltage 
node moves nearer and nearer the output 
end. The result of having a transmission 
line terminated in capacitance is to in- 
crease the effective electrical length. 
This is the same type of change that 
results from end effect on an antenna. 
The effect is equivalent to adding an 
open -end section of line which is less than 
a quarter-wavelength long. 

(6) Terminating the line in an inductance 
results in the standing waves shown in 
B of figure 67 . Compared to the open-end 
termination, the current wave is shifted 
an eighth-wavelength back toward the 
source. This results in an input imped- 
ance consisting of inductive reactance 
(voltage leads current). Placing an in- 
ductance across the output is equivalent 
to adding a closed-end section of line less 
than a quarter-wavelength long. De- 
creasing the inductance moves the volt- 
age node nearer the output end. 

h. Other Line Terminations . 

(1) It is possible to connect either the genera- 
tor or the load at other points along the 
transmission line besides the ends. So 
long as the line is not terminated in a 
resistive load equal to Z , standing waves 

72 



I E 




h % H 

TM 666-70 

Figure 66. Varying resistive load from zero to infinity on 
the half-wave line. 



EI IE 




A B 

TM 666-71 

Figure 67. Half-wave lines terminated in capacitance and 
inductance. 

exist. This results in a line impedance 
which varies all along the line. The 
generator then can be connected to work 
into different impedances at different 
points. 

(2) Assume an open-end half-wavelength line 
(fig. 68). The impedance curve along 
the line not only changes its value in 
ohms, but also changes in type of imped- 
ance. If the generator is connected a 



x 

2 




A 

Figure 68. Line impedance changes on a half-wave 



distance between a quarter- and a half- 
wavelength from the load, the input 
impedance consists of inductive react- 
ance. Connecting the generator to the 
same point on the shorted line in B of 
figure 68 results in an input impedance 
consisting of capacitive reactance. 

(3) If the generator in A of figure 68 is con- 
nected exactly a quarter-wavelength from 
the open end, the input impedance is a 
very low resistance. The equivalent 
lumped-constant circuit, however, is not 
simply a small resistor, but a series- 
resonant circuit. In other words, the 
transmission line has resonant lengths 
like an antenna. If the generator in B 
is connected a quarter- wavelength from 
the load, the input impedance consists 
of a very high resistance. The equiva- 
lent circuit in this case is a parallel- 
resonant circuit. 

(4) It is possible to connect the load at differ- 
ent points along the line. In figure 69, 
the load is connected a sixteenth-wave- 
length from the end of a shorted quarter- 
wavelength line. The generator is con- 
nected at a point of high voltage and low 



2 




B 

TM 666-72 

,-end line and on a half-wave closed-end line. 

73 





TM 666-73 

Figure 89. Open and closed lines as step-up and step-down transformers. 



current. The load is connected to a 
point of relatively lower voltage and 
higher current. The equivalent circuit 
for the line, therefore, is a step-down 
transformer. The equivalent of a step-up 
transformer can be obtained by connect- 
ing the load to the same point on an 
open-end quarter-wavelength line, as in 
B of figure 69. 
i. Changing Line Length. 

(1) For open- and closed-end lines, the 
impedance seen by the source is given in 
figure 70. In A, different lengths of 
open-end line are given. These vary 
from less than a quarter-wavelength to 
1 wavelength. The impedance seen by 
the source is repeated in half-wave steps. 
For example, the series-resonant circuit 
seen at 1 quarter-wavelength is seen 
again at 3 quarter-wavelengths. It also 
is to be seen at the fifth and seventh 
quarter-wavelengths, and for every 
length measured in odd quarters. On the 
other hand, the generator sees a parallel- 
resonant circuit at a half-wavelength, 1 
wavelength, and 3 half-wavelengths and 
so on. In other words, the input imped- 



ance consists of a parallel-resonant circuit 
on open-end lines an even number of 
quarter-waves long. 
(2) In J? of figure 70 different lengths of 
shorted line are given. For shorted 
lines, series resonance occurs at even 
quarters and parallel resonance at odd 
quarters. This is the reverse of the 
situation with open-end lines. For 
lengths of line shorter and longer than 
even or odd quarters, the input imped- 
ance is indicated as either inductive or 
capacitive. The generator also can be 
inserted at different points along any of 
these lines. The input impedance then 
is simply the line impedance at that 
point. 

j. Standing-Wave Ratio. 

(1) The actual loads connected to the trans- 
mission line usually have both resistive 
and reactive components. The result is 
the formation of standing waves on the 
line. Considering the standing wave of 
voltage, the ratio of maximum to mini- 
mum voltage along the line is called the 
standing-wave ratio. This ratio also can 
be obtained by measuring maximum and 



74 



GENERATOR SEES CAPAClTlVE REACTANCE 
BETWEEN ZERO AND 

4 



GENERATOR SEES SERIES-RESONANT 

CIRCUIT AT A 



GENERATOR SEES INDUCTIVE REACTANCE 
BETWEEN £ AND £ 



GENERATOR SEES PARALLEL - 
RESONANT CIRCUIT AT £ 



GENERATOR SEES CAPAClTlVE 
REACTANCE BETWEEN £ 
ANO ^ 



GENERATOR SEES SERIES - 
RESONANT CIRCUIT AT 



GENERATOR SEES 
INDUCTIVE REACTANCE 
BETWEEN 
ANO X 



GENERATOR SEES\*L 
PARALLEL* 
RESONANT 

CIRCUIT AT X 




GENERATOR SEES INDUCTIVE REACTANCE 

BETWEEN ZERO AND h 
4 



GENERATOR SEES PARALLEL- RESONANT 
CIRCUIT AT ^ 



GENERATOR SEES CAPAClTlVE REACTANCE 
BETWEEN £ AND £ 



GENERATOR SEES SERIES - 
RESONANT CIRCUIT AT A 




GENERATOR SEES INDUCTIVE 
REACTANCE BETWEEN 
ANO 

4 



GENERATOR SEES 

PARALLEL-RESONANT 

CIRCUIT AT li. 

4 



GENERATOR SEES 
CAPAClTlVE 
REACTANCE 
BETWEEN 
ANO X 



GENERATOR SEES *C/ \><L 
SERIES-RESONANT 
CIRCUIT AT X 



TM ••• 



B 



Figure 70. Changing line length changes the impedance seen by source. 



75 



minimum current along the line. The 
standing-wave ratio provides a measure 
of the energy reflected. 

(2) When the line is terminated in a resistance 
equal to Z , the maximum and minimum 
values of current are the same. The 
SWR (standing wave ratio) is 1 to 1 , or 1 . 
In this condition the load is said to be 
matched to the line. All of the energy is 
absorbed by the load (neglecting line 
losses), and there are no standing waves. 
Such a line is called flat, since the 
impedance is the same value, Z B , all 
along the line. 

(3) If standing waves occur on the line with 
a given load, the SWR is a measure of 
the degree of mismatch between load and 
line. For example, assume that a 
resistive load of 500 ohms is used to 
terminate a line with Z„ of 50 ohms. 
If the SWR is measured, it is found to 
be 10. This is the same as dividing 500 
by 50. 

k. Resonance. 

(1) If a load resistance equal to Z e terminates 
the line, then the line is matched, or flat. 
There are no reflections, and the SWR is 
unity. By definition, a matched line 
is nonresonant. This means that there is 
no reflection of energy and there are no 
standing waves, resulting in maximum 
transmission of energy. 

(2) Assume a section of line which is termi- 
nated in a load not the same as Z . 
This line has standing waves which are 
greater than 1, indicating the degree of 
mismatch. This line is called resonant. 
Depending on the type of load and the 
length of line used, the transmission line 
can be represented by a series- or a 
parallel-resonant circuit. 

(3) The nonresonant, or matched, line is used 
principally in fixed or semifixed opera- 
tion, since its design and construction 
take into consideration a great many fac- 
tors. Even in fixed installations, how- 
ever, only a need for very high efficiency 
warrants the use of a nonresonant line. 
The resonant line, on the other hand, is 
simple in construction and flexible in 
operation, and therefore, is used in most 
field or mobile installations. Although a 
maximum SWR exists on a resonant line, 



it still can be used with relatively high 
efficiency. 

I. Impedance Matching. Assume that a trans- 
mission line has a characteristic impedance of 100 
ohms. This line must be used to feed an antenna 
with 75, ohms of resistance and 30 ohms of indue- 
tive reactance at th« operating frequency. Since 
a mismatch occurs if the line is connected directly 
to the antenna, an intermediate element must be 
used between the line and the antenna. Such an 
element is called an impedance-matching device. 
For this purpose, a carefully constructed section 
of transmission line can be used. This is possible 
because a line exhibits an impedance which varies 
with length. If the proper length is selected, this 
length then can serve as the matching element. 

m. Velocity of Propagation. Wave travel on 
the transmission line is retarded in the same man- 
ner as on the antenna. Consequently, the 
electrical length and the physical length are not 
the same. The electrical length depends directly 
on the dielectric medium, the physical dimensions 
of the conductors, and the space between them. 
All the factors on which the electrical length de- 
pends can be reduced to a single constant factor 
for a specific type of line. The electrical length 
then can be expressed as — 

Z=246 k/f 

where I is the electrical length in feet, k is the 
factor for the specific line, and / is the frequency 
of operation in megacycles. Typical values of k 
range from 0.56 to 0.975. 

46. Types of Transmission Line 

Until now, only one type of transmission line 
has been discussed. This is the type using two 
parallel conductors, uniform in every dimension. 
So far as general theory is concerned, the previous 
discussion applies to all types of transmission 
lines. Physically, however, transmission lines 
differ considerably in their construction and 
specific characteristics. The various types are 
discussed below. 

a. Single-Wire Line. This is the simplest type 
of transmission line. A single-wire conductor 
links the transmitter to the antenna. The return 
path completing the circuit is ground. Because 
there is only one metallic conductor, the line is 
unbalanced. This condition leads to large radia- 
tion losses, which is a definite disadvantage. 
Another disadvantage is the lack of a constant 



76 



INSULATING SPACERS 




tn 

2 

x 
o 

o 

N 



800 
700 
600 
500 
400 
300 
200 
100 



1 1 1 1 1 1 






















- a — 
1 


r • 













































































































































































































1 



2 5 10 50 100 
RATIO b/a (TWO-WIRE LINE) 



500 



B 

TM 666-75 

Figure 71. Variation in Z t with changes in b/a ratio, open two-wire line. 



physical relationship between wire and ground, 
which leads to a varying characteristic impedance, 
making the line difficult to match to the antenna. 
Because of these two disadvantages, the single- 
wire line is used rarely. It is found where its 
advantage of easy installation outweighs its 
disadvantages. 

b. Open Two-Wire Line. 

(1) Because it uses two parallel conductors, 
this is called also the parallel-conductor 
line, and because the dielectric medium 
is air, it is known also as an open-wire 
line (fig. 71). The construction and 
installation of the open two-wire line 
are nearly as simple as for the single- 
wire line. Although the balanced con- 
ductors act to reduce radiation loss, the 
balance is critical, and nearby metallic 
objects tend to unbalance the line and 
cause large radiation losses. The two 
wires used in this line are kept at a con- 
stant distance from each other by means 
of insulating bars called spacers, or 



spreaders, shown in A of figure 71. The 
actual distance used between the con- 
ductors depends on the impedance re- 
quired, the diameter of the wire, and the 
frequency of operation. 
(2) The characteristic impedance of a two- 
wire line is relatively constant. For a 
two-wire line having air as a dielectric 
medium, Z„ is given by the formula 



Z„=276 log 10 - 

Of 



where b equals the space between the 
conductors measured from their centers 
and a equals the radius of the wire used. 
This formula is sufficiently accurate 
provided that the ratio b/a is 4 or greater. 
The graph in B of figure 71, demonstrates 
the variation in Z produced by changing 
the ratio, b/a. 
(3) Currents flow through the two parallel 
conductors in opposite directions. If 



77 



the two currents are exactly 180° out of 
phase, the fields nearly cancel and the 
radiation loss approaches zero. At 
relatively low frequencies, this condition 
can be approached. As the frequency 
of operation is raised, however, the two 
currents tend to be more and more out 
of phase, causing considerable radiation 
loss. The loss can be reduced by moving 
the conductors closer together. Because 
of radiation, the distance between the 
conductors should never exceed 0.01 X, 
where X is 1 wavelength at the operating 
frequency. 

(4) Moving the conductors closer together 
lowers the characteristic impedance of 
the line. This can be seen from the 
equation given above. In order to have 
a relatively high impedance and close 
spacing, it is necessary to reduce the 
size of the conductors. Reduction in 
size, however, decreases the power capa- 
bilities of the conductors. The higher 
the frequency of operation, the more 
difficult these problems become. A 
practical limit to the use of two-wire 
lines having air as the dielectric medium 
is reached at approximately 200 mc. 
c. Insulated Two-Wire Line. 

(1) Instead of having air as a dielectric 
medium, the two-wire line can be in- 
cased in a solid dielectric. This type of 
line has several advantages over the 
open line. Installation is simplified 
considerably because of its flexibility. 
For example, it is difficult to run the 
open-wire line around a corner without 
changing the spacing between the con- 




Figure 72. Insulated two-wire line. 

78 




BRAID SHIELD 

TM ««6-77 

Figure 7S. Shielded pair. 

ductors. In the insulated type, the 
dielectric is solid enough to keep the 
conductors evenly spaced, but flexible 
enough to bend easily around corners. 
(2) In one type of insulated line, the two 
conductors are molded into the edges of 
a plastic ribbon called polyethylene 
(fig. 72). The dielectric losses are 




i 2 4 6 8 10 

RATIO b/0 (CONCENTRIC OR COAXIAL LINE) 



B 

TM 666-78 

Figure 74. Variations in Z. with changes in the b/a ratio, 
open coaxial line. 



higher than in a comparable open-wire 
line, and the higher dielectric constant 
lowers the characteristic impedance. 

d. Shielded Pair. A further development of 
the insulated two-wire line is the shielded pair 
(fig. 73). The two parallel conductors are im- 
bedded in a solid dielectric, such as the plastic 
copaline. The insulated pair then is inclosed in a 
tube made of braided copper. The entire assembly 
is given a weatherproof coating. The principal 
advantage of the shielded pair over other types 
of two-wire lines is its low radiation loss. This 
is true because the shield provides a uniform 
ground for both conductors, resulting in a well- 
balanced line. Furthermore, the shield provides 
protection from stray pickup in the presence of 
external fields. 

e. Twisted Pair. If two insulated wires are 
twisted together, a flexible transmission line 
results without the use of spacers. This type is 
limited to use as a short untuned line because of 
its high losses. It should not be used at frequen- 
cies above 15 mc. 

/. Coaxial Lines. 

(1) It is possible to place one conductor 
inside the other to form a transmission 
line. Such a line is called coaxial, or 
concentric. The open line (air dielec- 
tric) is shown in A of figure 74. Usually, 
it consists of a wire conductor placed 
inside a flexible metal tube which serves 
as the second conductor. The inner 
wire is fixed along the central axis of the 
outer tube by spacers, usually plastic 
beads. The open coaxial line is used to 
provide efficient operation at relatively 
high frequencies. There is little radi- 
ation loss from this type of line because 

Section III. BASIC 

47. Introduction 

The transmission line used to couple the trans- 
mitter to the antenna sometimes is called a 
feeder. Because two general types of trans- 
mission line are used (resonant and nonresonant), 
methods of feed can be divided into two classes — 
tuned (resonant) and untuned (nonresonant or 
matched). Both types of line involve a trans- 
mitter, means for coupling the transmitter to the 
line, the transmission line, means for coupling the 
line to the antenna (including an impedance 



the outer conductor confines radiation 
to the space inside the line. External 
objects consequently have no effect on 
transmission, making this line definitely 
superior to the two-wire type. Instead 
of air, the line can be filled with a 
flexible, plastic dielectric, forming a solid, 
coaxial line which has the advantage of 
greater flexibility compared with an open 
coaxial line. The use of a solid di- 
electric, however, increases the dielectric 
losses. 

(2) The characteristic impedance of the open 
coaxial line can be calculated from the 
formula, 

Z(,= 138 log 10 bfa 

where b equals the inner diameter of the 
outer conductor and a equals the outer 
diameter of the inner conductor. Vari- 
ations in Z„ with changes in the ratio 
bja are given in the graph shown in B, 
figure 74, which includes open coaxial 
lines. The formula for Z of a solid 
coaxial line is given by 

,,138, b 
Z„= log, - 

€ a 

where e is the dielectric constant of the 
dielectric material used . The other quan- 
tities (b and a) are the same as for the 
open coaxial line. If « is equal to 1, 
then the two formulas become identical. 
From both formulas, it can be seen that 
a high ratio of b/a means a high Z„, and 
conversely, a low ratio of b/a means a 
low Z t . 

FEEDER SYSTEMS 

matching device if necessary), and the antenna 
itself. 

48. Transmitter-to-Line Coupling 

a. Transmitter Coupling to Untuned Lines. 
(1) On the untuned line, the SWR is 1.5 or 
less, and the line is considered properly 
terminated. For a single- wire feeder, 
two types of transmitter coupling are 
shown in A and B of figure 75. In A, 
the matched line is coupled capacitively 



79 



to a tap on the coil in the output circuit 
of the final amplifier. The tap is ad- 
justed for maximum power transfer. 
Although this method has the advantage 
of simplicity, it allows a great deal of 
harmonic radiation from the antenna. 
In B, a link is used to couple the trans- 
mitter output circuit to the feeder input 
circuit. The coils forming the links can 
be connected by a low-impedance section 
of line. This allows the transmission 
line to be moved a distance from the 
transmitter. This method is relatively 
simple and efficient, permitting little 
harmonic radiation. 
(2) The coupling used with two-wire systems 
depends in part on whether the trans- 



LINE 




mitter output circuit is single-ended or 
double-ended. It also depends on the 
input impedance of the two-wire feeder. 
If the impedance is low and the output 
circuit is single-ended (unbalanced), then 
the circuit in C can be used. A more 
flexible coupling arrangement is shown in 
D. This link circuit permits coupling to 
a high- or low-impedance two-wire feeder 
from the single-ended output circuit. 
(3) When the output circuit of the transmitter 
is double-ended (balanced), the two-wire 
feeder can be coupled capacitively to the 
output tank, as in E. This method 
usually is found with open two-wire 
lines. A link arrangement, as shown in 
F, also is available for balanced output 

LINE 




B 




LINE 




D 





TM 666-79 



80 



Figure 75. Various methods of coupling the transmitter to untuned transmission lines. 




B 



TM 666-80 



Figure 76. Two methods of coupling the transmitter to tuned transmission lines. 



circuits. The line taps in both cases are 
placed symmetrically in respect to the 
center of the coil. 

b. Transmitter Coupling to Tuned Lines. 

(1) If a resonant transmission line is used, 
the type of coupling depends on whether 
the line input impedance is the equivalent 
of a series- or a parallel-resonant circuit. 
The circuit in A of figure 76, is an arrange- 
ment used with a line presenting a low 
impedance to the transmitter. The extra 
link serves to reduce harmonic radiation 
from the antenna. 

(2) An arrangement that can be used for 
either series- or parallel-tuned lines is 
shown in B. In this circuit, the link 
reduces harmonic radiation to a mini- 
mum. Except for the method of employ- 
ing the link, the circuits in A and B are 
almost identical. 

c. Pi-Section Coupling Network. The pi-section 
circuit is a versatile network which can be used 
with both tuned and untuned lines. It also can 
be used with balanced and unbalanced final- 
amplifier circuits over a wide range of line imped- 
ances. A single pi-section is used for coupling to 
a single-wire feeder (A of fig. 77). For a two- 
wire feeder, a double pi-section provides the 
necessary coupling, as in B. 



49. Methods of Feed 

a. Line Conditions. 

(1 ) Between the transmitter coupling network 
and the antenna input terminals, certain 
factors must be taken into consideration 
with regard to the transmission line. 
These factors apply equally to both tuned 
and untuned lines, and they deal with the 
problem of minimizing radiation loss 
from the line. 

(2) A minimum of radiation loss is attained 
when the fields surrounding the two con- 
ductors forming the line are equal and 
opposite in phase, and hence cancel each 
other. In this condition, a transmission 
line is called balanced. If the line is unbal- 
anced, excessive radiation loss occurs. 
The radiation loss is directly proportional 
to the amount of line unbalance. Com- 
mon causes of line unbalance are — 

(a) Unequal size, length, or spacing of the 
conductors. 

(6) Unequal loading of the two conductors. 
This can result either from the antenna, 
or from coupling between one conduc- 
tor and a nearby conducting surface. 

(c) Acute angle in the line. 

(3) Various methods are available for bal- 



6 



-I 

7"- 



TO 



-I 

7^ 



LINE 
-i 

7- 




B 



Figure 77. Pi-section coupling to tuned or untuned lines. 



TM 666-81 

81 



v v V 



r 




B 



TM 666-82 

Figure 78. Devices for balancing transmission lines. 

ancing different types of transmission 
lines. Open-wire and shielded-pair lines 
have constant dimensions. If the line is 
a wavelength long, it usually is balanced. 
If an open-wire line longer than 1 wave- 
length is used, transposition blocks can 
be substituted for spacers. The positions 
of the conductors are reversed at regular 
intervals by means of these blocks (A of 
fig. 78). This serves to balance the line 
to ground and nearby objects. 
(4) The properties of a quarter-wave section 
of line can be utilized to balance unequal 
line currents in the transmission line 
proper. The balancing section used with 
a coaxial line, as in B, is called a bazooka. 
It serves to make the impedance of the 
outer conductor equal to that of the 
inner conductor. If the unbalance is 10 
percent or less, the line can be considered 
balanced for all practical purposes. 
b. Feeding the Antenna. Two important factors 
must be considered in answering the question of 
how to connect the line to the antenna. An 
antenna at resonance has standing waves on it, 
presenting a varying impedance along its length. 
For maximum transfer of energy, the impedance 
at the output end of the line must match the 
antenna input impedance. One factor to consider, 
therefore, is the antenna impedance. Another 
factor is the type of antenna used. Methods of 
feed are divided into two classes, tuned and un- 
tuned, corresponding to the two classes of trans- 
mission line. If it is no longer than 1 wavelength, 
a resonant line works with relative efficiency. If 
it is longer, a matched line should be used. 

50. Tuned Methods of Feed 

a. Although the resonant line can have the same 



physical construction as the nonresonant line, 
standing waves are present on the resonant type. 
Since standing waves are present on the antenna, 
so far as the transmitter is concerned the antenna 
is merely an extension of the transmission line. 

6. For all practical purposes, there are two 
points at which a resonant line can be used to feed 
a half-wave antenna. The point of highest im- 
pedance occurs at either end; the point of lowest 
impedance occurs at the center. If the antenna 
is a half -wavelength at the operating frequency, 
voltage loops occur at its ends and a current loop 
occurs at its center. If this antenna is fed at one 
end, it is being fed at a point of high impedance 
(about 2,500 ohms) and, also, at a voltage loop. 
If it is fed at the center, the connection is to a low- 
impedance point (about 73 ohms), which is also a 
current loop in this case. 

<•. An open-end resonant line can be either an 
odd or an even number of quarter- wavelengths. 
If it is an odd number, it acts as an impedance 
transformer. If it is an even number, it has the 
same input and output impedance. Either of 
these types can be connected to the center or the 
end of the half-wave antenna. 

</. The four possible arrangements for feeding 
the half-wave antenna with tuned lines are shown 
in figure 79. In A. the antenna is center-fed. The 
impedance curve for the antenna shows that the 
antenna impedance, Z A . is low at this point. The 
output impedance, Z T , of the line, therefore, must 
be low. Since the line used is an even number of 
quarter-wave lengths, a low output impedance 
results in a low input impedance. This determines 
the type of transmitter-to-line coupling. A series- 
resonant circuit is used to provide the necessary 
low-impedance match. The arrangement in A also 
is called current-Jed, since the point of feed occurs 
at a current loop. 

e. In B of figure 79 the antenna is end-fed. The 
antenna therefore is being fed at a point of high Z A . 
In order to obtain a high Z T , the even quarter-wave 
length line is coupled to the transmitter by means 
of a high-impedance, parallel-resonant circuit. 
This type of feed results in an unbalanced line con- 
dition, and little radiation from the line occurs 
provided that the antenna is cut to a resonant 
length. An antenna far from resonance causes 
serious line radiation loss. The end-fed half-wave 
antenna is more common, and sometimes is 
referred to as Zepp-fed. 

f. By comparison, center feed is a better arrange- 
ment than end or Zepp feed. In the center-fed 



82 



method, equal lengths of the antenna are added 
to each side of the line. Even if the antenna is 
actually not at resonance, the line remains near a 
balanced condition, and there is little line radia- 
tion loss. The symmetry of the center-fed method 
allows the antenna to be used over a wide fre- 
quency range, since it does not have to be near 
resonance to prevent line radiation loss. 

g. The center-fed antenna using a tuned line an 
odd number of quarter-wavelengths is shown in C 
of figure 79. The distinguishing feature of this 
length of line is its impedance transforming action. 
A low impedance across the output terminals is 
reflected as a high impedance at the input ter- 
minals. The antenna in C is being fed at a point 
of low impedance. The resulting high impedance 
reflected at the input end necessitates the use of a 
high-impedance, parallel-resonant circuit for cou- 
pling to the transmitter. If the same line is used 
to end-feed the antenna, as in D of figure 79, the 
high Z A is reflected as a low input impedance. A 
series-resonant circuit, therefore, is used for cou- 
pling to the transmitter. As in B, end feed results 
in a condition of line unbalance which must be 
kept below 10 percent. 

51 . Untuned Methods of Feed 

a. General. For all practical purposes, a non- 
resonant transmission line is defined as a line with 
SWR of 1.5 or less. To obtain this low standing- 
wave ratio, the lin^ must be terminated in an 
impedance very close to its characteristic imped- 
ance. The antenna impedance may vary con- 
siderably from this value. Consequently, an 



impedance matching device sometimes must be 
inserted between the line and the antenna. The 
input impedance of this device matches the charac- 
teristic impedance of the line, and its output 
impedance matches the antenna impedance. 
6. Single-Wire Feed. 

(1) The single-wire feeder can be used as a 
nonresonant line to feed the resonant 
half-wave antenna. For simplicity of 
construction, an impedance-matching de- 
vice generally is not used. The charac- 
teristic impedance of this line ranges from 
500 to 600 ohms. Since the antenna 
impedance varies from 73 ohms at the 
center to 2,500 ohms at the end, by con- 
necting the line to a suitable point 
between the end and the center, an 
impedance match can be made which 
results in a relatively flat line. This type 
of connection is called off-center feed 
(A of fig. 80). 

(2) The impedance point on the antenna 
which matches approximately the char- 
acteristic impedance of the line occurs 
about one-half of a half-wavelength from 
the center of the antenna. A precaution 
to be observed in bringing the feeder to 
the antenna is to keep the two elements 
at right angles to each other for at least 
one-third wavelength. This serves to 
prevent antenna coupling to the line. 

(3) The off-center feed method just described 
is limited in use, despite its simplicity, 
because it requires the presence of a 
highly conductive ground return, which 




TM 666-64 



Figure 80. Single-wire and twisted-pair feed systems. 



84 



serves as the line return to the tank. 
This coupling occurs through the an- 
tenna-to-ground capacitance, and it re- 
quires a well grounded tank circuit. An- 
other disadvantage is the excessive radia- 
tion loss from a single-wire feeder, since 
there is no parallel conductor to cancel its 
radiation field. 

c. Twisted-Pair Feed. An emergency method 
for center-feeding a half-wave antenna (B of fig. 
80) uses a twisted-pair line, the characteristic 
impedance of which is approximately 70 to 80 
ohms. The line in the figure is matched to the 
antenna. If Z is less than Z A , the ends of the line 
are spread, or fanned, to get the correct match. 
This follows from the principle that increasing the 
space between the conductors increases the an- 
tenna impedance seen between the terminals. 
This method normally is not used because of the 
high line losses associated with the twisted pair. 
In emergencies, however, the twisted pair forming 
a lamp cord or a field telephone wire can be used 
as a transmission line. 

d. Two-Wire Feed Using Delta Match. 

(1) A common system for feeding the half • 
wave antenna uses a balanced open-wire 
transmission line. Because of construc- 
tional difficulties, the open two-wire line 
usually cannot have a characteristic im- 
pedance to match the antenna input 
impedance. A practical line of this type 
has a Z of 400 to 700 ohms. If this line is 
used to center-feed a half-wave antenna 
having a 73-ohm input impedance, some 
type of impedance transformation is nec- 
essary. A method similar to fanning the 

ANTENNA 



DELTA 
MATCHING 
SECTION 



OPEN 
TWO-WIRE 
FEEDER 



/ 



CENTER 
LINE 



twisted pair is used. In this case, it is 
called a delta match. 

(2) An example of the use of a delta match is 
illustrated in figure 81. The delta match 
is obtained by spreading the transmission 
line as it approaches the antenna. As- 
sume that the line has a characteristic 
impedance of 600 ohms. The antenna 
impedance at its center is 73 ohms. The 
delta section therefore must have an in- 
put impedance of 600 ohms to terminate 
the line properly. Proceeding from the 
center to either end, a point is passed 
where the impedance equals the imped- 
ance at the output terminals of the delta 
section. The delta section then is con- 
nected this distance on either side of the 
center of the antenna. 

(3) Electrically speaking, the delta section is 
actually part of the antenna. The delta, 
therefore, introduces a radiation loss, 
which is one of its disadvantages. An- 
other disadvantage is that cut-and-try 
methods must be used to determine di- 
mensions A and D of the delta section 
(fig. 81). Since both A and D can be 
varied, adjustment is difficult. The ad- 
vantage of the delta section is that it 
permits the use of a balanced transmis- 
sion line. This means that there is a 
minimum of line radiation, which allows 
the mounting of other lines and antennas 
nearby with minimum effect. A group 
of antennas using deltas is shown in B 
of figure 81. Such a group is known as 
an antenna park, or an antenna farm. 

SEPARATE 
TRANSMITTERS 




B 

TM S66-8S 



Figure 81. Delta-matched antenna and antenna park. 



85 



e. Two-Wire Feed Using T Match. 

(1) A method of feed having some advantages 
over the delta-section method uses a T 
match {A of fig. 82). The essential dif- 
ference between the delta and the T is 
in the shape of the matching section. 
When the antenna consists of a tubular 
conductor, the 7"-match method is easier 
to construct than the delta section. 
Furthermore, its dimensions are not 
nearly as critical. 



Lfc=*=J 




T MATCHING SECTION 



TWO-WIRE LINE 



ADJUSTABLE SHORTING BARS 



ANTENNA 




T SECTION 



B 



TM 666-86 

Figure 82. T-matched antenna showing T-matching section. 

(2) A sketch of the physical appearance of a 
two-wire line feeding a T-matched an- 
tenna is shown in B of figure 82. In this 
arrangement, insulators can be used to 
support the assembly, and the line can 
be connected to the matching section by 
means of simple soldering lugs. The 
shorting bars permit dimension A to be 
adjusted for the best possible match. 
The radiation loss of the T section is less 
than that of the delta section. It must 
be borne in mind, however, that the T 
match also is part of the radiating ele- 



ment and not part of the nonresonant 
feeder. 

/. Two-Wire Feed Using J Match. Still another 
method for feeding the half-wave resonant antenna 
with a nonresonant line is by means of the J- 
matched section (fig. 83). The assembly consists 
of a shorted quarter-wave section of transmission 
line end-feeding a half-wave antenna. A shorted 
quarter-wave section of line acts as an impedance 
transformer. A high impedance is seen at the 
open end, and this end accordingly is connected 
to the high-impedance end of the antenna. The 
impedance of the lino is relatively low compared 
to the open end of the matching section. Conse- 
quently, the lines are connected near the shorted 
end of the J match, and are made adjustable for 
optimum performance. The J-matched antenna 
is in wide use in mobile installations, where sim- 
plicity and ease of operation are important. 



ADJUSTABLE 
CLAMPS 



TWO-WIRE LINE 




(ANTENNA) 



J MATCHING SECTION 



TM 666-87 



Figure 88. J-matched antenna. 



g. Two-Wire Feed Using Stub Matching. 

(1) A method of feed which uses sections of 
transmission line to connect a nonreso- 
nant line to the antenna is known as stub 
matching. The stub usually is a quarter- 
wave section of open or shorted line, al- 
though longer stubs can be used. In 
figure 84, two types of stub matching 
are shown. One uses the open stub, in 
A, and the other the shorted stub, in B. 
Assume that a 600-ohm nonresonant line 
is used to center-feed a half-wave antenna 



(2) 



86 



with input impedance of 70 ohms. The 
stub to be used therefore must have high 
input impedance and low output imped- 
ance, as shown in A of figure 84. One 
end of an open quarter-wave stub is con- 
nected to the low-impedance antenna ter- 
minals. From transmission-line theory, 
it is known that a quarter-wave line acts 
as an impedance transformer. The im- 
pedance along this stub rises to a maxi- 
mum at the open end. At some point 
along the stub, the impedance matches 
that of the line. In the case given, this 
point occurs a short distance from the 
antenna terminals, and the transmission 
line is connected to the stub at this point. 

(3) A shorted quarter-wave stub can be used, 
as in A, to end-feed a half-wave antenna. 
As in B, the line is connected to the stub 
near the low-impedance end. This type 
leads to a certain amount of unbalance in 
loading the transmission line which 
should not exceed 10 percent. 

(4) If the line cannot be brought close enough 
to the antenna to use a quarter-wave 
stub, longer stubs can be used. For ex- 
ample, any open stub an odd number of 
quarter-waves long can be substituted 
for the quarter-wave stub shown in A. 
In B, any shorted stub an odd number of 
quarter-waves long can be substituted. 
The impedance match to the line is made 
in the last quarter-wave section of the 
stub. Other arrangements also are pos- 
sible, as explained previously. 

(5) The stub-matching system has the dis- 
advantage of limiting the frequency of 



operation, and the antenna must be near 
a resonant length to operate properly. 
This is especially true of the center-fed 
antenna, where the current loop present- 
ing a low impedance can become a node 
presenting a high impedance. This must 
be borne in mind when determining what 
type of impedance-matching device is to 
be used. 

h. Two-Wire Feed Using Q Match. 

(1) In the stub-matching system, the line is 
connected to taps on the stub and the 
correct point at which to connect the line 
to obtain an impedance match is deter- 
mined by trial-and-error methods. The 
(2-matching method eliminates the neces- 
sity for finding the correct impedance 
match by means of taps. The Q-match- 
ing device is an open quarter-wave section 
of line placed in series with the untuned 
transmission line. It has an input im- 
pedance matching the characteristic im- 
pedance of the line, and an output 
impedance matching the antenna input 
impedance. 

(2) Since Z„ is the characteristic impedance of 
the line and Z A is the terminal impedance 
of the antenna, then 

where Z Q is the characteristic impedance 
of the line used for the matching section. 
For example, if a 600-ohm line is used to 
feed an antenna with input impedance of 
70 ohms, then 




TM S66-88 

Figure 84- Matching with the shorted and open quarter-wave stubs. 



87 



Z Q =V(600)(70) 

=V42,000 

=205 ohms 

If a quarter-wave section of line having a 
characteristic impedance of 205 ohms is 
inserted between the transmission line 
and the antenna terminals, a perfect 
match results. 
(3) The Q match is flexible in that any type of 
line can be used for the matching section, 
provided only that it has the character- 
istic impedance necessary. The quarter- 
wave matching section can have either 
air or solid dielectric. The Q match, 
however, loses its effectiveness rapidly as 
the frequency of operation swings away 
from the resonant frequency of the 
antenna. To lessen this effect, the 
impedances can be matched in two steps 



1 W 1 

HIGH Z ZjZ LCC Z A 

1 <SISISU 1 



A 



instead of one. This requires two 
quarter-wave sections. 
(4) In the example given above, a 205-ohm 
quarter-wave section is used to match 
a 600-ohm line to a 70-ohm antenna 
impedance. In using two sections, an 
intermediate impedance is selected, and 
the first-section characteristic impedance 
found from the formula. For example, 
the lirst section matches the 600-ohm line 
to an impedance of 300 ohms. Then 

=V(600)(300) 



=Vl80,000 

= 425 ohms 

The characteristic impedance of the first 
section is 425 ohms. Its input impedance 
is 600 ohms, and its output impedance is 
300 ohms. The second quarter-wave 




B 



LOW Z 



X 



HIGH Z A 



LOW Z 



HIGH Z A 



c 

Figure 85. Artificial-line matching to antennas. 



D 

TM 666-89 



section then must have an input im- 
pedance of 300 ohms and an output 
impedance of 70 ohms. Its character- 
istic impedance therefore must be 

Z Q2 = V(300)(70) 
=V2 1,000 
= 145 ohms 

The second section then can be connected 
directly to the antenna terminals. As 
stated previously, there is no limitation 
on the type of line used to form the 
matching section. 
i. Coaxial Line Feed. Coaxial cable using a solid 
dielectric can be constructed with a characteristic 
impedance equal to the input impedance of a 
center-fed antenna. This provides an extremely 
simple method of feed. The inner conductor is 
connected to one leg of the antenna, and the outer 
conductor to the other end. The line is un- 
balanced in this arrangement, and a bazooka sec- 
tion, therefore, is used to restore balance. If the 
coaxial cable is not a direct match for the antenna, 
stub matching can be used in the same manner as 
with the two-wire stubs. This is not practical, 
because the line must be tapped on the stub to 
obtain the correct impedance match. A simple 
method is to use a Q-matching section of coaxial 
line. Open coaxial lines also can be used to feed 
antennas. These are found in special transmitting 
applications. Usually, the line is filled with a gas 
dielectric and sealed to preserve a uniform 
impedance. 
j. Artificial-Line Matching. 

(1) Matching systems are used because a mis- 
match between the line and the antenna 
causes standing waves to appear on the 



line. This condition lowers the efficiency 
of energy transfer from the transmitter to 
the antenna. Thus far in this paragraph, 
the methods evolved for matching a 
nonresonant line to an antenna have 
involved transmission-line sections, such 
as stubs or quarter-wave transformers. 
Since transmission lines have distributed 
constants, it is possible to replace the 
distributed constants of the line section 
with the lumped constants of coils and 
capacitors to obtain impedance matches. 
Methods using lumped constants are 
classified under the heading of artificial- 
line matching. 

(2) The chief advantage of the artificial line 
over the transmission-line section in im- 
impedance matching is the negligible 
amount of space occupied by lumped 
constants. Coils or capacitors in sealed 
weatherproof containers can be mounted 
right at the input terminals of the 
antenna. The main use of this type of 
line is at the lower frequencies, where 
equivalent transmission-line sections 
would have to be of excessive length. 

(3) Several typical feed systems using artifi- 
cial-line matching are shown in figure 85. 
In A and B, high impedance lines are 
used to feed a resonant antenna having 
low input impedance. The coil is divided 
equally between the two line conductors 
in order to maintain line balance. In C 
and D, the low-impedance line is used to 
feed a resonant antenna having high 
input impedance. The coils again are 
divided in order to preserve line balance, 
with the capacitor being placed across 
the line. 



Section IV. BASIC RADIATION PATTERNS 



52. Introduction 

The first three sections in chapter 3 covered the 
various circuits necessary to transfer energy from 
the transmitter to the antenna. This energy is 
radiated from the antenna and forms a field having 
a definite pattern, depending on the type of 
antenna used. It is desirable, therefore, to be able 
to put the radiation pattern of an antenna on 
paper where it can be examined. In fact, the 
antenna usually is designed to have a specific 



pattern for use with a particular installation 
The radiation pattern is a measure of the energy 
radiated from an antenna taken at various angles 
and at a constant distance from the antenna. 

53. Radiation Types and Patterns 

a. General. 

(1) A source of radiant energy such as the sun 
radiates light in all directions equally. A 
radiator of this type is known as an 



89 



2 



3 



i 
I 
I 

4 ♦ 

I 
\ 
\ 



10 
9 

8 - 



♦ 



5 X 



ALL POSITIONS ARE EQUAL 
DISTANCES FROM THE SUN 



O 

< 
a 
< 

a 

UJ 

<r 
< 

UJ 



7 
6 
5 

4 - 

3 - 
2 - 



1 1 1 

1 2 3 
POSITION ON CIRCLE 



— r~ 
4 



TM S66-90 



Figure 86. The sun as an isotropic source of radiation. 



isotropic source of radiation. This sim- 
ply means that the energy coming from 
the source is found to be constant at a 
fixed distance from whatever angle it is 
measured. Assume that a measuring 
device is moved in a circle around the 
sun. At any point along the circle, the 
distance from the measuring device to 
the sun is the same. The measured radi- 
ation also remains the same. 
(2) If the measured radiation is plotted 
against various positions taken along the 
circle around the sun, the result is the 
graph in figure 86. Assume that the 
radiation is measured on a scale of to 
10 units, and that the radiation meas- 



ured is shown to be seven units at each 
position. The graph of the measured 
radiation, therefore, is a straight line 
plotted against positions along the circle. 
(3) The graph of measured radiation just 
described is a form of radiation pattern. 
The straight line represents the radiation 
pattern of an isotropic source taken in 
the plane of the measuring circle. It is 
possible to have a radiator which emits 
stronger radiation in one direction than 
in another. Such sources are called 
anisotropic. An example, shown in figure 
87, is the ordinary flashlight. The beam 
illuminates only a portion of the total 
space surrounding the flashlight. If a 




< 
o 
< 
tr 

o 

UJ 
£E 

Z3 
V) 

< 

UJ 

s 



MAXIMUM RADIATION 




I 23456789 10 It 12 13 14 15 16 



POSITION ON CIRCLE 



TM 666-91 



Figure 87. Flashlight as anisotropic source of radiation. 



90 



circle is drawn having the light source as 
center, the radiation can be measured 
at different positions along the circle. 
Each position used for measurement is 
the same distance from the light source. 
In other words, conditions are exactly 
those used in measuring the light radiated 
from the isotropic source. 

(4) At position on the circle, which is 
directly behind the light source, the 
radiation measured is negligible. A zero 
value accordingly is assigned to this 
position on the graph at the right. 
Until position 4 is reached, the radiation 
remains negligible. Between 4 and 6, 
the circle passes from comparative dark- 
ness into the flashlight beam. This is 
an area of sharp transition from darkness 
to brightness, as can be observed easily 
on the graph. Radiation is relatively 
constant moving from positions 6 to 10, 
reaching a maximum at position 8, 
which is directly in the path of the beam. 
Between 10 and 12, the measured radia- 
tion falls off sharply, becoming and 
remaining negligible from 13 to 16. 

(5) Radiation from a light source and radia- 
tion from an antenna are both in the 
form of electromagnetic waves. The 
measurement of radiation from an an- 



tenna, therefore, follows the same basic 
procedure as the one just described for 
the sun and the flashlight. These meas- 
urements can be graphed to obtain a 
radiation pattern for the antenna. 
Before proceeding with the study of 
antenna patterns, however, it is desirable 
to understand in detail the various 
methods used to graph measured values 
of radiation. 
b. Rectangular-Coordinate Pattern. 

(1) In figures 86 and 87, the rectangular- 
coordinate type of graph is used to plot 
the measured value of radiation against 
the position at which the measurement is 
taken. For convenience, the graph of 
figure 86 is reproduced in A of figure 88. 
The numbered positions along the circle 
are laid out along the horizontal axis of 
the graph from to 7. The units of 
measured radiation are laid out along the 
vertical axis from to 10. Units on both 
axes usually are chosen so that the 
pattern occupies a convenient area of 
the graph. 

(2) The horizontal and vertical axt>s are at 
right angles to each other. The point 
at which the axes cross each other is 
called the origin. In this case, the origin 
has the value of zero on both axes. 




Figure 88. Comparison of rectangular-coordinate and polar-coordinate graphs for isotropic source. 



Now, assume that a radiation value of 
seven units is measured at position 2. 
From position 2 on the horizontal axis, a 
line (vertical dashes) is projected running 
parallel to the vertical axis. From 7 on 
the vertical scale, a line (horizontal 
dashes) is projected running parallel to 
the horizontal axis. The point at which 
the two lines intersect represents the 
value of seven radiation units at position 
2. Note that this is the only point on 
the graph that can represent this value. 

(3) The vertical and horizontal axes plus the 
two dashed lines used to plot the point 
inclose an area forming a rectangle 
(shaped area). It is for this reason that 
this type of graph is called a rectangular 
coordinate. A new rectangle is formed 
for each different point plotted. In the 
case given, all the points plotted lie 
along a straight line extending from 
seven units on the vertical scale to the 
projection of position 7 on the horizontal 
scale. The straight line, therefore, is the 
characteristic pattern in rectangular co- 
ordinates of an isotropic source of 
radiation. 

Polar-Coordinate Pattern. 

(1) Although the rectangular-coordinate 
method of graphical analysis is used 
widely, another method has proved to be 
of greater use in studying radiation pat- 
terns. This method is the polar-coordi- 
nate type of graphical analysis (B of fig. 
88). Note the great difference in the 
shape of the radiation pattern when it is 
transferred from the rectangular-coordi- 
nate to the polar-coordinate graph. 
The scale of values used in both graphs 
is identical, and the measurements taken 
are both the same. The basic difference, 
which results in the difference in physical 
appearance, is in the type of axes used. 

(2) In the rectangular-coordinate graph, 
points are located by means of projections 
from a pair of axes at right angles to each 
other. These axes remain stationary at 
all times. In the polar-coordinate graph, 
one axis consists of concentric circles, and 
the other axis consists of a rotating radius 
extending from the center of the concen- 
tric circles. Recall how radiation was 
measured by traveling in a circle around 



the sun. Assume a radius, R, drawn from 
the sun as center to position on the 
circle. Moving to position 1, tlue radius 
moves to position 1 ; moving to position 2, 
the radius also moves to position 2; and 
so on. This moving radius constitutes 
the moving axis of the polar-coordinate 
graph. 

(3) The positions of the radius are marked on 
the polar-coordinate graph for each posi- 
tion at which a measurement is taken. 
Note how the position of the radius indi- 
cates the actual direction from which the 
measurement was taken. This is a dis- 
tinct advantage over the rectangular- 
coordinate system, in which the position 
is indicated along a straight-line axis 
having no physical relation to the position 
on the circle. Having established the 
direction in which the measurement was 
taken by means of the rotating axis, it 
remains to devise means for indicating 
the measured radiation. 

(4) The rotating axis passes from the center 
of the graph to some position marked on 
the edge of the graph. In so doing, it 
intersects a set of concentric circles 
spaced at equal distances from each other. 
Going out from the center, the circles get 
larger and larger. These circles are used 
to indicate the measured radiation. 
They are numbered successively from 
the center outward, the center indicating 
a zero measurement. In the graph in 
B of figure 88, a radiation scale going 
from to 10 units is used. Consequently, 
10 concentric circles go from the center 
to the circumference of the graph. These 
circles are marked 1, 2, 3, and so on, with 
10 designating the largest circle. This 
scale corresponds to the scale marked on 
the vertical axis of the rectangular- 
coordinate graph in A. 

(5) Summing up, the rotating radius of the 
polar-coordinate graph serves the same 
purpose as the stationary horizontal axis 
of the rectangular-coordinate graph. It 
has the advantage of indicating the actual 
direction from which the measurement is 
taken. The concentric circles serve the 
same purpose as the verticle scale on 
the rectangular-coordinate graph. They 
allow the same scale to be used no matter 



what the position of the rotating radius. 
The distance from the source is constant 
for both types of graphs. 

(6) At position in B, the radius extends from 
the center outward to the right. The 
radiation measured is seven units in this 
position. This poiiit is recorded by go- 
ing out seven circles along the radius. 
The point is the place where the radius 
intersects the seventh circle. The re- 
cording of the radiation measured in 
position 1 follows the same procedure. 
Since the source is isotropic, the measured 
radiation again is seven units. The ra- 
dius is rotated to position 1, and its 
intersection with the seventh circle is 
marked. 

(7) When all points are recorded through 
position 7, it is found that they all lie on 
the seventh concentric circle. The radia- 
tion pattern of the isotropic source, there- 
fore, is a circle. This contrasts sharply 
with the straightline pattern obtained 
with a rectangular-coordinate- type graph. 
The advantages of the polar-coordinate 
graph are evident immediately. The 
source, which is at the center of the 
observation circle, is at the center of the 
graph. Also, the direction taken by the 
radiated energy can be seen directly from 
the graph. For these reasons, the polar- 



coordinate graph is more useful in 
plotting radiation patterns. 

(8) In figure 87, the radiation pattern of the 
common flashlight was graphed in rec- 
tangular coordinates. This graph is re- 
produced for convenience in A of figure 
89. From the physical picture of the 
flashlight beam, it is evident that the 
light source is anisotropic in nature. 
This is not evident in the radiation pat- 
tern traced on the rectangular-coordinate 
graph. Conversely, the radiation pat- 
tern of the flashlight shown in B of figure 
89 bears some physical resemblance to 
the actual beam. This is the same pat- 
tern, drawn using polar coordinates. 

(9) The positions on the circle marked off on 
the two polar-coordinate graphs given 
have been selected and numbered arbi- 
trarily. It is possible to mark off posi- 
tions around the circle in a standard way 
so that one radiation pattern can be 
compared easily with another. The 
standard method is based on the fact that 
a circle is divided into 360°. The radius 
extending from the center horizontally to 
the right (position in B of fig. 89) is 
designated 0°. Advancing to position 4 
rotates the radius until it is at right angles 
to the 0° radius. This radius position 
accordingly is marked 90°. Position 8 




TM 666-93 



t 



Figure 89. Comparison of rectangular-coordinate and polar-coordinate graphs for anisotropic source. 

93 



Is, therefore, 180°, position 12 is 270°, 
and position 16 is 360°, by the same 
reasoning. The various radii drawn on 
the graph all are marked according to the 
angle each radius makes with the refer- 
ence radius at 0°. 
(10) In B of figure 89, the polar-coordinate 
graph shows a definite area inclosed by 
the radiation pattern, indicating the 
general direction of radiation from the 
source. This area is called a lobe. Out- 
side of this area, no radiation is emitted 
in any direction. For example, at an 
angle of 45° (position 2), the radiation is 
zero. Such a point is called a null. 
Practically speaking, there is usually some 
radiation in every direction. A null, 
therefore, also is used to indicate direc- 
tions of minimum radiation. In the 



HORIZONTAL 
POLAR 
PATTERN 




ANTENNA 



HORIZONTAL 
PLANE 



ANTENNA 
(INSIDE) 



pattern given, there is one lobe and one 
continuous null. 

54. Dipole Antenna Radiation 

a. Definitions. In the following discussion, the 
term dipole is used to mean the basic half-wave 
antenna. The term doublet is used to indicate an 
antenna that is very short compared with the 
wavelength of the operating frequency. Physi- 
cally, it has the same shape as the dipole. 
6. Radiation Pattern of a Doublet. 

(1) The doublet is the simplest form of prac- 
tical antenna. Since it is a source of 
radiation, its radiation can be measured 
and a radiation pattern plotted in the 
manner in which a flashlight and its 
beam were shown in figure 87. Figure 
90 is a perspective view of the radiation 



VERTICAL 
PATTERN 



NULL 



LOBE 



SOLID 
PATTERN 




VERTICAL 
PLANE 



NULL' 



TM 666-94 



Figure 90. Development of vertical- and horizontal-plane polar patterns from solid radiation pattern. 



94 



pattern of a doublet. This is not a picture 
of the radiation, but a three-dimensional 
view of the pattern itself. In three 
dimensions, the pattern resembles a 
doughnut. 

(2) From this perspective view, two types of 
polar-coordinate patterns can be drawn. 
The first is obtained by getting a top 
view of the doughnut in a horizontal 
plane through its center. This plane is 
the same as that of the circle drawn with 
dashes in the solid pattern. Looking 
down on the horizontal plane, the antenna 
axis is seen head on, so that it becomes 
simply a dot in the horizontal polar 
pattern. It can be seen from the hori- 
zontal pattern that the radiation is con- 
stant in any direction along the hori- 
zontal plane. 

(3) A vertical plane view of the doughnut 
pattern can be drawn, from which is 
obtained a vertical polar pattern. To 
obtain this pattern, the doughnut is 
sliced in half along a vertical plane 



(4) 



through the antenna. This can be seen 
in the figure to the right of the solid 
pattern. Note how the vertical plane 
view of the radiation pattern differs 
sharply from the horizontal plane view. 
The vertical pattern exhibits two lobes 
and two nulls. The difference in the 
vertical pattern is caused by the fact that 
no radiation is emitted from the ends of 
the doublet. Also, there is maximum 
radiation from the doublet in a direction 
perpendicular to the antenna axis. This 
type of radiation pattern is both non- 
directional (in a horizontal plane) and 
directional (in a vertical plane). 
From a practical viewpoint, the antenna 
can be mounted vertically or horizon- 
tally. The doublet shown in figure 90 is 
mounted vertically, and the radiated 
energy spreads out about the antenna 
nondirectionally in the horizontal plane. 
Since this usually is the useful plane, this 
arrangement is termed nondirectional, 
and its directional characteristic in other 




HORIZONTAL 
PLANE 
PATTERN 



VERTICAL 

PLANE 
PATTERN 



180* 



SOLID PATTERN 




TM 666-95 



Figure 91. Radiation pattern of dipole (half-wave) antenna. 



95 



planes is ignored. If the doublet is 
mounted horizontally, it has the effect of 
turning the pattern on edge, reversing the 
patterns given in figure 90. The antenna 
is now directional in the horizontal plane. 
The terms nondirectional, directional, 
and so on are used for convenience in 
describing specific radiation patterns. 
A complete description always involves 
a figure in three dimensions, as in the 
solid pattern of figure 90. 
c. Radiation Pattern of a Dipole. The radiation 
pattern of a dipole is similar to that of the doublet. 
Increasing the length of the doublet until it is a 
half-wavelength has the effect of flattening out the 
doughnut pattern (fig. 91), or forming a flattened 
figure 8. The radiation pattern in the horizontal 
plane is a larger circle than in the doublet. The 
vertical radiation-pattern lobes no longer art 
circular. They are flattened out, and the radia- 
tion intensity is relatively greater. The elonga- 
tion of the pattern is greatest perpendicular to the 
antenna axis. On the vertical radiation pattern, 
0° is used to indicate a position off one end of the 
antenna. All angular measurements around the 
graph are made from this point. A position at 
right angles to the source of radiation accordingly 
is designated 90° or 270°. A position off the 
opposite end of the antenna is marked 180°, and 
so on. This method of starting at one end of the 
antenna axis is conventional. 

55. Using Radiation Pattern 

a. Antennas usually are constructed to obtain a 
specific radiation pattern. Actual tests then are 
conducted to discover whether the practical 
antenna radiation pattern conforms to the pattern 
desired. Field tests of this type usually are 
carried out by measuring the relative field strength 
of the antenna in terms of microvolts per meter. 
This measurement utilizes the E field of the 
antenna. Since power is directly proportional to 
the square of voltage (P=E 2 /R), the measure- 
ments obtained can be used to plot a radiation 
pattern. This is the simplest and most common 
method for taking measurements of antenna field 
strength. Measurements involving actual power 
require elaborate equipment. 

6. Since two types of pattern can be obtained, 
the voltage measurements can be used to plot a 
field-strength pattern, or can be squared to plot a 
power pattern. These patterns indicate relative 



270 




HALF- POWER 
POINTS 



0* 270 




180" 
RELATIVE 
FIELD STRENGTH 



180° 
RELATIVE 
POWER 



B 



TM 866-96 



Figure 92. Beam width measured on relative field strength 
and relative power patterns. 

field strength and relative power. In practice, 
there is little difference between the patterns, and 
both are used. Relative field-strength and rela- 
tive power patterns for the same source of radia- 
tion are illustrated in figure 92. Maximum 
radiation is taken to be 1, and radiation in every 
other direction is expressed in terms of fractions of 
the maximum. 

c. Although the radiation pattern consists of 
areas inclosed by curves, it must be remembered 
that the actual radiation streams out from the 
source for great distances. The radiation takes 
the general directions indicated by the radiation- 
pattern lobes. There is little or no radiation in 
the directions indicated by nulls. For convenience 
in indicating the general direction of radiation, the 
term beam width is used. The radiation beam is 
considered to leave the antenna between the points 
where the field strength falls off to 0.707 of maxi- 
mum, or where the power falls off to 0.5 of 
maximum. 

d. In A of figure 92, the field strength is maxi- 
mum at 90° and 180°. There are nulls at 0° and 
270°. Between each null and maximum, the field 
strength rises from zero to one. The angles at 
which the voltage is 0.707 are marked in the 
figure. The beam is assumed to be contained 
within the total angle from 0.707 through a maxi- 
mum to 0.707. This total angle is the beam width. 
In this case, the beam is 90° wide, as indicated in 
the figure. 

e. The field strength pattern in A is related 



96 



directly to the power pattern in B. This is 
because the power is directly proportional to the 
square of the voltage. The same points can be 
found on the power pattern by taking the square 
of the voltage values used to determine the beam 
width. 

::(0.707) 2 
:: .5 

where P, is the relative power and E T is the relative 



voltage. The beam width on the relative power 
pattern, therefore, is found by locating the half- 
power points. The beam width in B is also 90°. 
In other words, the beam width is constant, which- 
ever type of graph is used. 

/. When looking at the plane patterns in figure 
92, bear in mind that they are cross sections of 
solid radiation patterns. Although there are two 
lobes in the plane radiation pattern, there is only 
one lobe in the solid pattern, and only one beam. 
It is possible to have more than one beam, as will 
be shown in chapter 4. 



Section V. PRACTICAL HALF-WAVE ANTENNAS 



56. Introduction 

q. The half- wave antenna has been discussed 
previously without reference to the effect produced 
by the presence of ground on the radiation pattern. 
Since all practical antennas are erected over the 
earth and not out in free space, it is necessary to 
determine just what effect the ground produces. 
The presence of ground may alter completely the 
radiation pattern produced by the antenna, and 
ground also will have an effect on some of the 
electrical characteristics of the antenna. 

b. In general, the ground has the greatest effect 
on those antennas which must be mounted fairly 
close to it in terms of wavelength. For example, 
medium- and high-frequency antennas elevated 
above ground by only a fraction of a wavelength 
will have radiation patterns that are quite different 
from the free-space patterns. 

c. In addition to ground effects, several examples 
of practical half-wave antennas are discussed in 
this section. These include the conventional 
single-wire, half-wave dipole, the folded dipole, 
the coaxial antenna, and the conical antenna. 

57. Ground Effects 

a. Assume that a horizontal half-wave antenna 
is erected at a vertical distance, H, from a ground 
plane, as illustrated in figure 93, where B is the end 
view. Some of the energy that leaves the antenna 
travels directly to some distant point, P. This 
energy is referred to as the direct wave in the 
figure. The direction followed by the direct 
wave makes a certain angle, A, in respect to the 
horizontal. 

b. Some of the energy leaving the antenna 
travels downward toward the ground plane. Since 
this ground plane is a good conductor, the down- 



ward traveling wave from the horizontal antenna 
is reflected with practically no loss and a reflected 
wave of energy travels outward toward the distant 
point, P. The angle between the reflected wave 
and the perpendicular is exactly equal to the angle 
between the downward traveling wave from the 
antenna and the perpendicular. 

c. During the actual reflection process, a 180 
phase shift takes place so that the reflected wave 
is 180° out of phase with the direct wave. The 
highly conducting ground plane then cannot 
sustain the horizontal lines of electric force pro- 
duced by the antennas. In order to produce zero 
voltage along the ground, an electric field is 
assumed to be produced that is equal in amplitude 
but opposite in direction to that produced by the 
antenna. If a vertical antenna is used, the electric 
field is vertical. Under these conditions, no phase 
reversal takes place during the actual reflection. 

d. Regardless of whether a phase shift is pro- 
duced, the distant point, P, is acted on by .wo 
waves. One of these waves is the direct wave and 
the other is the reflected wave. The reflected 
wave travels a greater distance than does the 
direct wave. Therefore, there is an additional 
phase shift so far as the reflected wave is concerned, 
because of the greater distance it must travel. 
For example, if the reflected wave travels a distance 
to point P that is a half-wavelength longer than 
the distance traveled by the direct wave, an 
additional 180° phase shift will result from the 
greater path length. 

e. The signal strength at point P depends on 
the amplitudes and phase relations of the direct 
and reflected waves. If the ground is a good 
conductor so that very little absorption of energy 
occurs during the reflection process, the reflected 
wave has the same amplitude as the direct wave. 



97 




Ttt «««-»T 

Figure 93. Reflection produced by ground plane. 



If these two equal-amplitude waves arrive at the 
distant point in phase, the resultant signal strength 
is twice that of the direct wave alone. On the 
other hand, if these waves arrive 180° out of 
phase, the resultant signal strength is zero. Inter- 
mediate values of signal strength occur with 
intermediate phase relations between the reflected 
and the direct wave. 

/. Assume that point P is so located that it 
receives twice the signal strength. Now assume 
that a second point, Q, is located slightly below 
point P. The distances to point Q might be such 
that the direct and reflected waves arrive 180° out 
of phase. As a result, cancelation occurs, the 
received signal strength is zero, and a null is 
produced. Because of ground reflections, then, 
it is possible that the radiation pattern may be 
broken up into a series of lobes. The signal 
strength at the center of the lobes will be about 
twice that which would be received if the antenna 
were not in the vicinity of a ground plane. These 
lobes are separated by nulls where the received 
signal strength is zero. 

g. It is sometimes convenient in making calcu- 
lations to use the idea of an image antenna. This 
is an imaginary antenna assumed to be located 
the same distance, H, below ground as the actual 
antenna is located above ground. The reflected 
wave is assumed to come from the image antenna, 
as shown in B of figure 93. When a horizontal 
antenna is used, to take into account the phase 
reversal that takes place when reflection occurs, 
the current in the image antenna is assumed to be 



180° out of phase with the current in the actual 
antenna. When a vertical antenna is used, the 
current in the image antenna is considered to flow 
in the same direction the current flows in the 
actual antenna. 

58. Ground-Affected Radiation Patterns 

a. Reflection Factor. 

(1) The reflection factor is a term by which 
the free-space radiation pattern of an 
antenna must be multiplied in order to 
determine the radiated field strength of a 
practical antenna at a given vertical 
angle. The maximum value of the 
reflection factor is 2. At those vertical 
angles, the direct and reflected waves are 
in phase, and twice the free-space signal 
strength occurs. The minimum value of 
the reflection factor is 0. At those ver- 
tical angles, the direct and reflected 
waves are of opposite phase, and com- 
plete cancelation occurs. The reflection 
factor then may vary from to 2 at 
vertical angles measured above the plane 
of the ground. Reflection factors are not 
given for angles below the surface of the 
earth. 

(2) The value of the reflection factor depends 
on the height of the antenna above the 
ground plane as well as the orientation. 
The following chart gives the value of 
the factor for horizontal half-wave anten- 



98 



nas at various vertical angles when the 
antenna is located a quarter-wavelength 
above ground: 



Vertical angle 
(degrees) 


Reflection 
factor 


Vertical angle 
(degrees) 


Reflection 
factor 







. 5 
1. 
1. 5 
1. 75 


50 


1. 8 

1. 95 

2. 
2. 
2. 


10 


60 


20 


70 


30 


80 


40 


90 







(3) When the horizontal antenna is located 
a half-wavelength above ground, the follow- 
ing chart can be used to obtain the re- 
flection factor: 



Vertical angle 
(degrees) 


Reflection 
factor 


Vertical angle 
(degrees) 


Reflection 
factor 







1. 

1. 75 

2. 
1. 75 


50 


1. 4 
. 75 
. 4 
. 1 




10 


60 


20. 


70 


30 


80 


40 


90 







(4) When the horizontal antenna is located 
S quarter-wavelengths above ground, the 
following chart is used to obtain the re- 
flection factor: 



Vertical angle 
(degrees) 


Reflection 
factor 


Vertical angle 
(degrees) 


Reflection 
factor 







1. 5 

2. 
1. 5 



50 


1. 

1. 7 

1. 9 

2. 
2. 


10 


60 


20 , 


70 


30 


80 


40 


90 







(5) When a height of 1 wavelength above ground 
is used, the following chart shows the re- 
flection factor: 



Vertical angle 
(degrees) 


Reflection 
factor 


Vertical angle 
(degrees) 


Reflection 
factor 







1. 8 
1. 6 


1. 6 


50 


1. 95 
1. 4 

. 6 

. 1 




10- 


60 


20 


70 


30. 


80 


40 


90 







(6) These charts also can be used with a 
half-wave vertical antenna. The height 
above ground is measured from the center 
of the vertical antenna. It is necessary, 
however, to subtract the given values of 
reflection factor from 2. Then, if the 
reflection factor given in the charts for a 
certain height and vertical angle is 1, 
the reflection factor for a vertical antenna 
is 2 minus 1, or 1. If the reflection 
factor for the horizontal antenna is 2, 
the factor for the vertical half-wave 
antenna is 2 minus 2, or 0. If the re- 
flection factor for the horizontal antenna 
is 0, then the vertical antenna reflection 
factor is 2 minus 0, or 2. 
b. Horizontal Half-Wave Antenna. 

(1) When the foregoing reflection factors are 
applied to the free-space radiation pat- 
tern of a horizontal half-wave antenna, 
the patterns shown in figure 94 are pro- 
duced. Patterns A, C, E, and G are the 
vertical radiation patterns in the plane 
of the antenna itself. B, D, F, and H 
are the vertical radiation patterns in the 
plane which is at right angles to the 
antenna. Patterns A and B are for 
antenna heights of a quarter-wavelength; 
C and D are for antenna heights of a 
half -wavelength; E and F are for heights 
of 3 quarter-wavelengths; 6 and H are 
for heights of 1 wavelength. 

(2) Figure 95 permits a better visualization 
of the radiation pattern produced. Here 
the actual solid radiation pattern is 
shown for a horizontal half-wave antenna 
located a half-wavelength above ground. 
In the vertical plane at right angles to 
the antenna, D of figure 94 shows two 
large lobes the maximum values of which 
occur at an angle of 30° with the hori- 
zontal. This pattern is reproduced in 
perspective at the upper left of figure 95. 
In the vertical plane which included the 
antenna, reference to C of figure 94, 
shows two small lobes with maximum 
values occurring at an angle of about 
40° with the horizontal. This pattern 
is reproduced in perspective at the upper 
right of figure 95. If these two plane 
views are connected smoothly, the solid 
pattern shown in the center of figure 95 
is produced. 



99 




100 



Figure 94- Vertical-plane radiation patterns of horizontal half-wave antennas. 



TM S66-SB 



Figure 95. Solid pattern produced by horizontal half-wave antenna located a half-wavelength above ground. 



(3) In a similar manner, solid radiation pat- 
terns can be visualized from the plane 
views shown in figure 94. Picture the 
pattern as being produced by the smooth 
transition from one vertical plane view 
shown to the other vertical plane view 
shown as an angle of 90° is covered. 

(4) Although vertical patterns are shown for 
only four specific heights above ground, 
it is not too difficult to predict the pat- 
terns produced at intermediate heights. 
This is true since the patterns do not 
change abruptly as the height of the 
antenna is increased gradually. Instead, 
there must be a smooth transition from 
the pattern shown for a height of a 
quarter-wavelength, to the pattern shown 
for a height of a half- wavelength. 

(5) At heights less than a quarter-wavelength 

above ground, the vertical patterns pro- 
duced by a horizontal half-wave antenna 
are almost perfectly circular. As the 
antenna is raised, the vertical pattern is 
flattened somewhat at its top, at a ver- 
tical angle of 90° (B of fig. 94). As the 
height is increased above a quarter-wave- 
length, a depression begins to appear at 
the top of the pattern, and the pattern 
width increases. The depression grows 
deeper and deeper as the antenna height 



approaches a half-wavelength. Finally, 
at a height of a half-wavelength, the 
pattern splits into two separate lobes. 
The radiation at a vertical angle of 90° 
(straight up) is zero at this height, as in 
D of figure 94. As the antenna height 
increases still more, a lobe of radiation 
begins to grow out of the center of the 
pattern at a vertical angle of 90°. As this 
lobe increases in amplitude with increas- 
ing antenna height, the two side lobes are 
spread farther apart so that their maxima 
occur at lower vertical angles. This ver- 
tical lobe has its maximum amplitude 
and begins to flatten somewhat (F of fig. 
94) at an antenna height of 3 quarter- 
wavelengths. As the antenna height is 
increased still more, the vertical lobe 
develops a depression that grows deeper 
as the height is increased. Finally, at a 
height of 1 wavelength, the center lobe 
splits into two separate lobes and the 
radiation at a vertical angle of 90° is 
again zero. Now four distinct lobes exist 
(Hoi fig. 94). 
(6) The patterns that are produced at antenna 
heights in excess of 1 wavelength also can 
be determined by studying figure 94. 
When the height of the horizontal an- 
tenna is an odd number of quarter-wave- 



101 



lengths above ground, a lobe of maximum 
radiation is produced at a vertical angle 
of 90° straight up. 

(7) Consider an antenna with a height of 1 
quarter-wavelength above ground. As- 
sume that the instantaneous electric field 
immediately around the antenna is maxi- 
mum in a given direction, designated as 
positive. A portion of this field moves 
downward a distance of a quarter-wave- 
length to the ground. Upon being re- 
flected, a 180° phase shift occurs, and the 
instantaneous electric field is now maxi- 
mum in the opposite direction, designated 
as negative. This negative field now 
moves upward from the ground for a 
distance of a quarter- wavelength. By the 
time the reflected field returns to the 
antenna, a total distance of a half-wave- 
length has been covered. Meanwhile, 
since one-half cycle of operation has 
elapsed during the time required for the 
downgoing wave to move from the an- 
tenna to the ground and from the ground 
back to the antenna again, the polarity 
of the energy on the antenna itself has 
reversed. As a result, the reflected wave 
arrives back at the antenna in exactly the 
right phase to reinforce the direct wave. 
The reinforcement occurs not only at 
heights of a quarter-wavelength above 
ground, but also at heights of 3 quarter- 
wavelengths, 5 quarter-wavelengths, 7 
quarter-wavelengths, and so on. Con- 
sequently, a lobe of maximum radiation 
is produced at a 90° vertical angle for all 
antenna heights which are an odd number 
of quarter-wavelengths from ground. 

(8) When the height of the antenna is an 
even number of quarter-wavelengths 
above ground, a null (zero radiated 
energy) occurs at the 90° vertical angle. 
Consider the action of the horizontal 
half-wave antenna that is located at a 
distance of a half-wavelength above 
ground. The portion of the radiated 
field from this antenna which travels 
downward toward the ground must cover 
a total distance of 1 wavelength before 
it arrives back at the antenna. The direc- 
tion of this field is reversed by the reflec- 
tion process. During the time that is 
required for the reflected wave to cover 



this distance, the field immediately sur- 
rounding the antenna has ggne through 1 
complete cycle and is now back to its 
original direction or polarity. The re- 
flected wave, therefore, with its field 
reversed by the reflection process, be- 
comes 180° out of phase with the direct 
wave from the antenna. As a result, 
cancelation occurs at the vertical angle 
of 90°, and a null is produced. This 
cancelation, as described above, occurs 
not only at heights of a half-wavelength 
above ground, but also at heights of 1 
wavelength, 1% wavelengths, and so on. 
Consequently, null is produced at a 90° 
vertical angle for all antenna heights 
which are an even number of quarter- 
wavelengths from ground. 
(9) One other factor can be observed from 
the patterns in figure 94 which can be 
used to determine the vertical radiation 
pattern of a horizontal half-wave an- 
tenna at heights greater than are shown. 
At a height of 1 quarter-wavelength 
above ground, the radiation pattern is 
seen to consist of one lobe only. At a 
height of 2 quarter-wavelengths (X/2) 
above ground, the radiation pattern con- 
sists of two lobes. At a height of 3 
quarter-wavelengths, the pattern con- 
sists of three lobes. At a height of 4 
quarter-wavelengths (X) above ground, 
the radiation pattern consists of four 
lobes. Consequently, the number of 
vertical lobes produced is numerically 
equal to the height of the antenna above 
ground in quarter-wavelengths and con- 
tinues for any antenna height. It is pos- 
sible to get a fairly good idea of the 
vertical radiation pattern of a horizontal 
half-wave antenna at any height above 
ground. For example, if the antenna is 
located at a height of 2 wavelengths 
above ground, which is an even number 
of quarter-waves, a null is produced at a 
vertical angle of 90°. Then, since 2 
wavelengths represent 8 quarter-wave- 
lengths, the radiation pattern consists of 
eight lobes. 
c. Vertical Half -Wave Antenna. 

(1) When the proper reflection factors are 
applied to the free-space radiation pat- 
tern of a vertical half-wave antenna, the 



102 



A 



B 




Figure 97. Solid pattern produced by vertical half-wave antenna located a half-wavelength above ground. 

103 



patterns shown in figure 96 are produced. 
Only, a simple plane view need be shown 
here because the vertical half-wave an- 
tenna is nondirectional in the horizontal 
plane. Its free-space horizontal radia- 
tion pattern is a circle. Therefore, the 
effect of the reflection factor is the same 
in all horizontal directions. 

(2) To visualize more clearly the solid radia- 
tion pattern, it is necessary only to 
picture the plane patterns shown in 
figure 96 being rotated. One such solid 
radiation pattern is shown in figure 97, 
where the pattern is produced by a 
vertical half-wave antenna the center of 
which is a half-wavelength above ground. 

(3) In general, two effects are shown when the 
patterns of figure 96 are observed. First, 
there is always a null at the vertical angle 
of 90° because there is no radiation from 
the end of the vertical antenna. There- 
fore, regardless of the value of the reflec- 
tion factor at this angle, no radiation occurs 
directly upward. At all antenna heights, 
then, the vertical half-wave antenna 
produces a null at 90°. The second 
effect noted is that, as the antenna is 
raised above ground, a greater number 
of lobes appear in the pattern. At a 
height of 1 quarter-wavelength, for 
example, two lobes appear (A of fig. 96) . 
When the antenna is raised to 1 half- 
wavelength, four lobes appear, as in B. 
The amplitude of the upper lobes is 
much smaller than that of the lobes 
which lie along the ground. At a 
height of 3 quarter-wavelengths, there 
are still four lobes, but the amplitude 
of the upper lobes has increased, as 
shown in C. When the antenna is 
raised to a height of a full wavelength, 
as in D, six lobes appear. 

59. Changes in Radiation Resistance 

a. The radiation resistance measured at the 
center of a half-wave antenna in free space is 
73 ohms. However, the radiation resistance of a 
practical half-wave antenna located over a ground 
plane may have any value of radiation resistance 
from to almost 100 ohms. The exact value 
of radiation resistance depends on the height 
of the antenna above ground. 



b. The change in radiation resistance occurs 
because of the effect of the wave reflected up 
from the ground plane. This reflected wave 
induces a voltage which causes a current to flow 
in the antenna. The phase of this induced 
current in respect to the current in the antenna 
produced bv the transmitter depends on the 
height of the antenna and its orientation. At 
some antenna heights, it is possible for the two 
currents to be in phase so that the total antenna 
current is greater than it would be if no ground 
reflection had taken place. At other antenna 
heights, the two currents may be 180° out of 
phase so that the total antenna current is less 
than it would be if no ground reflection had 
occurred. Intermediate antenna heights result 
in induced currents having different phase rela- 
tions. Therefore, the total antenna current varies 
widely, depending on the antenna height. 

c. If a given input power is applied to an 
antenna and the antenna current increases, the 
antenna behaves as though its resistance were 
reduced. Since the ohmie resistance of the 
antenna does not change, the radiation resistance 
is lowered effectively. Similarly, if the antenna 
height is such that the total antenna current 
decreases, the antenna radiation resistance is 
increased. 

d. The actual variation in radiation resistance 
of a half-wave antenna at various heights above 
ground is shown in the graph in figure 98. The 
solid curve shows the radiation resistance of a 
horizontal half-wave antenna, and the dashed 
curve shows the radiation resistance of a vertical 
half-wave antenna. The radiation resistance of 
the horizontal antenna rises steadily to a maxi- 
mum value of 98 ohms at a height of about 
3 eighths-wavelengths. Then the radiation resist- 
ance falls steadily to 58 ohms at a height of about 
5 eighths-wavelengths. The resistance then con- 
tinues to rise and fall around an average value 
of 73 ohms, which is the free-space value. As the 
height is increased, the amount of variation keeps 
decreasing. The curve is similar to a damped 
oscillation. 

e. The variation in radiation resistance of a ver- 
tical antenna is much less than that of the hori- 
zontal antenna. The radiation resistance is a 
maximum value of 100 ohms when the center of 
the antenna is a quarter-wavelength above ground. 
The value falls steadily to a minimum value of 70 
ohms at a height of a half-wavelength above 
ground. The value then rises and falls by several 



104 



RADIATION RESISTANCE OP 
"FREE- SPACE X/2 ANTENNA 



UJ 

o 
z 
< 

<n 
U> 

UJ 

il 

< 

a 
< 



100 
90 
80 
70 
60 
50 
40 
30 
20 
10 



/ 


V \ 
















A \ 
















1 — s — V 

\ \ 
































/I 
















r 
















VERT 


ICAL ANTENNA 


























"HORI 


ZONTAL ANTENNA 









































\ \ 3 \ * 5 \ *\ 2% 



HEIGHT OF \/Z ANTENNA ABOVE GROUND 
(WAVELENGTHS) 

TM 666-102 

Figure 98. Radiation resistance at various heights. 

ohms about an average value slightly above the 
free-space value of a horizontal half-wave antenna. 

/. Because of the variation in antenna current 
and radiation resistance at various antenna heights, 
the field intensity produced by a given antenna 
also changes. In general, as the radiation re- 
sistance is reduced, the field intensity increases. 
An increase in radiation resistance results in a drop 
in the radiated field intensity. 

60. Effects of Practical Grounds 

a. General. 

(1) Up to this point, all of the effects pro- 
duced have been the result of a ground 
which has a uniform high conductivity. 
In practice, the nature of the ground over 
which the antenna is erected is subject to 
considerable variation. This results not 
only from the ground material itself but 
also the manner in which it is found. 
For example, the antenna may be erected 
over a ground which has high or low con- 
ductivity, and over a ground with uni- 
form or nonuniform conductivity in the 
vicinity of the antenna. All of these 
characteristics of the ground have an 
effect on the radiation patterns and re- 
sistance of the antenna. 

(2) Table VI gives the approximate relative 
conductivities of various surfaces that 
may be found under an antenna. Note 
the wide variation in conductivity that 
occurs. It is not strange then that a 



considerable variation in ground effects 
occurs over different types of surfaces. 

Table VI. Ground Material Conductivity. 



Oround material 



Sea water 

Flat, rich soil 

Average flat soil 

Fresh water lakes... 

Rocky hills 

Dry, sandy, flat soil. 
City residential area 
City industrial area. 



Relative 
conductivity 



4, 500 
15 
7 
6 
2 
2 
2 
1 



Ground Losses. 

(1) Unless the ground behaves as a nearly 
perfect conductor, the amplitude of the 
ground-reflected wave will be much less 
than the amplitude of the wave before 
reflection. A portion of the wave which 
ordinarily would be reflected is absorbed 
by the resistance of the ground. Such 
absorption by the ground constitutes 
ground / losses. This means that the 
value of the ground reflection factor 
will be reduced considerably. 

(2) The maximum value of the ground 
reflection factor is 2 (par. 58a(l)). 
At those values of vertical angle at 
which the factor is 2, the total signal 
strength (resulting from the direct and 
ground-reflected waves) is twice the 
value of the direct wave alone. If the 
ground is a poor conductor, the maximum 
value of the ground reflection factor is 
much less than 2. As a result, the 
maximum value of the lobes never 
rises to twice the value without ground 
reflections. In addition, no nulls are 
produced over an imperfect ground. 
To produce a null, the ground-reflected 
wave must be 180° out of phase with the 
direct wave and must be of equal ampli- 
tude. As a result, complete cancelation 
occurs and a null results. If the ground 
is imperfect, some of the wave that 
would be reflected is absorbed instead. 
Ground losses occur, and the amplitude 
of the ground-reflected wave is reduced. 
Consequently, complete cancelation can- 
not occur at a given vertical angle. 

105 



Therefore, instead of an actual null being 
produced, a reduction in resultant signal 
strength occurs. 
(3) The value of ground reflection factor 
over a perfectly conducting ground 
(that is, one with no ground losses) 
varies from (I to 2. Nulls are produced 
at those values of vertical angle where 
the factor is 0, and lobes of double 
signal st rength occur where the factor is 2. 
Over a moderately conducting ground, 
the factor may increase to a maximum 
value of only 1.5 and may drop to a 
minimum value of 0.5. As a result, the 
vertical radiation pattern shows a series 
of high and low signal strengths rather 
than a series of double-amplitude lobes 
separated by well defined nulls. When 
the ground acts as a very poor conductor, 
practically all of the energy directed 
down toward the ground is absorbed 
and ground reflections do not occur. 
The ground reflection factor is then 1 
over a wide range of vertical angles, 
resulting in a free-space pattern. 

c. Frequency /effects. 

(1) From the preceding discussion of the 
effects of imperfect grounds and from 
the relatively low conductivities of all 
surfaces (table VI), it would appear 
that the entire previous discussion con- 
cerning ground effects is not too impor- 
tant. However, such is not the case. 

(2) At low and medium frequencies, the 
radio wave that strikes the ground 
causes ground currents to flow which 
penetrate the ground to a depth of 50 
feet or more. In general, a greater 
penetration occurs when the top layer 
of ground has a low conductivity. 
Consequently, even though the actual 
conductivity of the soil itself may not 
be great, the volume of soil in which 
current can flow is considerable. As a 
result, the resistance of the ground is 
low, and, for all practical purposes, most 
types of soil act as rather good reflectors. 
Only a relatively small amount of ground 
loss occurs and the ground reflection 
factors vary from to 2. The vertical 
radiation patterns shown previously in 
this chapter then apply almost exactly 



when low and medium frequencies are 
used. 

(3) At, higher frequencies (3 to 30 mc), the 
depth of penetration of a radio wave into 
the earth is limited to about 5 to 10 feet. 
Unless the antenna is erected over salt 
water or over a very highly conducting 
soil, considerable ground losses occur and 
much absorption of radiated energy oc- 
curs at vertical angles less than about 
10° to 12°. At very low vertical angles 
(approximately 1° to 3°) so much ab- 
sorption occurs because of ground losses 
that the ground reflection factor is re- 
duced to a very low value, and the 
vertical-plane radiation patterns shown 
in figure 96 must be modified to take 
this factor into account. These patterns 
should show little or no radiated field 
intensity at very low vertical angles. 
The large lobe of radiation which lies 
along the ground plane then is reduced 
in amplitude, and a null is produced 
along the ground. 

d. Radiation Resistance Effects. 

(1 ) The graph in figure 98 shows the variation 
in radiation resistance for half-wave 
antennas at various heights above 
ground. The curves in this figure have 
been plotted for antennas erected above 
a highly conducting ground. If an im- 
perfect!} 7 conducting ground is used, the 
curves shown must be modified. 

(2) In general, the use of an imperfectly 
conducting ground shifts the curves 
shown slightly toward the left. In addi- 
tion, the curves do not rise to as high 
values nor do they fall to as low values 
as when a highly conducting ground is 
used. The effect is to smooth out the 
curves and reduce the amount of change 
in radiation resistance as the antenna 
height is increased above ground. 

e. Antenna Height. 

(1) It is not possible always to answer ac- 
curately the question of what determines 
the exact height of a given antenna above 
ground. It might be assumed that it is 
necessary simply to measure the distance 
between the antenna itself and the surface 
of the ground. This method, however, 
may not give accurate results so far as 
reflection factors and vertical-plane pat- 



106 



terns are concerned. Instead, several 
feet must be added to the actual meas- 
ured height. It is just as though ground 
reflections take place from a plane located 
a few feet below the surface of the ground. 
(2) If the ground were a perfect conductor, 
no penetration of the radio wave would 
occur. Under these conditions, reflection 
takes place at the surface of the ground, 
and the actual height above ground can 
be used. In an imperfectly conducting 
ground, reflection seems to occur from a 
plane that is located below the surface 
and the actual depth of the reflecting 
plane is determined largely by the nature 
of the ground and the frequency used. 
This depth may be considerably greater 
than the few feet mentioned above. 
Since it is difficult to determine the actual 
depth of the ground reflecting plane, the 
exact effect on the radiation patterns 
and resistance of the antenna cannot be 
determined precisely, but sufficiently ac- 
curate results are obtained for all prac- 
tical purposes. 

61 . Ground Screens 

a. A ground screen consists of a fairly large area 
of metal mesh or screen which is laid on the surface 
of the ground under the antenna. Special mesh 
made of copper, Copperweld, or galvanized iron 
is available in large sheets for this purpose. 
Ordinary chicken wire also can be used although it 
will introduce some losses. When sheets of mesh 
are used, they must be bonded together at several 
places to keep the over-all resistance of the ground 
screen low. Although a ground screen simply can 
be laid on the surface of the ground, lower losses 
occur if it is connected to the earth by means of 
ground rods. The rods are driven into the earth to 
a depth of 4 to 8 feet. The metal screen then is 
bonded to the rods. Sometimes a ground screen is 
laid on a wooden framework that is erected 8 to 
12 feet off the ground. The metal mesh usually is 
stapled to the framework; the grounding wires 
are run from the mesh to ground rods. 

b. The purpose of the ground screen is to simu- 
late to some extent the effect of a perfectly con- 
ducting ground under the antenna. The screen 
should extend for a considerable distance in every 
direction from the antenna. In practice, however, 
the ground screen seldom extends more than a 



half-wavelength or slightly less in all directions. 

c. Two specific advantages are to be gained 
when a ground screen is used. First, the ground 
screen reduces ground absorption losses which 
would occur when the antenna is erected over 
imperfectly conducting ground. Second, by using 
the ground screen, the height of the antenna above 
ground is set accurately. As a result, the radiation 
resistance of the antenna is known and the radia- 
tion patterns of the antenna can be predicted more 
accurately. 

d. For the ground screen to have any effect on 
the very low-angle radiation produced by the 
antenna, it would have to be unreasonably large, 
since ground reflections that effect such radiation 
take place at a considerable distance from the 
antenna. When a ground screen is used which 
extends for only a half-wavelength in every direc- 
tion from the antenna, only the high-angle radia- 
tion is affected. The ground reflections that con- 
tribute to such radiation take place rather close 
to the antenna itself. Therefore, the ground screen 
is effective in such cases. 

62. Polarization 

a. General. 

(1) Polarization of a radiated wave is de- 
termined by the direction of the lines of 
force making up the electric field. If the 
lines of electric force are at right angles 
to the surface of the earth, the wave is 
said to be vertically polarized. If the 
lines of electric force are parallel to the 
surface of the earth, the wave is said to 
be horizontally polarized. 

(2) When a single-wire antenna is used to 
extract energy from a passing radio wave, 
maximum pickup results if the antenna 
is so oriented that it lies in the same 
direction as the electric-field component. 
Thus, a vertical antenna is used for effi- 
cient reception of vertically polarized 
waves and a horizontal antenna is used 
for the reception of horizontally polarized 
waves. In some cases, the orientation of 
the electric field does not remain constant. 
Instead, the field rotates as the wave 
travels through space. Under these con- 
ditions, both horizontal and vertical 
components of the field exist and the 
wave is said to have elliptical polariza- 
tion. 



107 



(3) When a half-wave antenna is used for the 
radiation of energy, the electric lines of 
force are built up from one end of the an- 
tenna to the other. At a distance from the 
antenna, the lines of force have a direc- 
tion that is parallel to the direction of the 
radiating antenna. Therefore, the hori- 
zontal half-wave antenna produces a 
horizontally polarized radio wave and the 
vertical half-wave antenna produces a 
vertically polarized radio wave. 
b. Polarization Requirements for Various Fre- 
quencies. 

(1) At medium and low frequencies, ground- 
wave transmission is used widely, and it 
is necessary to use vertical polarization. 
The lines of electric force are perpen- 
dicular to the ground and the radio wave 
can travel a considerable distance along 
the ground surface with a minimum 
amount of attenuation. Horizontal po- 
larization cannot be used since, under 
these conditions, the electric lines that 
touch the earth do so parallel to it. 
Because the earth acts as a fairly good 
conductor at these low frequencies, the 
horizontal lines of electric force are 
shorted out, and little useful range can 
be covered with horizontal polarization. 

(2) At high frequencies, with sky-wave trans- 
mission, it makes little difference whether 
horizontal or vertical polarization is used. 
The sky wave that has been reflected by 
the ionosphere, arrives at the receiving 
antenna elliptic-ally polarized. This is 
the result of the effect of the earth's 
magnetic field on a wave traveling ob- 
liquely through it and striking the 
ionosphere. The radio wave is given a 
twisting motion and its orientation 
changes continually because of the un- 
stable nature of the ionosphere. The 
relative amplitudes and phase difference 
between the horizontal and vertical 
components of the received wave change 
at random, and the transmitting and 
receiving antennas therefore can be, 
mounted either horizontally or vertically. 

(3) One reason for the preference for hori- 
zontally polarized antennas in the high- 
frequency range is that less interference 
is experienced because of man-made 
noise source. Vehicular ignition systems, 



various rotating machinery, and many 
electrical appliances produce vertically 
polarized interference signals. The re- 
sponse to these signals is minimized by 
using horizontal polarization. Supports 
for horizontally polarized antennas are of 
more convenient size. There also is less 
absorption of radiated energy by building 
or wiring when horizontal polarization is 
used, 

(4) With frequencies in the very-high or ultra- 
high range, either horizontal or vertical 
polarization is satisfactory. The radio 
wave travels directly from transmitting 
antenna to receiving antenna and the 
ionosphere is not used. The original 
polarization produced at the transmitting 
antenna is maintained throughout the 
entire travel of the wave to the receiver. 
Therefore, if a horizontal half-wave an- 
tenna is used for transmitting, a hori- 
zontal antenna must be used for receiv- 
ing. If a vertical half-wave antenna is 
used for transmitting, a vertical antenna 
must be used for receiving. 

(5) In some cases, the orientation of the 
receiving antenna need not be the same 
as the transmitting antenna for vhf and 
uhf signals. For example, when duct 
transmission occurs, as described in 
chapter 2, the orientation of the wave 
may -change as the energy travels to the 
receiving antenna in much the same way 
as when a high-frequency sky wave is 
reflected from the ionosphere. Similar 
antenna orientations are not required 
when a large amount of the received 
energy arrives at the receiving antenna 
through diffraction or by reflection from 
irregular surfaces or from large flat 
oblique surfaces for example, when the 
receiving antenna is located in an urban 
area near large buildings. Since duct 
transmission is an abnormal condition 
and most military vhf and uhf antennas 
are located in the clear away from large 
reflecting surfaces, the same antenna 
orientation generally is used for the 
receiving antenna as for the transmitting 
antenna. 

c. Advantages of Vertical Polarization. 

(1) Simple, vertical, half-wave antennas can 
be used to provide omnidirectional (in 



108 



all directions) communication. This is 
advantageous when it is desired to com- 
municate with a moving vehicle. 

(2) When antenna heights are limited to 10 
feet or less over land, as in a vehicular 
installation, vertical polarization results 
in a stronger received signal at frequen- 
cies up to about 50 mc. From approxi- 
mately 50 to 100 mc, there is only a slight 
improvement over horizontal polariza- 
tion with antennas at the same height. 
Above 100 mc, the difference in signal 
strength is negligible. One polarization 
may produce a greater or smaller signal 
strength, depending on local conditions. 

(3) For transmission over sea water, vertical 
polarization is decidedly better than 
horizontal when antennas are below 
approximately 300 feet at 30 mc, but 
only 50 feet at 85 mc, and still lower at 
the higher frequencies. Therefore, at 
ordinary antenna mast heights of 40 feet, 
vertical polarization is advantageous for 
frequencies less than about 100 mc. 

(4) Radiation using vertical polarization is 
somewhat less affected by reflections 
from aircraft flying over the transmission 
path. With horizontal polarization, such 
reflections cause variations in received 
signal strength. This factor is important 
in locations where aircraft traffic is heavy. 

(5) When vertical polarization is used, less 
interference is produced or picked up 
because of strong vhf and uhf broadcast 
transmission and reception (television 
and f-m (frequency-modulation)), all of 
which use horizontal polarization. This 
factor is important when an antenna 
must be located in an urban area having 
several television and f-m broadcast 
stations. 

d. Advantages of Horizontal Polarization. 

(1) A simple horizontal half-wave antenna 
is bidirectional. This characteristic is 
useful if it is desired to minimize inter- 
ference from certain directions. 

(2) Horizontal antennas are less apt to pick 
up man-made interference, which ordin- 
arily is polarized vertically. 

(3) When antennas are located near dense 
forests, horizontally polarized waves suf- 
fer lower losses than vertically polarized 
waves, especially above about 100 mc. 



(4) Small changes in antenna location do not 
cause large variations in the field intensity 
of horizontally polarized waves when 
antennas are located among trees or 
buildings. When vertical polarization 
is used, a change of only a few feet in 
the antenna location may have a con- 
siderable effect on the received signal 
strength. This is the result of interfer- 
ence patterns which produce standing 
waves in space when spurious reflections 
from trees or buildings occur. 

(5) Since the interference patterns will vary 
even when the frequency is changed by 
only a small amount, considerable dis- 
tortion may occur when complex types 
of modulation are used, as with tele- 
vision signals or with certain types of 
pulse-modulation systems. Under these 
conditions, horizontal polarization is pre- 
ferred. 

(6) When simple half-wave antennas are 
used, the transmission line, usually ver- 
tical, is less affected by a horizontally 
mounted antenna. By keeping the an- 
tenna at right angles to the transmission 
line and using horizontal polarization, 
the line is kept out of the direct field of 
the antenna. As a result, the radiation 
pattern and electrical characteristics of 
the antenna are practically unaffected by 
the presence of the vertical transmission 
line. 

63. Reciprocity 

a. The half- wave antenna has the property of 
reciprocity since, when it is used for transmitting, 
it has the same characteristics as when used for 
receiving. 

6. The function of a transmitting antenna is 
to convert the output power delivered by a radio 
transmitter into an electromagnetic field that is 
radiated through space. As such, the transmit- 
ting antenna is a transducer which converts energy 
having one form to energy having another form. 
The receiving antenna is also a transducer; how- 
ever, it makes the energy conversion in the 
opposite direction. The function of the receiving 
antenna is to convert the electromagnetic field 
that sweeps by it into energy that is delivered to 
a radio receiver. In transmission, the antenna 
operates as the load for the transmitter; in recep- 



109 



tion, it operates as the power source for the 
receiver, which is the load. 

c. A half-wave antenna used for transmitting 
radiates its maximum energy at right angles to the 
antenna itself and no energy is radiated in the 
direction of the antenna elements. When such an 
antenna is used for reception, it receives maximum 
energy at right angles to the antenna and no energy 
is received in the direction of the antenna elements. 
A vertically mounted half-wave antenna radiates 
energy equally in all horizontal directions. A 
similar antenna used for reception receives energy 
equally in all horizontal directions. Consequently, 
a pattern which shows the radiation of an antenna 
can be used also to show the reception of that an- 
tenna. A radial line drawn at a given angle from 
the center of such a pattern to its edge not only can 
be used to indicate the magnitude of the radiated 
field traveling outward on the pattern from the an- 
tenna but also shows the magnitude of the received 
field traveling inward on the pattern toward the 
antenna. 

d. This reciprocity of radiation and reception 
applies not only to the half-wave antenna but also 
to more complicated antennas and arrays de- 
scribed later in this manual. In all cases, if an 
antenna is highly directive as a transmitting 
antenna, it will have exactly the same directivity 
as a receiving antenna. 

e. The gain of an antenna is the same regardless 
of whether the antenna is used for transmitting 
or for receiving (par. 82). The impedance and the 
distribution of standing waves of voltage and 
current are identical regardless of whether the 
half-wave antenna is used for transmitting or for 
receiving. 

/. If the reciprocity of an antenna is to be 
realized fully, it is necessary that the nature of the 
wave remain unchanged as it travels from trans- 
mitting antenna to receiving antenna. If it does 
not, two identical antennas similarly oriented may 
not act as though their characteristics were truly 
reciprocal. For example, assume that two such 
antennas, A and B, are used for communication by 
way of the ionosphere. When antenna A is trans- 
mitting to antenna B, the radiated energy follows 
a certain path up toward the ionosphere and then 
down toward antenna B. When antenna B is 
transmitting to antenna A, it is possible that a 
slightly different path is followed by the radiated 
energy in travelling toward antenna A. Under 
these conditions, the received wave at A will arrive 



at a slightly different vertical angle, and a different 
part of the directive pattern will be used. 

g. For complete reciprocity to exist, it is 
necessary that antennas be terminated similarly 
when transmitting and receiving. If proper 
impedance matching is used when an antenna is 
transmitting, a proper match must exist also when 
the antenna is used for receiving. If it does not, 
the antenna will have somewhat different char- 
acteristics when it is used for transmission than 
when it is used for reception. 

64. Single-wire Antenna 

a. General. 

(1) A single-wire, half-wave antenna is one 
constructed of a single conductor of 
proper length. In the high-frequency 
range, the conductor is usually a stranded 
copper-alloy wire which is suspended 
between two upright supports. In the 
vhf and uhf frequency ranges, aluminum 
tubing frequently is used, and the 
antenna length is sufficiently short that 
the tubing need be supported only at the 
center. 

(2) The single-wire antenna can be mounted 
either vertically to produce a vertically 
polarized radio wave, or horizontally, to 
produce a horizontally polarized wave. 
Any of the feeding methods previously 
described can be used, depending on the 
type of transmission line or the nature of 
the output circuit of the transmitter that 
is to be connected to the antenna. Most 
of the figures so far in this manual show 
single-wire, half-wave antennas. Details 
of erection and construction have been 
omitted since only the theory of the 
antenna operation was under discussion. 

b. Typical Military Antennas. 

(1) The typical military half-wave antenna 
in figure 99 is suitable for transmission 
and reception. It can be used in con- 
junction with a transmitter having an 
output power of less than 100 watts. 
All of the component parts required for 
the installation are furnished in kit form. 
When the antenna is disassembled, it is 
highly portable. 

(2) Sufficient antenna wire is provided to 
construct a half-wave antenna resonant 



no 




Figure 99. Typical military single-wire antenna. 



to a frequency as low as 1.5 mc, that is 
312 feet long. If the antenna is to 
operate on a frequency as high as 18 mc, 
the length is shortened to 24.3 feet. In 
actual practice, antenna wire forming 
one-half of the antenna is wound on a 
small reel. The wire is unwound from 
the reel until one-half the required 
antenna length is obtained. Then the 
free end of the wire is attached to an 
insulator at the center of the antenna. 
The reel then is attached to a halyard 
which has a strain insulator in it. The 



halyard is passed through a pulley at the 
top of one of the antenna supports with 
the free end made fast to a guy stake on 
the ground. Both halves of the antenna 
are made up in this way. The insulator 
at the center of the antenna is fitted 
with a female coaxial fitting. This 
accommodates a 72-ohm coaxial line 
which leads to either the transmitter or 
the receiver, providing the correct im- 
pedance for matching to the center of 
the half -wave antenna. 
(3) A single-wire transmission line also can 



111 



-10' 





10- 



12"— 




-WIRE SIZE,**6B 8 S 
APPROX 

TO STATION 




TM 666-104 



Figure 100. Delta-matched military antenr, 



be used with this antenna. Here, the (6) 

center insulator is not used and the two 

portions of the antenna are connected. 

The single-wire line is connected at a 

point 0.18-wave length from one end of 

the antenna, giving it a proper impedance 

match. 

(4) Another common high-frequency, single- 
wire antenna is illustrated in figure 100. 
This delta-matched military antenna 
often is used with transmitters having 
output powers up to several kilowatts (1) 
over distances of 500 miles or less for 
point-to-point, fixed-station communi- 
cation. For transmission up to about 

200 miles, the antenna height should be 
less than a quarter-wavelength in order to 
produce the required high angle of radi- 
ation. For longer distances, the antenna 
height may be as great as a half-wave- 
length to produce lower radiation angles. 

(5) The antenna consists of a single, hori- 
zontal wire of about one half-wavelength. 
The power from the station transmitter is 
transferred from a balanced 600-ohm 
transmission line to the antenna by means 

of a delta-matching section. The ends of (2) 

the wires of the transmission line are 

fanned out and attached to the antenna 

at equal distances from the center. The 

dimensions must be such as to match the 

line to the antenna so that the standing 

waves on the line are minimized. The 

exact dimensions depend on the antenna 

height, ground conditions, frequency, 

and the effect of structures near the 

antenna. 



A delta-matched, half-wave antenna used 
at a frequency of 5 mc would have the 
following dimensions (fig. 100): height, 
H, 70 feet, antenna length, A, 94 feet, 
maximum spread of matching section, B, 
23.6 feet, and length of matching section, 
C, 29.5 feet. 



65. Folded Dipole 

a. Operation. 



The folded-dipole antenna consists of an 
ordinary half-wave antenna (dipole) 
which has one or more additional con- 
ductors connected across the ends of the 
antenna. The additional conductors are 
mounted parallel to the dipole elements 
at a distance that is a very small fraction 
of a wavelength in which spacings of 
several inches are common (fig. 101). In 
A. the two-wire folded dipole is con- 
structed of metal tubing. In B, the 
three-wire folded dipole is made of wire. 
The electrical length of both antennas is 
a half-wave, which is about 95 percent 
of the free-space half-wavelength. 
Consider first the simple, two-wire folded 
dipole. Assume that the additional con- 
ductor is removed at points 1 and 2. 
Assume further that the charge remain- 
ing on the simple half-wave antenna is 
such that the end of the antenna at point 
1 is maximum positive and the end of the 
antenna at point 2 is maximum negative. 
Ordinarily, current then would start to 
flow from point 2 toward point 1. Now, 
if the additional conductor is connected 



112 




as shown in the figure, this current finds 
two paths available. Consequently, the 
current divides so that about half flows 
from right to left in the additional con- 
ductor, and the remaining half flows in 
the same direction in the lower conduc- 
tor making up the simple half-wave 
antenna. This occurs with no change in 
input power. 

(3) Since impedance varies inversely as the 
square of the current (Z—P/P), a reduc- 
tion in the current flowing in that branch 
of the folded dipole to which the trans- 
mission line is connected results in an in- 
crease in the input impedance of the 
antenna. As the current is reduced to 
one-half its original value, the impedance 
of the antenna increases to four times 73 
ohms, or close to 300 ohms. Therefore, 
a 300-ohm transmission line can be con- 
nected to the folded dipole, and a correct 
impedance match occurs. 

(4) If three conductors are used instead of 
two, a given input power will produce 
only one- third the original current in 
each conductor. As a result, the input 
impedance of the antenna rises to nine 
times 73 ohms, or about 600 ohms. This 
provides the proper impedance match 
for an ordinary 600-ohm transmission 
line, and the folded dipole antenna pro- 
vides an impedance step-up that can be 
used to affect an impedance match to 
common transmission lines. 

b. Effect of Different Conductor Sizes. 

(1) The folded dipoles discussed previously 
were constructed of conductors of equal 
diameter. If the added conductor has a 



larger diameter than the conductor to 
which the transmission line is connected, 
the impedance step-up produced by the 
folded dipole is increased. Under these 
conditions, the spacing between the two 
conductors has considerable effect on the 
impedance step-up. 
(2) Assume that a folded dipole (fig. 102) is 
constructed so that the diameter of con- 
ductor No. 1 is twice the diameter of 
conductor No. 2. The effect of various 
spacings (S) is shown in the following 
chart, which gives the input impedance 
of this particular folded dipole arrange- 
ment: 



Ratio of spacing (S) 


Input im- 


Ratio of spacing (S) 


Input im- 


to diameter of con- 


pedance 


to diameter of con- 


pedance 


ductor No. 2 


(ohms) 


ductor No. 2 


(ohms) 


2:1 


730 
440 
365 


12:1 


360 
330 


4:1 


25:1 


8:1 





c. Radiation Pattern and Frequency Response. 

(1) The radiation pattern produced by the 
folded dipole is similar to that produced 
by the conventional half-wave antenna. 
Maximum radiation occurs at right 
angles to the antenna itself and mini- 
mum radiation occurs off the antenna 
ends. 

(2) An advantage of the folded dipole is that 
its characteristics do not change with 
frequency as much as do those of an 
ordinary dipole. Therefore, the folded 
dipole has a broader frequency response. 
This increased bandwidth is the result 



113 



CONDUCTOR I 




TRANSMISSION 

LINE TO 
TRANSMITTER 



TM 666-106 

Figure 102. Folded dtpole constructed of different conductor 

.SiZC.S. 

of the greater cross-sectional area of the 
antenna which results from the folding. 
As a result, the folded dipole has greater 
capacitance and a smaller inductance per 
unit of length than does the ordinary 
dipole. The reduction in the L-C (in- 
ductance-capacitance) ratio of the an- 
tenna results in a lowered Q and a reduced 
f req ue ncy sel ec tivit y . 
(3) The greater bandwith of the folded dipole 
is explained by the fact that it acts not 
only as an antenna but as two short- 
circuited quarter-wave transmission lines 
connected end to end (at the antenna 
center). Since the reactance at the center 
of an antenna varies in the opposite 
direction to the reactance at the end of a 
quarter-wave section of transmission line, 
the result is a reduction in the rate of 
reactance change which decreases the Q 
of the antenna and increases the fre- 
quency range. 

66. Coaxial Antenna 

a. Description and O peration . 

(1 ) The coaxial or sleeve antenna is a common 
vertical radiator that is used in the vhf 
and uhf bands In figure 103, the typical 
military coaxial antenna consists of a 
vertical half-wave antenna so constructed 
as to provide a convenient, mechanical 
feed arrangement. The antenna is fed by 
means of flexible coaxial cable which runs 
up through the supporting staff. 

(2) The inner conductor of the coaxial cable 
is connected to the upper portion of the 

114 



antenna, designated as the whip. The 
outer conductor of the cable is connected 
through a shorting ring to the top of an 
outer skirt. This skirt or sleeve is a 
hollow metal cylinder that is mounted 
around the outside of the mounting staff 
which supports the antenna. A small air 
space exists between the skirt and the 
outer surface of the mounting staff except 
at the location of the shorting ring. The 
skirt then acts as the lower portion of 
the antenna. 
(3) The skirt has an additional function. In 
conjunction with the outer surface of the 
metal mounting staff, it forms a quarter- 
wave section of transmission line which is 
short-circuited at one end by the shorting 
ring. The impedance at the bottom end 
of the line so formed is very high. As a 
result, current flow is minimized on the 
mounting staff and on the outer con- 
ductor of the coaxial cable. Such current, 
if allowed to flow, would produce radi- 
ation at a high vertical angle. By reduc- 
ing this current to a minimum value, 
the radiation is reduced. In this way, the 
low-angle, line-of-sight transmission re- 
quired in the vhf and uhf bands is 
produced. 
b. Dimensions. 

(1) The dimensions of the whip and especially 

of the skirt are highly critical. The upper 
radiating portion (whip) {A, fig. 103) is 
made 95 percent of a free-space quarter- 
wavelength. The length of the lower 
radiating portion (skirt) (B) is made 
equal to a free-space quarter-wavelength. 
Actually, the skirt should be somewhat 
shorter than this value to produce maxi- 
mum efficiency as a radiator. Its length, 
however, is chosen for best operation 
as a quarter-wave line, which produces 
slightly higher radiation efficiency. Ad- 
justing clamps are provided both for the 
skirt and for the whip so that the antenna 
may be adjusted for any frequency in a 
given bank. 

(2) One particular military coaxial antenna 
has a frequency range of from 30 to 40 
mc. Another military coaxial antenna 
has a frequency range of 70 to 100 mc. 
Markings sometimes are provided on the 
elements themselves to show the correct 



ADJUST.' 9LE PORTION 
OF WHIP 




PROTECTING SPRING 

COAXIAL CABLE 

Figure 103. Military coaxial antenna. 



TM 666-107 



whip and sleeve lengths for various reso- 
nant frequencies. For example, a coaxial 
antenna suitable for 35 mc would have a 
whip length of 80 inches and a skirt 
length of 84 inches. 
c. Characteristics. The horizontal radiation 
pattern produced by the coaxial antenna is circular 
and, therefore, provides omnidirectional radiation 
or reception. Maximum radiation occurs at right 
angles to the antenna itself. The input impedance 
of a coaxial antenna is about 50 ohms, and a proper 
impedance match is produced when 50-ohm 
coaxial line is used. 



67. Conical Antenna 

a. Description and Characteristics. 

(1) The conical antenna is one of a large num- 
ber of special antennas that have been 
developed to operate satisfactorily over a 
wide frequency band. One type of coni- 
cal antenna, constructed of two solid 
metal cones, is shown in figure 104. Fre- 
quently the conical antenna is con- 
structed of metal mesh or is simply a 
framework of metal rods that forms the 
required shape. The cones are arranged 



115 




TRANSMISSION 

^ TM 666-108 

Figure 101,. Simple conical antenna, 

on a common horizontal or vertical axis, 
depending on whether horizontal or 
vertical polarization is required. 
(2) If the conical antenna is to operate as a 
half-wave antenna, its over-all length 
must be considerably shorter than a 
free-space half-wavelength. This is the 
result of the large end effect produced by 
the bases of the cones forming the an- 
tenna. As the apex angle, A, is increased, 
the length of antenna required is reduced. 
For example, if angle A is 10°, the over- 
all length required is about 75 percent of 
a free-space half-wavelength. With angle 
A at 20°, the length is only about 70 per- 



cent of a free-space ha]f-w r avelength. 
When such short leugths are used, the 
input impedance is approximately 40 
ohms. When the conical antenna is 
operated as a full-wave antenna, the over- 
all length commonly is made 73 percent 
of a free-space wavelength and the input 
impedance is several hundred ohms. 
When such an antenna has an apex angle 
of 10°, the input impedance is 950 ohms; 
with an apex angle of 20°, the input 
impedance is 600 ohms; with an apex 
angle of 30°, the input impedance is 300 
ohms. The value may be reduced by 
using large apex angles, in excess of 30°. 
(3) The large, cross-sectional area of the 
conical antenna accounts for its wide 
frequency response. Like the folded 
dipole, the conical antenna has a large 
capacitance but a small inductance per 
unit length. This reduces the effective Q 
of the antenna and causes its characteris- 
tics to change more slowly as the fre- 
quency is varied away from resonance. 
As the apex angle, A, increases, the band- 
width of the conical antenna increases. 
The radiation pattern of the conical 
antenna is similar to that produced by an 




TM 666-109 



Figure 105. Typical military conical antenna assembly. 



116 



ordinary half-wave antenna which is 
similarly oriented. 

b. Typical Military Antenna. 

(1) The typical military conical antenna as- 
sembly shown in figure 105 consists of 
two modified conical antennas mounted 
at opposite ends of a cross arm atop a 
mast. The antennas can be mounted 
either as shown for the reception of verti- 
cally polarized signals, or they can be 
rotated through 90° for the reception of 
horizontally polarized signals. Each of 
these antennas has its own transmission 
line. These lines lead to separate input 
circuits of a vhf, uhf receiver. 

(2) One antenna has an over-all length of 
slightly more than 23 inches, with a maxi- 
mum cone diameter of 8% inches. The 
other antenna has a length of about 16% 
inches with a cone diameter of 6% inches. 
The larger antenna operates over a fre- 
quency range from 145 to 300 mc, the 
smaller antenna over a frequency range 
from 300 to 600 mc. Because of the 
large apex angle (about 60°), the varia- 
tion in the input impedance of the an- 
tenna is very small over an extremely 
wide frequency range, and the antenna 
length is not particularly critical. 

(3) The dimensions of the conical antenna 
were computed to maintain an imped- 
ance centered at about 95 ohms over the 
frequency range for which they were de- 
signed. Two shielded, two-wire lines 
having a characteristic impedance of 95 
ohms are used to connect the antennas to 
the two input circuits of the receiver. 

68. Microwave Antenna 

a. The half-wave antenna can be used at any 
operating frequency. When microwave frequen- 
cies (at the upper part of the uhf band and higher) 
are used, the antenna length required is extremely 
small. For example, the microwave frequency of 
5,000 mc has a free-space, half-wavelength of only 
a little more than 1 inch. The length of the half- 
wave antenna at this frequency would have a 
value somewhat less than this small distance. 

b. The small size of a microwave, half-wave an- 
tenna is both a disadvantage and an advantage. 
The electrical characteristics of the microwave 



antenna are exactly like those of its larger, lower- 
frequency counterpart. It has about the same 
radiation pattern, the same distribution of stand- 
ing waves along its length, and the same radiation 
resistance for similar conditions. However, the 
amount of signal pick-up when such a small an- 
tenna is used for receiving is reduced greatly. Any 
receiving antenna is able to pick up energy from a 
section of an incoming wave front that extends less 
than a quarter-wavelength away from the antenna. 
Therefore, a receiving antenna that is, say, 50 to 
100 feet long is able to pick up a far greater amount 
of energy than can a microwave antenna only 
about an inch in length. So poor is the signal 
pick-up that a simple, half- wave antenna rarely 
is able to pick up enough microwave energy to 
overcome the noises generated within the receiver 
itself. As a result, a simple, half-wave antenna 
seldom is used alone in the microwave range. 

c. The great advantage of the small size of 
microwave, half-wave antennas is that it becomes 
convenient to use a large number of these together 
to form an array of antennas. All antenna arrays 
have two things in common: First, an array 
produces a concentration of radiated energy in 
certain directions; that is, the array is highly 
directional — it has gain. Second, an array occupies 
a greater space than does the single, half- wave 
antenna, since it is made up of a number of half- 
wave antennas, and the greater the number of 
individual antennas that make up the array, the 
greater are the directivity and gain. In the 
microwav.e range, the construction of very elab- 
orate arrays of half-wave antennas can be ac- 
complished in a reasonably small space. Some 
microwave arrays are made up of as many as 
32, 64, 100, or even 250 individual half-wave 
dipoles. 

d. Other microwave antennas are composed 
of a single half-wave dipole or an array that is 
used in conjunction with specially shaped re- 
flectors. These operate in much the same way 
as the parasitic reflectors (pars. 94 through 109) 
except that these reflectors are much larger than 
the dipole and they are specially shaped. 

e. One commonly used reflector, shown in A 
of figure 106, is the corner-reflector type. The 
reflector is composed of two flat, metal sheets 
which meet at an angle to form a corner. Wire 
mesh or metal tubing sometimes is used instead 
of the solid metal. The half -wave, microwave 
antenna is located so that it bisects the corner 
angle because maximum radiation occurs out of 



117 



MAXIMUM 
RADIATION 



FEEDERS 




COAXIAL OR 
WAVEGUIDE 
FEED 



MAXIMUM 
RADIATION 

/// 




DIPOLE 
ANTENNA 



DIPOLE- v/ I 
ANTENNA \l ^ 



HEMISPHERICAL 
REFLECTOR 



B 



SHEET REFLECTORS 



Figure 106. Half-wave antennas used with special reflertm 



the corner. The field strength produced by such 
an arrangement is considerably more than would 
be produced by the antenna alone. 

/. Another commonly used reflector, shown in 
B, is the paraboloidal type. The shape is 
similar to that of reflectors used in searchlights 
that concentrate energy from a light bulb into a 
narrow, well defined beam. The reflector is 
constructed of solid metal or metal mesh. A 
half-wave, microwave antenna is located at the 



focal point of the paraboloid. Energy arriving 
at any angle from the antenna is reflected by t*>e 
paraboloid in parallel rays. This results in a very 
narrow beam of radio energy in the direction 
shown. A small, metal, hemispherical reflector 
prevents direct radiation from the half-wave 
antenna from interfering with the beam produced 
by the paraboloidal reflector. The small re- 
flector causes all of the energy from the antenna 
to be directed back into the paraboloid. 



Section VI. GROUNDED ANTENNAS 



69. Quarter-wave Antenna 

a. The ground is a fairly good conductor for 
medium and low frequencies and acts as a large 
mirror for the radiated energy. This results in 
the ground reflecting a large amount of energy 
that is radiated downward from an antenna 
mounted over it. It is just as though a mirror 
image of the antenna is produced, the image 
being located the same distance below the surface 
of the ground as the actual antenna is located 
above it. Even in the high-frequency range and 
higher, many ground reflections occur, especially 
if the antenna is erected over highly conducting 
earth, salt water, or a ground screen. 

b. Utilizing this characteristic of the ground, 
an antenna only a quarter-wavelength long can 
be made into the equivalent of a half-wave 
antenna. If such an antenna is erected vertically 
and its lower end is connected electrically to the 
ground (fig. 107), the quarter-wave antenna 
behaves like a half-wave antenna. Here, the 



ground takes the place of the missing quarter- 
wavelength, and the reflections supply that part 
of the radiated energy that normally would be 
supplied by the lower half of an ungrounded 
half-wave antenna. 

c. When the charge on the quarter-wave 
grounded antenna is maximum positive at its upper 
end, the charge at the lower end of the image 
antenna is maximum negative. Current begins to 
flow toward the positive end of the quarter-wave 
antenna, as illustrated by the arrow in the figure. 
Current in the image antenna begins to flow away 
from the negative end of the image. Note that 
the current flow is up in both eases. This is 
similar to conditions in a vertical half-wave 
antenna that has negative polarity at the bottom 
and positive polarity at the top. It is just as 
though the half-wave antenna were driven halfway 
into the earth. 

d. From figure 107, it must not be assumed that 
if a hole were dug into the earth under the antenna 
to a depth of a quarter-wave, conditions of voltage, 



118 



-QUARTER-WAVE 
ANTENNA 



MIRROR 



REFLECTED RA* 



ir 
it 



IMAGE ANTENNA' 



tm «a«-in 

Figure 107. Quartet -wave antenna connected to ground. 

and current such as described above would be 
found. Actually, this is not true, since the fields 
produced by the grounded quarter-wave antenna 
terminate only a short distance below the ground 
surface. None of the radiated field from the 
antenna penetrates the earth to any great extent. 
The following assumption with reference to a 
lighted flashlight will help to clarify the concept. 

e. Assume that a soldier is standing before a 
mirror with a lighted flashlight in his hand (fig. 
108). He holds the flashlight in such a way that 
its light is reflected by the mirror into his eyes; 
that is, the head of the actual flashlight is pointed 
away from him, whereas the head of the image 
flashlight is pointed toward him. This is a 180° 
shift in position. The effect is the same as if a 
flashlight were located behind the mirror the same 
distance as the actual flashlight is located before 
the mirror, without the mirror in the way. The 
image flashlight in the mirror is shining directly 
into the eyes of the soldier, although it is not a 
physical object as drawn in the figure, and if he 
looked behind the mirror he would find no flash- 
light. If the mirror were removed, there would 
be no reflected ray and the effect would be - as 
though the image flashlight had disappeared. 

j. The idea of an image flashlight can be applied 
to an image antenna formed by the ground. No 
antenna actually is located deep in the ground, 
but, because of the reflection of energy, conditions 
are similar to those that would occur without the 
reflecting surface and with a source of energy 
located as shown by the dashed lines in figures 107 
and 108. Just as the position of the image flash- 
light is reversed, the polarity of charge on the 
image antenna is opposite to that of the actual 
antenna. 

g. At medium and low frequencies, ground-wave 




IMAGE FLASHLIGHT 



tm cee-112 

Figure 108. Formation of image flashlight. 

transmission is used, which requires vertical polar- 
ization, and vertical antennas are necessary. If 
the half-wave antenna were used, an extremely 
tall structure would be required. At 1,000 kc, 
for example, the length of a vertical half -wave 
antenna would be almost 500 feet. When oper- 
ated as a grounded type, the antenna need be only 
half that length, and loading devices make possible 
the use of even shorter lengths. These will be 
described below. Even in the high-frequency or 
very-high-frequency range, where the length of a 
half-wave antenna is much shorter, the use of the 
grounded quarter-wave antenna is common for 
portable and mobile installations. Here, resonant 
antennas of minimum length are required so that 
they can be carried easily by vehicles or by hand. 

h. The term Marconi antenna sometimes is used 
to designate the grounded quarter-wave antenna, 
but this term is being replaced gradually by more 
descriptive terms which relate to specific types of 
grounded quarter-wave antennas. The term 
Hertz antenna, occasionally used to designate the 
ungrounded half-wave antenna, also is being 
replaced by more descriptive terms to designate 
specific antenna types. 

70. Current and Voltage Distribution 

a. The distribution of the standing waves of 
voltage and current on a grounded quarter- wave 
antenna is the same as the distribution on one-half 
of a half-wave antenna. The voltage is maximum 
and the current is zero at the end of the antenna 
farther from ground. As a result, the impedance 
which is the ratio of voltage to current, is highest 
at this point. The voltage and current vary in 
amplitude sinusoidally along the antenna; at the 



119 



grounded end of the antenna, the voltage is zero 
and the current is at its maximum value. As a 
result, the impedance is lowest at this point. The 
distribution of the standing waves of voltage and 
current is shown in A of figure 109. 



VOLTAGE 




CURRENT 



A 




CURRENT LOADING COIL 



B 

"M 666-113 



Figure 109. Distribution of voltage and current on grounded 
(iiinrter- wa ve ante n n a . 



71. Polarization 

■'i. The polarization of the radiated field pro- 
duced by the grounded quarter-wave antenna is 
always vertical. Figure I 10 shows the electric 
lines of force existing around the antenna at a 
particular instant of time. 



ELECTRIC LINES- 




TM 666-114 

Figure 110. Eleelrie Jield around a grounded quarter-wave 
antenna. 



b. Frequently, loading coils are used at the base 
of the grounded quarter-wave antenna. The 
added inductance of the loading coil reduces the 
resonant frequency of the antenna. This causes 
the antenna to appear as though it were longer 
electrically than it is physically, and grounded 
antennas which are shorter than a quarter-wave- 
length may be made to operate as quarter-wave 
antennas. The distribution of voltage and current 
on such an antenna are as shown in B. The 
current is zero at the ungrounded end of the 
antenna and rises sinusoidally along the antenna 
until the position of the loading coil is reached. 
The current then remains relatively uniform 
throughout the length of the coil. The voltage is 
at its maximum value at the ungrounded end of 
the antenna, falls sinusoidally until the coil is 
reached, and then falls uniformly to zero at the 
ground connection. 

c. If a loading coii having a higher value of 
inductance is used, the length of the antenna can 
be reduced further. However, even though an 
extremely short antenna can be brought to 
resonance by a sufficiently large loading coil, the 
effective radiation of such an antenna is lowered 
and the ratio between the amount of power 
radiated and the amount of power dissipated in the 
resistance and ground connection of the coil is 
reduced considerably. 



b. In some grounded antennas, a portion of the 
antenna is mounted horizontally. Even here, 
when short antenna lengths are, used, a vertically 
polarized radio wave is produced, since the electric 
field is built up between the antenna and the 
ground rather than between one horizontal portion 
of the antenna and another horizontal portion. 

72. Radiation Characteristics 

a. General. 

(1) The radiation pattern produced by a 
grounded quarter-wave antenna is shown 
in figure 111. It resembles the pattern 
of a vertieal half-wave antenna in free 
space except that the latter pattern is cut 
in two horizontally. Maximum radia- 
tion (or reception) of energy occurs at 
right angles to the antenna and along the 
surface of the ground. The radiation 
falls off as the vertical angle is increased, 
until directly over the antenna (at a 
vertical angle of 90°), no radiation of 
energy occurs. A true semicircle is 
shown so that a comparison can be made 
with the radiation pattern. 

(2) A top view of the radiation pattern shows 
that this pattern is circular. The an- 
tenna, therefore, is omnidirectional in the 



120 




SIDE VIEW 



Figure 111. Radiation pattern produced 

horizontal plane, and radiates equally 
in all horizontal directions. The solid 
radiation pattern of the grounded 
quarter-wave antenna also is shown. 
Other Radiation Patterns. 

(1) Although most grounded vertical an- 
tennas are a quarter-wave length, it is 
common to find grounded antennas used 
with loading coils so that their length can 
be made shorter than a quarter-wave. 
The radiation patterns produced by such 
antennas are similar to the pattern 
shown in the previous illustration except 
that the amount of radiation is reduced 
and the side view of the pattern is 
practically a true semicircle. 

(2) Consider next the vertical-plane radiation 
patterns of vertical antennas which are 
longer than a quarter-wave length. As 
the antenna length is increased to a half- 
wavelength, the amount of radiation is 
increased at very low vertical angles and 
the pattern becomes flatter (A of fig. 
1 12) . As the antenna length is increased 
still more toward 5 eighth-wavelengths, 
even greater radiation occurs at very 
low vertical angles. However, small 
minor lobes begin to appear at vertical 
angles of about 60°, as shown in B. 

(3) If the antenna length is increased still 



SOLID PATTERN 



TM 666-115 

by a grounded quarter-wave antenna. 

more, the intensity of radiation at higher 
vertical angles is increased, as shown in 
C and D. Such antennas are not suit- 
able for ground-wave transmission or for 
transmission of large amounts of energy 
in the horizontal direction. 
c. Radiation Resistance. 

(1) The radiation resistance of a grounded 
quarter-wave antenna is just half that of 
the ungrounded half-wave antenna. For 
very thin antennas, the value of the 
radiation resistance is 36 ohms. If 
large-diameter tubing or wide towers are 
used, the value is reduced. 

(2) When grounded vertical antennas are 
used that are shorter than a quarter- 
wavelength, the radiation resistance is 
reduced still more. This can be seen in 
the following chart, where the value of 
radiation resistance is shown for vertical 
grounded antennas of various lengths. 



Antenna lengths 
(wavelengths) 


Radiation 
resistance 
(ohms) 


Antenna lengths 
(wavelengths) 


Radiation 
resistance 
(ohms) 


0.30 


60 
36 
20 


0.15 


8 
2 
1 


0.25 


0.10 


0.20 


0.05 







(3) When very short antennas are used, it is 



121 



ANTENNA LENGTH 

X 
2 




c 

Figure 112. Vertical-plane radiation patterns prod 

possible that the ohmic resistance of the 
antenna is considerably greater than the 
radiation resistance. For example, the 
ohmic resistance of the loading coil used 
might be several ohms, the resistance of 
the ground connection might be several 
ohms, and the resistance of the antenna 
itself might be 1 or 2 ohms, or more. 
The power that is dissipated in all these 
resistances may be considerably greater 
than the power that is radiated into 
space. If an antenna length used is .1 
wavelength the radiation resistance is 
only 2 ohms, and the total ohmic re- 
sistance is 8 ohms. 

(4) The power that is applied to this antenna 
is applied effectively to two resistances, 
one having a value of 2 ohms and the 
other having a value of 8 ohms. The 
input power is applied to a total re- 
sistance of 10 ohms. Since the power 
dissipated or radiated is directly propor- 
tional to the resistance, eight-tenths of 
the total power applied is used to over- 
come the antenna losses and only two- 
tenths of the total power applied pro- 
duces useful radiation. Therefore, it 
can be seen how important it is to keep 
the ohmic resistance of the antenna as 
low as possible, to use a good ground 
connection, and to use a well designed 
loading coil. 

(5) Once all the resistances that dissipate 



ANTENNA LENGTH 
8 




B 



ANTENNA LENGTH 




TM 666 116 

■erf by grounded vertical antennas of various lengths. 

power uselessly have been reduced to an 
absolute minimum, the amount of energy 
radiated can be increased by increasing 
the radiation resistance of the antenna. 
One method of accomplishing this is to 
increase the length of the antenna toward 
a half-wavelength. At this length, the 
radiation resistance is about 100 ohms. 
The current, maximum is no longer at the 
grounded end of the antenna but is 
instead halfway up the antenna. As a 
result, more energy is concentrated along 
the ground level and less energy is radi- 
ated at high vert ical angles (A of fig. 106). 
In addition, since a greater portion of the 
antenna is carrying high current, a 
greater radiated field strength is pro- 
duced. 

73. Types of Grounds 

a. General. 

(1) When grounded antennas are used, it is 
especially important that the ground 
have as high a conductivity as possible. 
This is necessary to reduce ground losses 
and to provide the best possible reflecting 
surface for the down-going radiated 
energy from the antenna. Since at low 
and medium frequencies the ground acts 
as a sufficiently good conductor, the 
problem is how to make connection to the 
ground in such a way as to introduce the 



122 



least possible amount of resistance in the 
ground connection. At higher frequen- 
cies, artificial grounds constructed of 
large metal surfaces are common. 

(2) The ground connection takes many forms, 
depending on the type of installation and 
the loss that can be tolerated. For fixed 
station installations, very elaborate 
ground systems are used. These fre- 
quently are arranged over very large 
areas so that they operate as part of the 
reflecting surface in addition to making 
the connection to ground itself. In 
many simple field installations, the 
ground connection is made by means of 
one or more metal rods driven into the 
earth. Where more satisfactory arrange- 
ments cannot be made, it may be possible 
to make ground connections to existing 
devices which are themselves grounded. 
Metal structures or underground pipe 
systems (such as water pipes) commonly 
are used as ground connections. In an 
emergency, a ground connection can be 
made by plunging one or more bayonets 
into the earth. 

(3) Sometimes, when an antenna must be 
erected over soil having a very low 
conductivity, it is advisable to treat the 
soil directly to reduce its resistance. 
Occasionally, the soil is mixed with a 
quantity of coal dust for this purpose or 
it can be treated with substances which 
are highly conductive when in solution. 
Some of these substances, listed in order 
of preference, are sodium chloride (com- 
mon salt), calcium chloride, copper 
sulphate (blue vitriol), magnesium sul- 
phate (Epsom salt), and potassium 
nitrate (saltpeter). The amount required 
depends on the type of soil and its 
moisture content. When these sub- 
stances are used, it is important that 
they do not get into nearby drinking 
water supplies. 

b. Radial Grounds. 

(1) The most common ground system used 
with vertical grounded antennas at fixed 
stations is the radial ground. This 
consists of a number of bare conductors 
arranged radially and connected. The 
conductors, which may be from a tenth- 
to a half-wavelength or more, are buried 



a short distance beneath the surface of 
the earth. Sometimes one radial ground 
system serves two vertical antennas 
which operate over different frequency 
ranges (fig. 113). 

(2) Both antennas are constructed of several 
conductors connected to form a cage. 
Antenna A is used over a frequency range 
of 7 to 12 mc; antenna B is used over a 
frequency range of 2.5 to 7 mc. Each 
antenna is connected to the transmitter 
by means of a feed-through insulator 
(mounted on the wall of the transmitter 
house). Horn gaps, mounted on the in- 
sulators and connected to ground rods, 
provide lightning protection for the an- 
tennas and transmitter. The guys and 
horizontal supporting wire are broken 
up by insulators to prevent resonance ef- 
fects which would cause absorption of 
power from the antennas. All of the 
radials forming the ground system are 
bonded together at the center and con- 
nected to the transmitter ground. 

(3) A common military ground system kit 
which is supplied for use with grounded 
vertical radiators consists of 36 radials 
made of #12 Copperweld wire. These 
radials, spaced every 10°, extend out- 
ward for a distance of at least 350 feet 
from a common terminal near the lower 
end of the antenna. The conductors are 
buried in trenches, 6 to 8 inches deep. 
Soldered to the free end of each radial 
conductor a 6-foot ground rod is driven 
into the earth. 

6. Ground Rods. 

(1) With a less elaborate ground system, a 
number of ground rods can be used. 
These rods usually are made of galvanized 
iron, steel, or copperplated steel in 
lengths up to 8 feet. One end of the 
rod is pointed so that it can be driven 
easily into the earth. The other end 
frequently is fitted with some type of 
clamp so that the ground lead can be at- 
tached. Some ground rods are supplied 
with a length of ground lead already 
attached. 

(2) A fairly good ground connection can be 
made by using several ground rods, 6 to 
10 feet apart, connected in parallel. If 
possible, the rods should be located in a 



123 




^~ GROUND SYSTEM 
SEE SEPARATE VIEW 





124 



GROUND 

Figure 113. Giound syste 1 



SYSTEM 

•m for two vertical antennas. 



T M 666-117 



moist section of ground or in a depression 
which will collect moisture. Ground re- 
sistance can be reduced considerably by- 
treating the soil with any of the sub- 
stances previously mentioned. A trench 
about a foot deep is dug around each 
ground rod and filled with some common 
rock salt, Epsom salt, or any of the other 
materials mentioned. The trench then 
is flooded with water, after which it is 
covered with earth. To remain effec- 
tive, this treatment should be renewed 
every few years. 
(3) For simple installations, a single ground 
rod can be fabricated in the field from 
pipe or conduit. It is important that a 
low resistance connection be made be- 
tween the ground wire and the ground 
rod. The rod should be cleaned thor- 
oughly by scraping and sandpapering at 
the point where the connection is to be 
made, and a clean ground clamp installed. 
A ground wire then can be soldered or 
joined to the clamp. The joint should 
be covered with tape to prevent an in- 
crease in resistance caused by oxidation. 
d. Counterpoise. 

(1) When an actual ground connection cannot 
be used because of the high resistance of 
the soil or because a large buried ground 
system is not practicable, a counterpoise 
may replace the usual direct ground 
connection in which current actually 
flows to and from the antenna through 
the ground itself. The counterpoise con- 
sists of a structure made of wire erected 
a short distance off the ground and in- 
sulated from the ground. The size of 
the counterpoise should be at least equal 
to and preferably larger than the size of 
the antenna. 

(2) The counterpoise operates by virtue of 
its capacitance to ground. Because of 
this capacitance, the ground currents 
which flow normally and usually are col- 
lected by conduction now are collected 
in the form of charge and discharge cur- 
rents. The end of the antenna which 
normally is connected directly to ground 
now is connected to ground through the 
large capacitance formed by the counter- 
poise. If the counterpoise is not well in- 
sulated from ground, the effect is much 



the same as that of a leaky capacitor. 
Leakage currents flow between the coun- 
terpoise and ground so that a poorly in- 
sulated counterpoise introduces more 
losses than no counterpoise at all. 

(3) The shape and size of the counterpoise are 
not particularly critical. In some field 
antenna installations, a type of grounded 
antenna is used in which a large portion 
of the antenna is folded into a horizontal 
position. The counterpoise used with 
such an antenna has the same shape 
and approximate dimensions as does 
the antenna itself. This counterpoise is 
mounted directly under the antenna at a 
height of about 8 to 12 feet off the ground. 

(4) When the antenna is mounted vertically, 
the counterpoise is made to have any sim- 
ple geometrical pattern such as those 
shown in figure 114. Although perfect 




TM 666-118 

Figure 11 4- Wire counterpoises. 



symmetry is not necessary, the counter- 
poise should extend for equal distances at 
all angles from the antenna. The area 
covered by the counterpoise should be as 
great as possible, although very little is 
gained by extending the counterpoise 
more than a half-wavelength from the 
lower end of the antenna. The distance 
between parallel adjacent wires making 
up the counterpoise should be about 
equal to the height of the counterpoise 
above ground, because long conductors 
become resonant and absorb power from 
the antennas. To avoid this, the use of 
short jumpers between the conductors 
causes the counterpoise to behave only 
as a capacitance to ground. Smaller 

125 



separations should be used with small 
counterpoises. 
(5) In some vhf antenna installations on 
vehicles, the metal roof of the vehicle 
is used as a counterpoise for the antenna. 
Small counterpoises made of metal mesh 
sometimes are used with special vhf an- 
tennas that must be located a consider- 
able distance off the ground. This 
counterpoise provides an artificial ground 
which helps to produce the required 
radiation pattern. 

Bent Antenna 

Description and Operation. 

(1) A bent antenna is a grounded antenna 
so constructed that a portion of it is 
mounted horizontally. Such an antenna 
takes the form of an inverted L or T. 
In an inverted-L antenna, a fairly long 
horizontal portion, or flattop, is used, 
and the vertical downlead, which forms 
an important part of the radiating sys- 
tem, is connected to one end of the flat- 
top. The length of the antenna is 
measured f»om the far end of the flattop 
to the point at which the downlead is 
connected to the transmitter. In the 
T antenna, a horizontal portion, or flat- 
top, also is used. Here the downlead, 
which is a part of the radiating system, 
is connected to the center of the flattop. 
The over-all length of the T antenna is 
equal to the entire length of the down- 
lead plus one-half the length of the 
flattop. 

(2) The purpose of the bent antenna is to 
afford satisfactory operation when it is 
not convenient to erect tall vertical 
antennas. This is particularly necessary 
wlen operation at low frequencies is 
required. 

(3) When the length of the flattop of a bent 
antenna (the entire length of the in- 
verted L or one-half the entire length of 
the T) is 1 quarter-wavelength long, a 
full quarter-wave of current appears as 
a standing wave on the flattop. The 
current is minimum at the far end of the 
flattop and maximum at the point where 
the downlead is connected. Since the 
current maximum is no longer at the 



ground level (as it is in a grounded an- 
tenna of a quarter-wavelength or less), 
but is elevated above ground by the 
length of the vertical downlead, there 
are several advantages. First, the high- 
angle radiation is reduced and more 
energy is propagated along the surface 
of the earth. This is particularly im- 
portant when ground-wave transmission 
is used. Second, because more of the 
antenna is carrying a high value of cur- 
rent, a greater amount of radiation 
occurs. 

(4) When the vertical downlead of a bent 
antenna is approximately a quarter- 
wavelength, the current in the downlead 
falls to a minimum value at the end 
connected to the transmitter. Here the 
radiation resistance of the antenna has a 
high value because the ratio between the 
radiation resistance and the ohmic re- 
sistance of the antenna is a maximum, 
and a large proportion of the power 
applied to the antenna is radiated. 
When using a downlead of 1 auarter- 
wave, a parallel resonant circuit should 
be employed for antenna tuning which 
requires high impedance for proper im- 
pedance matching; but if the downlead 
is 1 eighth-wavelength or shorter, series 
tuning is used to provide the required 
low impedance. 

(5) Even when very short vertical downleads 
are used, the addition of a horizontal 
flattop to form a bent antenna produces 
the advantages mentioned above. Bent 
antennas of only a quarter-wavelength 
(including downlead) frequently are used 
in the field. The radiation produced by 
such antennas is considerably greater 
than would be produced by a simple 
vertical antenna having the same length 
as the height of the bent antenna. 

(6) When bent antennas are used at low 
frequencies, it is common to construct 
the flattop of several connected con- 
ductors. This increases c onsiderably the 
capacitance between the flattop and the 
ground. As a result, the resonant fre- 
quency of the antenna is reduced, and 
the antenna operates as a simple vertical 
antenna of much greater height. The 
higher capacitance produced by this type 



SET 



TM «««-!■• 



Figure 115. Inverted-L military antenna. 



of flattop raises the position of the current 
maximum still higher above the ground, 
with the resultant advantages previously 
pointed out. 
6. Military Types. 

(1) One common military bent antenna used 
at frequencies from about 1.5 to 12.5 mc 
(fig. 115) is an inverted-L type with a 
single-wire counterpoise. This antenna 
is designed to operate with a total length 
of about 1 quarter-wavelength at a lower 
portion of its frequency range and 3 
quarter-wavelengths at the upper por- 
tion. It affords a low-impedance load 
on the radio set with which it is used. 



(2) 



INSULATOI 



This antenna is particularly suitable for 
ground-wave transmission, although it 
is very efficient for sky-wave use. 
Small jumpers are provided at the various 
insulators so that these may be shorted 
out if it is required to increase the length 
of the flattop and counterpoise. Clip 
leads at E and Z connect the counterpoise 
and the flattop to the leads which run to 
the radio set. When operation from 1.5 
to 2 mc or from 4.5 to 6 mc is required, all 
jumpers are connected so that the lengths 
of the antenna and the counterpoise are 
each 100 feet. When operation from 2 
to 3 mc or from 6 to 9 mc is required, the 




MAST 



GUYS 



Figure 116. Crowfoot antenna. 



127 



connections at A and X are broken. The 
antenna and the counterpoise then are 
each 80 feet long. When operation from 
3 to 4.5 mc or from 9 to 12.5 mc is re- 
quired, the connections at A, B, C, D, X, 
and Y are broken, making the length of 
the antenna 60 feet and the length of the 
counterpoise 45 feet. 

(3) Another inverted-L antenna used for 
military communication in the frequency 
range below 800 kc consists of a flattop 
constructed of five parallel conductors 
from 250 to 400 feet long and separated 
from each other by about 3 feet. The 
vertical downlead is connected to each 
of the flattop conductors at one end of 
the antenna. An extensive underground 
radial ground system is used with this 
antenna. 

(4) Still another inverted-L antenna used for 
low-frequency military communication 
is shown in figure 116. The flattop is 
seen to consist of three conductors, each 
100 feet long, which are joined at one end 
and fan out at the other end to a maxi- 
mum separation of about 30 feet. A 
counterpoise is used with this antenna, 
which is shaped like the flattop. Because 
of the appearance of this antenna, it 
commonly is referred to as a crowfoot 
antenna. 

75. Folded-top Antenna 

a. A folded-top antenna is a modified bent an- 
tenna in which the flattop is folded in such a way 
that it prevents radiation. If radiation from the 
flattop is prevented, or at least reduced consider- 
ably, more energy can be radiated from the vertical 
downlead of the antenna. The main advantage 
of preventing radiation from the horizontal flattop 
is that this part of the antenna may produce con- 
siderable radiation that is horizontally polarized. 
In addition, this energy is radiated at high vertical 
angles. Since the radiation does not add to the 
vertically polarized ground wave required, its 
elimination will improve the operation of the 
antenna. Another advantage of the folded-top 
antenna is that less horizontal space is required for 
its erection. 

6. The simplest method of preventing radia- 
tion of energy from the flattop portion of the an- 
tenna is to fold the flattop in such a way that ad- 



jacent sections carry current flowing in opposite 
directions. In this way, the field produced around 
one section is opposite in direction to that pro- 
duced around the adjacent section. As a result, 
almost complete cancellation of fields occurs, and 
radiation is largely prevented. Unless an even 
number of sections is used, appreciable cancella- 
tion will not occur. 

<:. Two folded-top antennas are shown in figure 
117. In both cases, the quarter-wave flattop 
which is required to bring the current maximum 
to the top of the vertical downlead is folded in 
such a way as to prevent radiation from the flat- 
top. In A, the downlead to the transmitter is 
connected to one end of the folded section. This 
section consists of a quarter-wavelength of wire 
that is doubled back on itself so that the over-all 
length of the folded section is an eighth-wavelength. 
No te that the two wires forming the folded section 
are connected at the left and are not connected at 
the right. At any instant, the current in the two 
wires flows in opposite directions as shown by the 
arrows. Consequently, the field produced by one 
conductor is opposite in direction to that produced 
by the other conductor. Therefore, negligible 
radiation occurs from the folded section. 

d. In B, the downlead is connected to the center 
of the folded flattop. Note that the downlead is 
connected to the left half of one of the two con- 
ductors forming the folded top, and the direction 
of current flew in the folded quarter- wave section 
is as indicated by the arrows. Other arrange- 
ments can be used in addition to those shown. 
For example, the top quarter-wave section can be 
folded into four lengths, each of which is a six- 
teenth-wavelength long. 

76. Top-loaded Antenna 

a. Both the bent and folded top antennas oper- 
ate at frequencies that are much lower than the 
length of the vertical portion of the antenna would 
seem to indicate. The length not supplied by the 
vertical downlead is furnished b}' the bent or 
folded flattop. As a result, the standing wave of 
current appears higher on the vertical section of 
the antenna and its radiation resistance rises. 
Consequently, greater effective radiation occurs 
and at smaller vertical angles. 

b. Another method of increasing the effective 
length of a vertical antenna to obtain these ad- 
vantages is to use top loading. This usually is ac- 
complished by adding a concentrated amount of 



128 



/ 

FOLDED 
FLATTOP 



VERTICAL' 
DOWNLEAD 



t 

TO 

TRANSMITTER 



1— <CZ 

/ 

FOLDED 
FLATTOP 



VERTICAL 
DOWNLEAD 



TO 

TRANSMITTER 



B 



TM 666-121 

Figure 117. Folded-top antennas. 



capacitance or inductance at or near the top of the 
vertical antenna. Such an antenna is a top-loaded 
antenna. 

c. When inductance is used, the inductor simply 
is inserted in series with the antenna near the top. 
When capacitance is used, a capacitor cannot be 
inserted, in series with the antenna, since this 
would reduce the total capacitance of the antenna, 
making it appear electrically shorter rather than 
longer as is desired. Shunt capacitance must be 
used instead so that the total capacitance be- 
tween the antenna and ground is increased. The 
most common method of producing the required 
shunt capacitance is to use a disk or a hat made of 
sheet metal, mesh, or wire skeleton. The disk is 
centered on the top of the antenna and mounted 
at right angles to it. Such an arrangement pro- 
vides an added capacitance of about 1 /i/xf (micro- 
microfarads) for each inch of disk diameter. 

d. Sometimes both inductance and capacitance 
are used. The hat then must be insulated from 
the top of the vertical antenna and an inductor 
inserted between the hat and the antenna. The 
inductor frequently is made variable so that an 
adjustment of the amount of the top-loading is 
possible. This is more convenient than trying to 
make such an adjustment by varying the amount 
of shunt capacitance, since this would involve 
a change in the size of the top-loading disk. 

e. In some portable or mobile installations 
where top loading is used, the top-loading coil is 
inserted near the top of the antenna and a large 
metal shield is installed around the coil. The 
shield not only affords protection for the coil but 
also provides some shunt capacitance for top- 
loading the antenna. Where the top-loading coil 
and its shield would cause the antenna to be un- 
stable physically because of top-heaviness, the 



coil and shield are moved to the center of the 
antenna. Although this is not as effective as 
true top-loading, the radiation produced by such 
a center-loaded antenna is better than would be 
produced by a single loading coil at the base of 
the antenna or in the transmitter itself. 

/. The low-frequency bent antennas already 
discussed actually use some top-loading to inert.. 
the electrical length of the antenna and to produce 
the advantages mentioned above. The antennas 
shown in figure 116 utilize two or more insulator? 
as the flattop to provide an increase in shunt 
capacitance. The term top-loaded antenna, how- 
ever, usually refers only to those vertical antennas 
in which the shunt capacitance is supplied by a 
structure (such as a disk) the size of which is small 
compared with the length of the antenna. Be- 
cause of the small size of the top-loading disk, 
little radiation is produced. 

g. Figure 118 shows the current distribution sa 
top-loaded antennas which are somewhat shorter 
than a quarter-wavelength. The current is the 
same as would be produced if the top-loading disk 
or coil were removed and the actual height of the 
antenna extended as shown. 

h. In general, as the ground resistance is 




TM 666-122 

Figure 118. Top-loaded antennas. 

129 



increased, the size of the top-loading disk can be 
reduced. However, large ground resistances re- 
quire top-loading coils with higher inductance 
values. As the size of the top-loading disk is 
increased, the effective length of the antenna is 
increased. For example, the length of the antenna 
shown in A can be reduced further physically if 
the top-loading disk is increased in size. The 
actual size of the top-loading disk or coil required 
has been determined experimentally for a large 
number of special cases. 

i. It is desirable to increase the radiation resist- 
ance of an antenna so that a greater proportion of 
the input power can be radiated. Assume that a 
vertical antenna having a length of 0.2 wavelength, 
used without a top-loading disk, has a radiation 
resistance of 20 ohms. If a small top-loading 
disk is installed which has such a size as to increase 
the effective length of the antennas by about 0.05 
wavelength, the radiation resistance is increased 
to 34 ohms. If the size of the disk is increased so 
that the effective length of the antenna is increased 
by about 0.1 wavelength, the radiation resistance 
rises to 45 ohms. A further increase in disk size, 
so that the effective antenna length is increased 
by about 0.15 wavelength, causes the radiation 
resistance to rise to 50 ohms. 

77. Mast and Tower Radiators 

a. Description. 

(1) Most of the antennas discussed so far 
have been constructed of suitably sup- 
ported wire. Since vertical supports are 
required for such antennas, it seems 
reasonable to consider use of the support 
itself as the antenna. This support must 
be constructed of metal so that it can be 
used as the antenna. Metal masts are 
used in the field and large metal towers 
are used for fixed-station installation as 
vertical radiators. 

(2) Metal masts usually are made of aluminum 

alloy or steel tubing, sectionalized with 
metal coupling so that they can be taken 
apart for easy portability. Some short 
antenna masts, approximately 25 feet 
high, use a telescoping section construc- 
tion. Masts usually are supported by 
means of guys. 

(3) Metal antenna towers are designed to be 

either self-supporting or guyed. Both 
three-sided (triangular) and four-sided 



(square) construction is common. The 
self-supporting tower has a wide base so 
that no guying is needed. The guyed 
tower, on the other hand, usually has a 
fairly uniform cross section. 
(4) Mast and tower radiators can be sub- 
divided into insulated and noninsulated 
types. An insulated mast or tower uses 
special compression- type base insulators 
that carry the weight of the structure and 
handle the r-f voltage that exists. A 
spark gap is used across the base insulator 
for lightning protection (fig. 119). In 



CLEARANCE 




TM 666-123 

Figuie 119. Typical insulated guyed-touer radiato). 

noninsulated towers and masts, the base 
of the structure is in direct contact with 
the base support which is in the earth. 
When the insulated tower is used, the 
energy applied to the tower must be 
series-fed. Shunt feeding is used with the 
noninsulated tower. 
b Operation. 

(1) The operation and radiation of a mast or 
tower radiator are similar to the opera- 
tion and radiation of a grounded vertical 



130 



antenna constructed of wire. Because of 
the greater cross-sectional area of these 
radiators, reduction of ohmic resistance 
results in a slight increase in over-all 
efficiency and in electrical length. 

(2) When a self-supporting tower is used, the 

tapering construction of the tower alters 
the current distribution so that it is no 
longer sinusoidal. Instead of the current 
rising sinusoidally at increasing distances 
from the tower top, the current increase is 
somewhat more gradual. Since the elec- 
trical length of the tower is reduced, the 
radiation resistance falls, and increased 
high-angle radiation occurs. This con- 
dition is reduced by adding a top-loading 
disk or hat on top of the tower. 

(3) When insulated masts or towers are used, 

the output of the transmitter is applied 
between the lower end of the structure 
and ground. The transmitter output is, 
in effect, connected directly across the 
base insulator (S), as shown in A of 
figure 120, and is referred to as series 
feeding. 

(4) The output of the transmitter is con- 
nected through a capacitor to point X, 
about one-fifth of the way up the tower. 
The inclined wire usually makes an angle 
of about 45° with respect to ground. The 
exciting voltage from the transmitter is 
developed between point X and ground, 
across the lower section of the tower. 



This section can be considered to be a 
portion of a one-turn loop made up of the 
inclined wire (with capacitor) , the portion 
of the tower between point X and 
ground, and the ground return between 
the bottom of the tower and the trans- 
mission-line ground connection. Since 
the transmission line usually sees an in- 
ductive reactance in the direction of 
point X, a series capacitor is used to 
cancel out this reactance. Noninsulated 
masts or towers using this arrangement 
are shown in B. 
c. Military Types. 

(1) One type of tower radiator used for mili- 

tary communication involving trans- 
mitter powers of several kw (kilowatts) 
is 180 feet high and is guyed for support. 
The tower is an insulated type using a 
single compression insulator at its pointed 
base. The tower consists of nine 20-foot 
sections, triangular in shape and lattice 
braced, with guys at five different levels, 
three guys to each level. 

(2) Another military tower radiator used for 

transmitter powers up to about 1 kw is 
125 feet high and is self-supporting, a 
square construction with insulators being 
used at the tower base. 

(3) Still another military tower radiator used 
for transmitter powers up to about 1 kw 
is 90 feet high and is guyed for support. 
This tower is an insulated type with 




UNDERGROUND. 
RADIAL GROUND 



SERIES -FED INSULATED 
TOWER 



TO ^ 
TRANSMITTER 



•sssr 




UNDERGROUNO 
RADIAL GROUND 



t-f-TT- 
ROUND \ 



SHUNT-FED NONINSULATED 
TOWER 



B 



TM 666-124 



Figure 120. Series- and shunt-fed tower radiators. 



131 



triangular, lattice construction, equipped 
with one set of guys located 30 feet from 
the tower top. 

78. Ground-plane Antenna 

a. A ground-plane antenna consists of a quarter- 
wave vertical radiator which, in effect, carries its 
own artificial ground. The artificial ground or 
ground plane consists of a flat disk of metal or a 
number of metal rods or spokes located at the 
bottom of the radiator and usually at right angles 
to it (A of fig. 121). Since the metal disk or 



spokes are not connected directly to ground, they 
may be referred to as a counterpoise. This term 
is used rarely, however, this part of the antenna 
usually being called simply an elevated ground 
plane. 

b. The ground-plane antenna is used when 
nondirectional horizontal radiation or reception is 
required. It is particularly useful in the very- 
high-frequency range and higher. At these fre- 
quencies, the length of a vertical quarter-wave 
antenna is not great. Any desire to operate such 
an antenna in conjunction with the actual ground 
would create high ground losses and would prevent 



QUARTER -WAVE 
VERTICAL RADIATOR 




132 



efficient radiation or reception. The ground-plane 
antenna, on the other hand, is usually well elevated 
so that ground losses are minimized. 

c. The elevated ground plane also prevents 
circulating currents from flowing in a vertical 
metal mast that might be used to support the 
antenna. These currents, if not prevented, would 
cause the vertical support itself to radiate in the 
same manner as a long-wire antenna. As a result, 
undesired high-angle radiation would be produced. 

d. The radiation produced by a vertical quarter- 
wave grounded antenna erected adjacent to the 
earth itself is maximum along the surface of the 
earth (at a vertical angle of 0°). The intensity of 
the radiation falls off at higher vertical angles 
until, at a vertical angle of 90°, no radiation occurs. 
A side view of this radiation pattern is shown 
dashed in B of figure 121. Since this type of 
radiation occurs at all horizontal angles, a top 
view of the pattern would be circular. When a 
ground-plane antenna is used, the limited size of 
the elevated ground plane alters the radiation as 
shown, and maximum radiation is no longer along 
the horizontal plane but occurs at some angle 
above. 

e. When maximum radiation is required in the 
horizontal direction, it is common practice to bend 
down the spokes forming the elevated ground 
plane to an angle of about 50° below the hori- 
zontal. When the solid metal construction is 
used, the elevated ground plane takes the form of 
a cone, and the lobes of maximum radiation, shown 
in B, are pulled downward to a much lower 
vertical angle. 

/. In almost all cases, coaxial line is used to feed 
the ground-plane antenna. The inner conductor 
of the coaxial line is connected to the quarter-wave 
vertical radiator ; the outer conductor is connected 
to the elevated ground plane. 

g. The input impedance of a ground-plane 
antenna with elevated ground plane at right angles 
to the radiator is between 20 and 25 ohms. Since 
this is a lower value of impedance than is found in 
most coaxial lines, a quarter-wave matching sec- 
tion sometimes is inserted between the antenna 
and its transmission line. The matching section 
can be constructed of two quarter-wave sections 
of coaxial line connected in parallel to produce 
the required low impedance. In some ground- 
plane antennas, the radiator is folded back on 
itself so that it resembles one-half of a folded 
dipole. Under these conditions, the input imped- 
ance of the antenna is raised to about 80 ohms, so 



that a coaxial transmission line having a charac- 
teristic impedance near this value can be used. 
When the ground-plane rods are bent downward 
below the horizontal, the input impedance is 
raised to about 50 ohms. 

79. Whip Antenna 

a. The most common antenna used for tactical 
radio communication when relatively short dis- 
tances are to be covered is the whip antenna (fig. 
122). This term is applied to almost any type of 
flexible radiator used in conjunction with portable 
or mobile radio equipment. Whip antennas rang- 
ing in length up to 25 feet are mounted on vehicles. 
Shorter whip antennas are mounted on small 
hand-held radio sets or portable sets used in the 
field. 

b. Most whip antennas are constructed of tel- 
escoping sections of metal tubing which can be 
collapsed when not in use simply by pushing one 
section into another. In this way, the antenna 
has a minimum length and portability is increased. 
In certain lightweight portable equipment, the 
antenna can be collapsed completely into the 
equipment itself so that none of it is exposed. 

c. Sometimes a whip antenna mounted on a 
vehicle must be left fully extended so that it can 
be used instantly while the vehicle is in motion. 
In such antennas, the base mounting insulator of 
the whip is fitted with a coil spring attached to a 
mounting bracket on the • vehicle. The spring 
base allows the whip antenna to be held in a 
nearly horizontal position by insulated guy lines 
so that the vehicle can be driven under low 
bridges or obstructions, although the radiation 
produced under these conditions is reduced. An- 
other advantage of the spring base is that even if 
the antenna is vertical and it does hit an obstruc- 
tion, the whip usually will not break or be bent 
since most of the bending occurs at the spring 
base. 

d. One common mobile radio station installed 
in a 2%-ton, 6 by 6 cargo truck uses three such 
whip antennas — two 15-foot whips for receiving 
and a single 25-foot whip for transmitting. The 
truck chassis is used as a ground. 

e. When whip antennas are operated in the 
high-frequency range, their length usually is a 
small fraction of a wavelength. Here, large load- 
ing coils must be used to resonate the whip antenna 
properly. The radiation resistance of a short whip 



133 



I 



MAST SECTIONS 




TM 668- 1 2« 

Figure 182. Typical whip antennas. 



is very low and it is quite possible that the ohmic 
losses which result from the resistance of the load- 
ing coil, antenna connection, and the antenna 
itself may exceed the radiation resistance. The 
radiation efficiency of this antenna is low, being 
approximately 0.5 to 2 percent in the low end of 
the high-frequency range. If large whips or even 
higher frequencies are used, so that the whip length 
is a quarter- or half-wave, the output of the whip 



antenna is raised. A 25-foot whip, for example, 
is a half-wavelength long at about 18.5 mc. 

/. In the very-high-frequency band (and higher) 
short whip antennas can be a quarter- or a half- 
wavelength long. For example, a 4-foot whip is 
a quarter-wavelength long at about 60 mc and a 
half-wavelength long at about 120 mc. Under 
these conditions high radiation efficiencies are 
possible. 



134 



SECTION VII. SUMMARY AND QUESTIONS 



80. Summary 

a. The presence of an electric field about an 
antenna indicates the presence of a voltage on 
the antenna, resulting from the charge placed 
on the antenna by the transmitter. 

b. The presence of a magnetic field about an 
antenna indicates the presence of a current in 
the antenna. This current is the charge moving 
in the antenna. 

c. The electric and magne f ic fields that im- 
mediately surround an antenna, forming the 
induction field, are in space and time quadrature. 

d. Standing waves on an antenna are the result 
of incident and reflected traveling waves moving 
in opposite directions on the same conductor. 

e. The standing wave of voltages on a half- 
wave antenna is so distributed that it has maxi- 
mum amplitude at the antenna ends and minimum 
amplitude at the center of the antenna. 

/. The standing wave of current on a half- 
wave antenna is so distributed that it has maximum 
amplitude at the center of the antenna and 
minimum amplitude at the antenna ends. 

g. The velocity of wave travel on an antenna 
is lower than in free space. As a result, the 
physical length of a half-wave antenna is about 
5 percent shorter than its electrical length. 

h. A current flowing in an antenna must 
contend with radiation resistance, ohmic resis- 
tance, and leakage resistance. 

i. In the half-wave antenna, the radiation 
resistance is large compared to the other resist- 
ances, so that most of the available energy is 
radiated. 

j. If an antenna is cut to a length of exact 
resonance, the reactance is zero and the impedance 
of the antenna is purely resistive. If the antenna 
is made somewhat shorter, capacitive reactance 
is present; if the antenna is made longer, induc- 
tive reactance is present. 

k. A transmission line is a device for guiding 
electrical energy from one point to another. 
Such a line has electrical constants of inductance, 
capacitance, and resistance distributed along its 
length. 

A line terminated in a resistance equal to 
its characteristic impedance is said to be termi- 
nated properly. 

m. The ratio of maximum to minimum voltage 
along a transmission line is called the standing- 



wave ratio. This ratio provides a measure of 
the energy reflected. 

n. A properly matched line is nonresonant. 
It produces no reflection of energy and there are 
no standing waves. As a result, there is maxi- 
mum transmission of energy. 

o. A resonant line is terminated in a load not 
the same as Z . This line has standing waves 
so that the standing-wave ratio is greater than 1. 
Such a line, when cut to certain lengths, exhibits 
the properties of a resonant circuit. 

p. The transmission line used to couple the 
transmitter to the antenna sometimes is called 
a feeder. 

q. In determining how to connect a feeder line 
to the antenna, it is necessary to consider the 
antenna impedance at the connection point and 
the type of antenna that is used. 

r. A single-wire transmission line must be 
connected to a suitable point on the antenna 
between the end and the center in order to effect 
an impedance match. Two-wire and other lines 
usually are connected at the center or at the 
end of the antenna. 

s. A delta-matching section used with a two- 
wire line is made by fanning out the end of the 
transmission line as it approaches the antenna. 

t. T-matching and J-matching are two com- 
monly used impedance-matching systems with 
two-wire feeders. 

u. Stub matching, which utilizes sections of 
transmission line, frequently is used to connect 
a nonresonant feeder line to an antenna. 

v. The radiation pattern is a measure of the 
energy radiated from an antenna taken at various 
angles and at a constant distance from the 
antenna. 

to. Most radiation patterns are plotted in polar 
coordinates. This system of graphing has the 
advantage over the rectangular coordinate graph 
in that positions are indicated on the graph which 
are directly related to the actual positions at 
which measurements are taken around an antenna. 

x. The radiation pattern of a dipole is circular 
in a plane at right angles to the antenna. Broad- 
side, the pattern resembles a somewhat flattened 
figure 8. 

y. The beam width is measured between the 
points where the field strength falls off to 0.707 
of maximum, or where the power falls off to 0.5 
of maximum. 



135 



2. The presence of ground beneath an antenna 
affects the radiation pattern. 

aa. The energy received at a distant point is 
the sum of the direct wave and the ground- 
reflected wave. 

ab. The reflection factor is a term by which the 
free-space radiation pattern of an antenna must be 
multiplied in order to determine the radiated field 
strength of a practical antenna at a given vertical 
angle. 

ac. In general, ground reflections result in the 
breaking up of the vertical -plane radiation pattern 
into a number of separate lobes. 

ad. The radiation resistance of a practical, half- 
wave antenna located over a ground plane may 
have any value of radiation resistance from to 
almost 100 ohms, depending on the exact height. 

ae. Because the ground has resistance and is not 
a perfect reflector, a portion of the wave which 
ordinarily would be reflected is absorbed in the 
ground resistance. This absorption constitutes 
ground losses. 

of. With an imperfectly conducting ground, the 
vertical radiation pattern shows a series of high 
and low signal strengths rather than a series of 
double-amplitude lobes separated by well defined 
nulls. 

ag. A ground screen can be used to establish 
accurately the location of the reflecting ground 
plane and to minimize ground losses. The ground 
screen consists of a fairly large area of metal mesh 
or screen which is laid on the surface of the ground 
under the antenna. 

ah. The polarization of a radio wave is deter- 
mined by the direction of the electric flux lines 
with respect to the surface of the earth. 

ax. In general, the characteristics of an antenna 
used for transmitting are much the same as when 
the antenna is used for receiving. The antenna 
receives best in those directions in which i t radiates 
best. 

aj. A single-wire, half-w r ave antenna is one 
which is constructed of a single conductor of proper 
length. 

ak. The folded-dipole antenna consists of an 
ordinary, half-wave antenna which has one or 
more additional conductors connected across the 
ends of the antenna. These are mounted parallel 
to the dipole elements at a distance that is a very 
small fraction of a wavelength. 

al. A coaxial or sleeve antenna consists of a 
quarter-wave vertical radiator with a quarter- 
wave sleeve that fits around the antenna support 



mounted just below it. The sleeve acts as the 
lower portion of the antenna. 

am. The conical antenna is a wide-band antenna 
whose elements are constructed in the form of 
cones. 

an. A quarter-wave antenna which operates in 
conjunction with ground operates as a resonant 
antenna. 

ao. The current in a quarter-wave, grounded 
antenna is maximum at the grounded end and the 
voltage is a minimum. The impedance of such an 
antenna is minimum at the grounded end. 

ap. The radiation resistance of a grounded, 
quarter -wave antenna is about 36 ohms. Shorter 
antennas have lower radiation resistance. 

aq. Loading coils are used frequently to increase 
the electrical length of short, grounded, quarter- 
wave antennas. 

ar. Maximum radiation from a grounded, 
quarter-wave antenna occurs at right angles to the 
antenna and along the surface of the ground. No 
radiation occurs directly over the antenna. 

as. With grounded antennas, it is especially 
important that the ground connection be good and 
that the ground conductivity is high. 

at. Large, underground, radial grounds are 
common for fixed-station installations. Ground 
rods also are useful in providing good ground con- 
nection. 

au. A counterpoise is utilized when an actual 
ground connection cannot be used because of the 
high resistance of the soil or because a large buried 
ground system is not practical. 

av. A bent antenna is a grounded antenna which 
is constructed so that a portion of it is horizontally 
mounted. 

aw. The advantages of a bent antenna are the 
reduction in high-angle radiation and an increase 
in the amount of radiation produced. 

ax. A folded-top antenna is a modified bent 
antenna in which the flattop is folded in such a 
way that it prevents radiation. 

ay. Top-loading increases the electrical length 
of the antenna. When top-loading is used, the 
radiation resistance of the antenna is increased and 
the radiation efficiency of the antenna rises. 

az. Metal masts and towers can be used as 
grounded vertical radiators. These are either 
self-supporting or guyed, insulated or noninsu- 
lated. 

6a. A ground-plane antenna consists of a quar- 
ter-wave vertical radiator which carries its own 
ground consisting of a flat, metal disk or a number 



136 



of metal rods or spokes located at the bottom of 
the radiator and at right angles to it. 

bb. The whip antenna, which is a short, flexible, 
vertical radiator, is the most common antenna used 
for tactical radio communication when short 
distances are to be covered. 

81. Review Questions 

a. What is the direction of the electric flux 
lines that surround an antenna? 

b. What is responsible for the magnetic field 
that surrounds an antenna? 

c. Explain the expression, space and time quad- 
rature. 

d. What is meant by a loop, a node"! 

e. Describe the distribution of the standing 
waves of voltage and current along a half-wave 
antenna. 

/. Give several factors that cause the velocity 
of wave travel on an antenna to be lower than in 
free space. 

g. What is the free-space length (in feet) of a 
half-wave at a frequency of 7 mc? 

h. How long is a half-wave antenna which is to 
operate at 7 mc? 

i. What causes radiation resistance? Ohmic 
resistance? Dielectric resistance? 

j. What is the value of the radiation resistance 
as measured at the center of a half-wave antenna? 

k. Why is it important to minimize the react- 
ance seen by a transmission line that is connected 
to an antenna? 

I. What is the purpose of a transmission line? 

m. Why is it advantageous to terminate a 
transmission line properly? 

n. Give three methods by which transmission 
lines dissipate power. 

o. Describe the current and voltage distribution 
on closed-end lines. On open-end lines. 

p. How does the standing-wave ratio indicate 
the amount of line mismatch? 

q. What is the purpose of an impedance-match- 
ing device? 

r. Describe several types of practical trans- 
mission lines. 

s. What is a tuned line? An untuned line? 

t. What lengths of open-end, resonant, trans- 
mission line are required to cause the line to act 
as an impedance transformer? 

u. Where is a current-fed, half-wave antenna 
fed? 

v. If voltage feeding is required, where should 



the feeders be connected to the half-wave antenna? 

w. Give some advantages and disadvantages 
of the single-wire feeder. 

x. Give some advantages and disadvantages of 
the delta-matching system. 

y. What are some characteristics of the T- 
matching system? 

z. Describe the J-matching system. 

aa. Give several examples of impedance match- 
ing by the use of stubs. 

ab. Describe the Q-matching system. 

ac. What is an isotropic radiator? Give an 
example. 

ad. What constitutes a lobe on a radiation 
pattern? A null? 

ae. Do radiation patterns actually picture the 
radiation produced by an antenna? 

of. Distinguish between a dipole and a doublet 
as used in this section. 

ag. Why does the presence of ground beneath 
the antenna affect the radiation pattern? 

ah. What is meant by an image antenna? 

ai. What are the minimum and maximum 
values of reflection factor? 

aj. What determines the value of the reflection 
factor? 

ak. Describe the vertical-plane radiation pat- 
tern produced by a horizontal half-wave antenna 
when its height above ground is increased gradu- 
ally. 

al. W"hy does the radiation resistance of an 
antenna change at differing heights above ground? 

am. How does the frequency of the radiation 
from an antenna affect the amount of ground 
losses? 

an. What effect does an imperfect ground have 
on the radiation resistance of an antenna? 

ao. Give two specific advantages that are gained 
by the use of a ground screen. 

ap. How does the orientation of an antenna 
affect the polarization of the radiation produced? 

aq. Give some advantages of horizontal polari- 
zation. 

ar. Give some advantages of vertical polariza- 
tion. 

as. What is meant by the reciprocity of an 
antenna? 

at. What is the input impedance of a folded 
dipole antenna? 

au. Explain the greater bandwidth of the folded 
dipole. 

av. What is the function of the sleeve or skirt 
on the coaxial antenna? 



137 



aw. What are the characteristics of a coaxial 
antenna? 

ax. Describe some simple microwave antennas. 

ay. What is a Marconi antenna? 

az. What is the current and voltage distribution 
on a grounded, quarter-wave antenna? 

ba. What is the function of a loading coil? 

66. Describe the radiation pattern produced by 
a vertical, half-wave antenna. 

6c. Compare the radiation resistance of a quar- 
ter-wave antenna with an antenna that is only a 
tenth-wavelength long. 

bd. Describe a radial ground. 

be. What is the purpose of a ground rod, and 
how does chemical treatment affect the operation 
of a ground rod? 

bf. For what purpose is a counterpoise used? 

bg. When are bent antennas used? 

bh. What lengths are desirable for the flattop in 
a bent antenna? 



bi. Why do low-frequency, bent antennas use 
flattops constructed of several connected conduc- 
tors? 

bj. What is the crowfoot antenna? 

bk. Why is it advantageous to minimize radia- 
tion from the flattop of an antenna? 

bl. Describe some devices commonly used for 
top»-loading an antenna. 

bm. Distinguish between an insulated and a 
noninsulated tower. 

bn. When is a tower radiator series-fed? Shunt- 
fed? 

bo. What is the radiation pattern produced by 
a ground-plane antenna? 

bp. What advantages occur when the rods form- 
ing the elevated ground plane in a ground-plane 
antenna are bent downward below the horizontal? 

bq. Why do some whip antennas have spring 
bases? 



136 



CHAPTER 4 
LONG- WIRE ANTENNAS 



82. Introduction 

Long-wire antennas are long single wires, 
longer than 1 half-wavelength, in which the 
current in adjacent half-wave sections flows in 
opposite directions. Such antennas have two 
basic advantages over the antennas discussed in 
the previous chapters. These advantages are 
increased gain and directivity. 

a. Antenna Gain. 

(1) All of the antennas discussed so far 
have been basic half- and quarter- 
wave antennas that radiate equally in 
all directions. Greatest amounts of 
power are radiated in directions that 
are broadside to the antenna itself, and 
very little power is radiated off the 
antenna ends. Consequently, the basic 
antennas already discussed have a certain 
degree of directivity, which is the ability 
to radiate and receive energy better in some 
directions than in others. 

(2) An isotropic antenna is one that radiates 
equally in all directions. In actual 
practice, every antenna radiates more 
energy in certain directions than in 
others. The imaginary isotropic an- 
tenna, however, can be used only as a 
standard for comparison. 

(3) Assume that a certain amount of power 
is applied to an isotropic radiator. 
This produces a field having a certain 
strength at a distant receiving antenna. 
If the same amount of power is applied 
to a half-wave antenna that is far 
removed from ground, this antenna will 
produce a field the strength of which is 
greater in certain directions than that 
produced by the isotropic radiator. 
This increase in field strength in some 
directions can be produced only at the 
expense of field strength in other di- 
rections; that is, an increase in field 
strength in certain directions must be 
accompanied by a decrease in field 



strength in other directions. As a 
result, a distant receiving antenna will 
have a greater or lesser amount of 
induced voltage depending on its position 
with respect to the orientation of the 
radiating antenna. The half-wave an- 
tenna produces an increase or gain in 
field strength in the direction at right 
angles to itself, and a decrease or loss in 
field strength in other directions. In 
the direction of maximum radiation, 
the antenna produces an increased field 
strength (field strength gain) that is 
about 1.28 times that produced by the 
isotropic radiator. This is equivalent 
to an increase in power (power gain) 
of 1.64 times that obtained from the 
isotropic radiator. In other words, if 
the power applied to the isotropic 
radiator were increased 1.64 times, ex- 
actly the same field strength would be 
produced for all directions as is obtained 
from the basic half-wave antenna in the 
direction of maximum radiation. 
(4) Gain frequently is expressed in terms of 
the logarithmic ratio, the decibel. In 
order to convert the figures to decibels, 
it is necessary only to use the formulas 
below: 

Gain (in db) = 20 logi x field-strength ratio 
Gain (in db) = 10 log I0 x power ratio. 

When either or both of these formulas are 
solved, it is found that the gain of the 
basic half- wave antenna is 2.15 db over 
that of the isotropic radiator. 
b. Calculation of Gain. 

(1) Since no antenna is truly isotropic, it is 
common practice to use a basic half-wave 
antenna as a standard for reference. 
The reference field strength is the field 
intensity at a fixed point produced by the 
half-wave antenna in the direction of 



139 



its maximum radiation. The reference 
power is the power applied to the stand- 
ard antenna. If any antenna produces 
a greater field strength at the same fixed 
point than does the standard antenna, it 
is said to have gain with respect to the 
standard. Conversely, if the antenna 
produces less field strength at the fixed 
point than does the standard antenna, it 
is said to have loss with respect to the 
standard. 

(2) In actual practice, the procedure is first 
to set up the antenna to be checked. A 
given amount of input power is applied 
and the field strength is measured at a 
distant receiving point. Then, the half- 
wave antenna is set up at the same posi- 
tion and height above the earth, and 
oriented so that a field having the same 
polarization as the original antenna is 
produced. Exactly the same power is 
applied to the half-wave antenna as was 
applied to the other antenna. The field 
strength then is measured at the same 
distant receiving point. Comparison be- 
tween the two field strengths indicates 
whether the antenna being checked pro- 
duces a gain or a loss in field strength 
compared with the reference antenna. 

(3) Another method used to measure the gain 
(or loss) of an antenna involves an actual 
measurement of the input power to the 
antenna. The amount of input power 
applied to the antenna to be checked is 
measured, the field strength at a certain 
distant point is noted, and the half -wave 
antenna is set up as in the preceding 
method. The power applied to this 
reference antenna then is adjusted until 
exactly the same field strength is pro- 
duced at the distant point. If more 
power must be applied to the reference 
antenna to produce the same field 
strength at the point as produced by the 
antenna under test, the antenna under 
test has a gain with respect to the 
reference antenna. On the other hand, 



if less power must be applied to the half- 
wave reference antenna to produce the 
same field strength, the antenna under 
test has a loss with respect to the refer- 
ence antenna. The ratio between the 
two input powers (reference antennas 
divided by tested antennas) gives the 
gain (or loss) of the antenna being tested. 
Because an accurate measurement of dif- 
ferent field strengths cannot be made as 
easily as an accurate measurement of 
different antenna input powers, this 
method is preferred to that described in 
(2) above. 

(4) If the antenna under test produces the 
same field strength at a certain distance 
with exactly the same power applied as is 
produced by the reference half-wave 
antenna, there is neither a gain nor a loss. 
The ratio between the two input powers 
is unity (1 to 1) as is the ratio between 
the two field strengths with the same 
input power. Since the logarithm of 1 is 
zero, the gain in decibels as calculated 
by the formulas previously given is db. 
This is simply another way of saying that 
there is neither a gain nor a loss. The 
reference level of field strength produced 
by the half-wave antenna, or a level 
equal to the reference level, is referred to 
as the zero db level. The standard half- 
wave antenna which produces the refer- 
ence field strength often is referred to as 
the zero db antenna. 

(5) If a certain antenna has a gain of 10 db, it 
produces a field strength that is over 
three times greater than that produced by 
the half-wave antenna with the same 
input power. This same antenna pro- 
duces the same field strength as that 
produced by the half-wave antenna when 
the power applied to the half-wave 
antenna is 10 times greater than that 
applied to the antenna under test. The 
chart given below gives the db gain or loss 
for various field strength and power 
ratios. 



140 



Db gain or loss 


Field- 
strength 
ratio 


Power ratio 


- — 


1. 


1. 


1 - 


1. 12 


1. 26 




1. 26 


1. 56 




1. 41 


1. 99 


4.5 — - 


1. 68 


2. 82 


6 


1. 99 


3. 98 


8 - - 


2. 51 


6. 31 


10 - - 


3. 16 


10. 


12.5 - -- 


4. 22 


17. 8 


15 - 


5. 62 


31. 6 


20 


10.0 


100.0 


30 -- 


31. 6 


1, 000. 


Aft 


100. 


10, 000. 







c. Directivity. 

(1) Since all antennas are directional to a 
certain degree, the term directional usu- 
ally is applied only to those antennas that 
are rather highly directional. The main 
advantage to be gained from the use of 
the long-wire antennas and arrays is in 
their greater directional qualities. These 
antennas all concentrate a larger amount 
of the available radiated energy into a 
smaller sector. 

(2) Some antennas are directional in some 
planes but practically nondirectional in 
others. Consider, for example, the basic 
half-wave antenna that is mounted in a 
vertical position. If a vertical plane is 
passed through the center of the antenna 
and the radiation pattern is drawn on that 
plane, the pattern would take the form of 
a figure 8. Maximum radiation occurs 
in the two directions that are at right 
angles to the antenna itself and no 
radiation occurs off the ends. The 
antenna is said to have two lobes of 
radiated energy and two nulls. Conse- 
quently, this antenna is said to be bi- 
directional (it radiates in two directions) 
in the vertical plane. If the horizontal 
plane is considered, however, it is seen 
that the antenna radiates equally in all 
directions. The antenna therefore is 
nondirectional in the horizontal plane. 
When highly directional antennas are 
used, it is important to know in which 
plane the desired directivity occurs. 

(3) Highly directional antennas are designed 



to produce a large increase in radiated 
(or received) energy in one direction. 
The idea, however, may be to prevent 
radiation (or reception) in a certain direc- 
tion. For example, assume that two 
powerful transmitters are located near 
each other. To prevent these trans- 
mitters from interfering with each other, 
it is necessary to use directional antennas 
with respective nulls pointing toward 
each other. Under these conditions, the 
antennas may be adjusted to produce the 
least amount of radiated energy in the 
direction of each other, rather than the 
greatest amount of energy in any given 
direction. 

83. General Characteristics of Long-wire An- 
tennas 

a. If the length of a long-wire antenna is such 
that two or more half-waves of energy are distrib- 
uted along it, it often is referred to as a harmonic 
antenna. Consider the half-wave antenna shown 
in A of figure 1 23. At a given instant, the polarity 
of the r-f generator connected to the center of the 
antenna is positive at its left-hand terminal and 
negative at its right-hand terminal. As a result, 
current in the left half of the antenna flows toward 
the generator, whereas current in the right half of 
the antenna flows away from the generator. In 
both halves of the half-wave antenna, current 
flows in the same direction, from left to right, as 
shown by the wave of current above the antenna 
wire. 

b. Now assume that the antenna just discussed 
is increased until it is 2 half- wavelengths, as in B. 
With the r-f generator still connected at the center 
and with the same instantaneous polarities as in 
A, current in the left side of the antenna must flow 
toward the generator, and current in the right 
side must flow away from the generator. Since 
the antenna is now 2 half-wavelengths, 2 half- 
waves of current can be accommodated on the 
antenna and the current polarity is the same in 
both halves of the antenna. It is important to 
note that this is not a true long.wire or harmoni- 
cally operated antenna since there is no reversal of 
current flow in adjacent half-wave sections. In- 
stead, this arrangement is simply 2 half- wave 
antennas operating in phase at their fundamental 
frequency. Such an arrangement is called a 
driven collinear array (paragraphs 94 through 109) 



141 




TM 66S-127 

Figure 128. Harmonic and nonharmonic antennas. 



and has characteristics quite different from those 
to be discussed for the true harmonically operated 
or long-wire antenna. 

c. The antenna in B can be converted into a 
true long-wire, harmonically operated antenna 
simply by moving the generator to a current loop 
as shown in C. With the r-f generator polarity as 
shown, current flows from left to right in the half- 
wave section of the antenna. The direction of 
current flow then is reversed in the second half- 
wave section. If the generator is moved to the 
extreme end of the antenna as shown in D, the 
antenna is also a long-wire antenna, and the cur- 
rent distribution on the antenna is exactly the 
same as in C. 



d. The harmonically operated antenna, there- 
fore, must be fed either at a current loop or at its 
end for proper operation. If the antenna is any 
odd number of half-wave lengths (1%, 2%, 3%, and 
so on) so that a current loop occurs at the center 
of the antenna, center feeding can be used. 

e. As the length of an antenna is increased, it is 
natural to expect a change in the radiation pattern 
produced by the antenna. A long-wire antenna 
can be considered to be one made up of a number 
of half-wave sections fed 180° out of phase and 
spaced a half -wavelength apart. As a result, there 
is no longer zero radiation off the ends of the 
antenna, but considerable radiation occurs in the 
direction of the long wire as a result of the com- 
bined fields produced by the individual half-wave 
sections. In addition, radiation also occurs broad- 
side to the long wire. Consequently, the resultant 
maximum radiation is neither completely at right 
angles to the long wire nor completely along the 
line of the long wire. Instead, the maximum 
radiation occurs at some acute angle in respect to 
the wire, the exact angle being determined by the 
length of the antenna. 

/. It will be shown that as the length of a long- 
wire antenna is increased, the following charac- 
teristic changes occur: First, the gain of the 
antenna increases considerably compared with 
that of the basic half-wave antenna, especially 
when the long wire is many wavelengths. Second, 
the direction along which maximum radiation 
occurs makes a smaller angle with respect to the 
wire itself. Consequently, as the antenna is made 
longer, its major lobe of radiation lies closer to the 
direction of the wire itself. Third, more minor 
lobes are produced as the antenna length is 
increased . 

84, Harmonically Operated Antennas in Free 
Space 

a. Calculation of Length. 

(1) It already has been pointed out (pars. 36 
through 81) that the electrical and physi- 
cal lengths of a half- wave antenna are not 
the same because of the reduction in wave 
velocity on the antenna resulting from 
its thickness and because of end effect. 
The main factor producing end effect is 
the use of insulators at the antenna ends. 
These introduce additional capacitances 
to the antenna which lower its resonant 
frequency and increase the electrical 



142 



length of the antenna. Consequently, 
the half-wave antenna is foreshortened by 
5 percent to compensate for these effects. 

(2) Since, in a long-wire antenna, the insu- 
lators are used at the ends and not 
between adjacent half-wave sections, it 
is only the half-wave sections at the 
antenna ends that are affected by end 
effects. Therefore, a harmonically oper- 
ated antenna of 1 wavelength is fore- 
shortened by only Vk percent over-all, 
one of 2 wavelengths is foreshortened by 
1% percent over-all, and so on. A con- 
venient formula that is used to determine 
the length in feet of a harmonic antenna 
for any given frequency in megacycles is 

. t . 492 (H— 0.05) 
lengths frequency 

where H is the number of half-w&ves on 
the antenna. 

(3) Note that the foreshortening of the long- 
wire antenna is less than for the simple 
half-wave antenna. For this reason, a 
long-wire antenna with, say, 3 half-waves 
on it is slightly longer than three times 
the length of a half -wave antenna. There- 
fore, the length of the half -wave antenna 
is not an exact submultiple of the length 
of the harmonic antenna. 



E I 




2 X FUNDAMENTAL 



E I 




4 X FUNDAMENTAL 



Figure 124- Standing 



b. Standing Waves and Impedances at Various 
"equencies. 

(1) As the length of the antenna is increased 
so that it operates on higher harmonic 
frequencies, or, as the frequency applied 
to an antenna of fixed length is raised, a 
greater number of half-waves of voltage 
and current occur on the antenna. This 
is shown in figure 124, where antennas 
operating on the second, third, fourth, 
and fifth harmonics appear. The 
standing waves of voltage and current 
are 90° out of phase. 

(2) The impedance of the harmonic antenna 
at any point is determined by the re- 
actance and the resistance of the antenna 
at that point. The impedance often is 
measured at a current loop, because this 
is where the feed line usually is attached. 
When the antenna length is such that 
exact harmonic resonance occurs or if the 
reactance is tuned out otherwise, only 
resistance remains. This resistance is 
largely radiation resistance, since the 
ohmic losses of the antenna usually are 
so low that they can be neglected com- 
pared with the value of radiation resist- 
ance. 

(3) The following chart shows the approxi- 
mate radiation resistance, measured at a 
current loop, of harmonic antennas of 
various lengths. As the antenna length 



e I 




3 X FUNDAMENTAL 



E I 




5 X FUNDAMENTAL 

TM «8«-128 

on harmonic antennas. 



143 



Antenna length 

(wavelengths) 



1... 

VA- 

2__. 
2M- 
3__. 
4... 



Radiation 
resistance 
(ohms) 



Antenna length 
(wavelengths) 



Radiation 
resistance 
(ohms) 



90 


S 5 


138 


100 


; 6 


| 144 


110 


8 


1 154 


117 


10 


I 162 


122 


13 


I 170 


130 







is increased, the value of radiation 
resistance also increases. 
(4) When the frequency applied to an 
antenna is no longer the resonant fre- 
quency, a considerable reactive com- 
ponent is present. Consider a half-wave 
antenna that is fed at the center. As 
the frequency of the r-f generator is 
increased above the resonant frequency 
of the antenna, the antenna becomes 
inductive with an inductive reactance. 
This inductive reactance reaches a maxi- 
mum value and then begins to fall off as 
the frequency is raised still more. When 
the frequency is such that the antenna 
length is slightly less than a full wave- 
length, the reactance is zero. An in- 
crease in frequency above this value 
causes the antenna to behave as a capac- 
itive reactance. As the frequency is 
raised still higher, the capacitive react- 
ance reaches a maximum value and then 
falls off toward zero. When the fre- 
quency is such that the antenna is 
slightly less than IK wavelengths, the 
reactance again is zero. An increase in 
frequency above this value causes the 
antenna to behave as a inductive react- 
ance. As the frequency is raised still 
higher, the inductive reactance reaches 
a maximum value and then falls off 
toward zero. When the frequency is 
such that the antenna is slightly less than 
2 wavelengths, the reactance again is 
zero. An increase in frequency above 
this value causes the antenna to behave 
as a capacitance once more. Conse- 
quently, a complete cycle of reactance 
changes occurs as the frequency is 
increased so that electrically the antenna 
is changed from a half-wavelength to 
]K wavelengths. A similar cvcle occurs 



as the antenna is changed from 1% to 
2Y 2 wavelengths, from 2V 2 to 3% wave- 
lengths, and so on. 

(5) During the time the reactance is going 
through a cycle, the resistive component 
of the impedance also varies with the 
change in frequency. It is at a mini- 
mum value when the frequency is such 
that the antenna is about a half-wave- 
length and reaches a maximum value 
when the antenna is about 1 wavelength, 
and back to a minimum value when the 
antenna length is approximately \}{ 
wavelength. These cyclic variations in 
reactance and resistance are shown in 
figure 125. As a result of these varia- 
tions, the impedance of the antenna also 
has a cyclic variation as the frequency 
is raised from one harmonic to another. 

(6) The rate at which the impedance varies 
is increased at the higher frequencies. 
The impedance at the center of an an- 
tenna varies from minimum to maximum 
when the frequency applied increases 
from the fundamental to the second 
harmonic. A similar variation must take 
place when the frequency applied in- 
creases from the fifth to the sixth har- 
monic. From the fundamental to the 
second harmonic, however, a frequency 
increase of 100 percent occurs, and from 
the fifth to the sixth harmonic, there is a 
frequency increase of only 20 percent. 
Therefore, the impedance must change 
five times faster in the vicinity of the 
fifth harmonic than in the vicinity of the 
fundamental. 

(7) As the frequency applied to a given an- 
tenna or as the length of an antenna is 
increased, the resistive component of 
its impedance rises, and as the radiation 
resistance of an antenna rises, the length 
and losses increase. As a result, as the 
antenna varies from a half-wavelength 
to wavelengths, and the resistance 
does not drop to as low a value at \K 
wavelengths as at a half-wavelength, 
nor does it rise to as high a value at 2 
wavelengths as at 1 wavelength (fig. 125). 
This is true also as the frequency in- 
creases to 5. 6, or more times the funda- 
mental, or as the antenna increases to 3 
wavelengths, 4 wavelengths, and so on. 



144 



z 





\ 

V- 


RFS 


STANCE 

V 




RE/ 


iCTANCE 


/ 


\ 


* 



FREQUENCY — *~ F 2F 3F 4F 5F 

LENGTH > llfcX 2\ Z'.fc'X 

TM 666-129 

Figure 125. Cyclic variations of reactance and resistance 
at center of antenna. 

Although cyclic changes in impedance 
still occur at long antenna lengths or at 
high applied frequencies, the impedance 
does not rise to as high a value nor does 
it fall to as low a value as occurs at low 
harmonic frequencies or short antenna 
lengths. 
c. Directivity and Gain. 

(1) The maximum radiation from a har- 
monic antenna forms a lobe which covers 
a smaller and smaller sector as the an- 
tenna length is increased. Since a 
greater amount of power is concentrated 
into a smaller sector, the harmonic an- 
tenna has a power gain with respect to 
the half- wave antenna. 

(2) The approximate power gain of harmonic 
antennas of various lengths is shown in 
the following chart: 



Antenna length 
(wavelengths) 


Power gain 


Antenna length 
(wavelengths) 


Power gain 


1 


1. 2 

1. 4 

2. 1 

3. 1 


8 


4. 3 

5. 6 
7. 2 


2 


10 


4 


12 


6 





(3) Very little gain occurs when the antenna 
is only a few wavelengths. When the 
length becomes appreciable, however, 
considerable power gains result, and 
increased power gain is accompanied by 
greater directivity. This is true since 
an increase of power in certain directions 
is attained by reduction of power in other 
directions. 

(4) The radiation produced by a harmonic 
antenna is neither completely at right 
angles to the antenna itself nor off the 



ends of the antenna. Instead, the maxi- 
mum radiation forms lobes which lie 
closer and closer to the direction of the 
antenna itself as the length of the 
antenna increases. The angle of maxi- 
mum radiation is the angle between the 
line running through the center of the 
lobe and the antenna wire. 
(5) The approximate angles of maximum 
radiation for harmonic antennas of vari- 
ous lengths are given in the following 
chart: 



Antenna length 
(wavelengths) 


Angle of matt- 
mum radiation 
(degrees) 


Antenna length 
(wavelengths) 


Angle of maxi- 
mum radiation 

(degrees) 


1 


54 
36 
25 
20 


8 


18 
17 
16 


2 


10 


4 


12 


6 







d. Radiation Patterns. 

(1) Figure 126 shows the radiation patterns 
of harmonic antennas up to 3 wave- 
lengths. The field strength produced by 
the half-waVe antenna is shown for com- 
parison. Note that as the antenna 
length is increased, more lobes are pro- 
duced. The 1 ^-wavelength antenna, 
which operates on the third harmonic, 
has three lobes — two major lobes and 
one minor lobe, the latter lying at right 
angles to the antenna. The 3-wave- 
length antenna, which operates on the 
sixth harmonic, has six lobes — two major 
lobes and four minor lobes. 

(2) The harmonic antennas which operate on 
the even harmonics (2d, 4th, and so on) 
have an even number of ha If -wave pat- 
terns distributed along their length. 
Since the adjacent half-wave sections 
have currents of opposite phase, a distant 
point in space located equidistant from 
the ends of the antenna is acted on by 
equal and opposite fields. Cancellation 
of fields occurs and » null is produced on 
a plane at right angles to the antenna, 
cutting it at the center. On the other 
hand, harmonic antennas which operate 
on the odd harmonics have an odd 
number of half -wave sections . Complete 
cancellation of radiated fields does not 



145 





occur at points equidistant from the ends 
of the antenna because of the odd half- 
wave section. This results in a minor 
lobe being produced in a direction that is 
perpendicular to the antenna, and com- 
ing from its center. 

85. Effects of Ground 

a. The radiation patterns of a harmonic antenna 
are modified considerably by the presence of the 
earth under the antenna. Some of the energy 
radiated from the antenna travels downward 
toward the earth, where it is reflected. If the 
reflected energy arrives at some distant point in 
phase with the direct energy from the antenna, 
then reinforcement of the signal strength occurs. 
On the other hand, if the reflected energy arrives 
180° out of phase with the direct energy, a reduc- 
tion or cancellation of signal strength takes place. 

b. Energy reflected from ground will induce a 
voltage into the harmonic antenna. This causes 
a current to flow which combines with the original 
antenna current. The total antenna current then 
will be greater or less than the original antenna 
current, depending on the height of the antenna. 
Consequently, the radiation resistance of the 
harmonic antenna varies, depending on the height 
above ground. In this respect, the behavior of 



the harmonic antenna is the same as the half-wave 
antenna. 

c. Some horizontal radiation patterns at various 
vertical angles above harmonic antennas parallel to 
the ground are shown in figure 127. As the 
vertical angle is reduced — that is, approaching a 
horizontal plane which includes the antenna — the 
pattern resembles those shown previously in 
figure 126. However, as the vertical angle in- 
creases toward the angle of the lobe maximum, the 
patterns become filled in. The nulls in the direc- 
tion of the antenna itself disappear. 

d. The shapes of the patterns are not altered by 
the earth. The effect of the earth is to change the 
relative amplitude of a pattern, which can be seen 
by comparing the 10° pattern with a 15° pattern, 
on A, B, or C. It is possible for the maximum lobe 
which occurs at one vertical angle to be reduced to 
zero, whereas the maximum lobe which occurs at 
another vertical angle can be increased to twice its 
normal free-space value. In order to note the 
effect of the ground on the radiation pattern, it is 
necessary to alter the patterns produced at various 
angles by taking into account the antenna height. 

86. Feeding Long-wire Antennas 

a. Both resonant and nonresonant lines can be 
used to feed long-wire antennas. The same 



146 






general principles apply here as in the half-wave 
antenna. Since a point on the antenna which is a 
current loop becomes a current node on the next 
higher harmonic, a current-fed antenna behaves 
as a true long wire only at odd harmonics of the 
original frequency. Therefore, for operation on 
all harmonics, end feeding is preferred. However, 
with end feeding, unbalanced transmission-line 
currents result and a nonsymmetrical radiation 
pattern is produced. The intensity of the lobes 
in the direction off the feeder end of the long wire 
is reduced, and the intensity of the lobes in the 
direction away from the feeder end is increased. 
When matching sections of line are used with 
nonresonant feeders, it must be realized that these 
operate over only a narrow band of frequencies. 

b. An end-fed long-wire antenna with a reso- 
nant feeder line is shown in figure 128. Operation 
on all harmonic frequencies is possible with this 
arrangement, provided the tuning unit at the 
input end of the resonant line has sufficient 
range to match the input impedance to the trans- 
mitter. Arrangements for using nonresonant 
lines are shown in B and C. In both, quarter- 
wave matching sections are used to match the 
nonresonant line to the long-wire antenna. In B, 
the feeder is tapped on the matching section at 
a point where an impedance match occurs. 
LONG-WIRE ANTENNA 

RESONANT 
LINE 



LONG-WIRE ANTENNA 




•5- MATCHING 
SECTION"* 



NONRESONANT 
LINE 




LONG-WIRE ANTENNA 



-j MATCHING 
SECTION"* 



CURRENT 
r LOOP 



NONRESONANT 
LINE 



L«2\ 




TM 666-131 

Figure 127. Horizontal patterns of harmonic antennas at 
various vertical angles. 



TM 666-132 



Figure 128. Feeding long-wire antennas. 



147 



In C, the feeder is connected to a Q-matching 
section the characteristic impedance of which is 
made equal to the square root of the product of 
the radiation resistance of the long-wire antenna 
and the impedance of the nonresonant line. 

87. Resonant and Nonresonant Antennas 

a. Only resonant antennas have been discussed 
heretofore in this manual. These have standing 
waves of voltage and current distributed along 
their length which are set up by the reflection of 
waves at the ends of the antenna. If one end of 
an antenna is terminated in a resistance that is 
equal to the characteristic impedance of the 
antenna, waves can travel in one direction only. 
As a result, no standing waves are set up. Instead, 
the current and voltage are distributed uniformly 
along the length of the antenna. Such an antenna 
is known as a nonresonant antenna. 

b. The radiation pattern of a nonresonant 
antenna is quite different from the pattern pro- 
duced by a resonant antenna. Consider the 
radiation patterns shown in figure 126. Assume 
that all these resonant antennas are made non- 
resonant by connecting a terminating resistor 
between the right end of each antenna and 
ground. All of the antennas then radiate only 
in the direction of the terminating resistor, or 
toward the right in the figure. The lobes of 
energy to the left are largely attenuated. Conse- 
quently, the major lobe takes the form of a 
single cone of radiation surrounding the antenna 
in the direction of the terminating resistor. The 
antennas are converted from bidirectional types 
(which produce maximum radiation in two di- 
rections) to unidirectional types (which produce 
maximum radiation in only one direction). If a 
radiation pattern were drawn to show the radi- 
ation at the vertical angle at which maximum 
radiation occurs, a single major lobe of radiation 
appears in the direction of the antenna itself 
and toward the terminating resistor. 

c. An important characteristic of a nonreso- 
nant antenna is that it radiates efficiently over a 
very wide frequency range. Therefore, it is not 
necessary to cut the antenna for any exact length 
so long as it is at least several wavelengths. 

88. Beverage or Wave Antennas 

a. Description and Design. 

(1) One type of nonresonant, long- wire an- 



tenna is the Beverage or wave antenna 
which consists of a single wire preferably 
of 2 or more wavelengths, parallel with 
the earth and supported on poles at a 
height of 10 to 20 feet above ground. The 
far end of the wire is connected to ground 
through a noninductive resistor of about 
500 ohms. This resistor must be able 
to dissipate about one-third of the power 
fed into the antenna. This is about the 
characteristic impedance of a single-wire 
transmission line with a ground return. 
A wave antenna is shown in figure 129. 
A reasonably good ground, such as a 
number of ground rods or a counter- 
poise, should be used at both ends of 
the antenna. 

(2) Sometimes two or more antenna wires 
are used in parallel instead of a single 
wire. This reduces the characteristic 
impedance of the antenna and ground- 
return circuit and permits a lower value 
of terminating resistance to be used. 
The input impedance of the antenna is 
reasonably constant with frequency, and 
the antenna may be used over a wide 
frequency range without changing its 
length. 

(3) The wave antenna is directional and is 
used primarily for either transmitting or 
receiving low-frequency signals. Maxi- 
mum reception or radiation is in line with 
the wire and off the terminated end. 
There is a minimum of radiation in the 
opposite direction if the antenna is ter- 
minated properly. The forward lobe 
may be made narrower and the gain in- 
creased by using a longer antenna wire. 
However, if extremely long-wave anten- 
nas are used, the forward gain falls off. 

(4) At frequencies below 800 kilocycles, a 
properly located wave antenna should 
give results equivalent to a vertical an- 
tenna several hundred feet high. One 
particular military wave antenna (fig. 
130) consists of three conductors ar- 
ranged in the form of an equilateral 
triangle 5 feet on a side, erected about 
15 feet above ground on short telephone 
poles, and usually of 2 wavelengths. At 
a frequency of 500 kilocycles, such an 
antenna would be almost 4,000 feet long. 
If ground space limitations prevent the 



148 



DIRECTION OF TRANSMISSION 




TM 5005-18 

Figure 129. Beverage or wave antenna. 



use of such a long antenna, an antenna 
under 1 wavelength can be used. A re- 
duction of forward gain will result under 
these conditions. 
b. Wave Tilt. 

(1) The operation of the wave antenna de- 



pends on a process known as wave tilt. 
When a vertically polarized radio wave 
travels over the surface of an imperfect 
conductor, such as the earth, the wave 
fronts lean forward in the direction of 
propagation. This is caused by the 



jar 

H 






j| TM 5005 -IS 



Figure ISO. Three-wire wave antenna. 



149 



slower propagation constant of the earth. 
The amount of forward tilt depends on 
the frequency of the r-f wave and the 
characteristics of the surface over which 
the wave is traveling. At the lower 
frequencies at which the wave antenna 
is used, the wave tilt is approximately 
proportional to the square root of the 
product of frequency and soil resistivity. 
As the resistance of the surface is in- 
creased, the wave travel along the sur- 
face is reduced and a greater wave tilt 
results. Consequently, over rocky and 
sandy soil, a considerable forward tilt re- 
sults, whereas over salt marshes and sea 
water, almost no tilt occurs. This means 
that wave antennas, which depend on 
wave tilt for proper operation, should 
not be installed over highly conducting 
surfaces, but, instead, should be installed 
only over poor or medium soil. Wave 
antennas also give good results when in- 
stalled over ground which has a perma- 
nent layer of ice (such as permafrost) a 
short distance below the surface, or over 
certain types of ground found in northern 
or polar regions which are very moist in 
the summer and have poor conductivity 
because of lack of mineral content. Ac- 
tually, it is the average ground conduc- 
tivity for a considerable distance below 
the surface that is important rather than 
the character of a thin top-soil layer. 
(2) The wave antenna operates in conjunc- 
tion with ground, so that a vertically 
polarized radio wave is radiated. How- 
ever, because of the forward wave tilt, 
there is a horizontal component of the 
electrical field. The vertical and hori- 
zontal components are not exactly in 
phase, and the resultant polarization of 
the radiated wave, therefore, is elliptical. 
The wave is radiated in the direction of 
the tilt which is off the end of the antenna 
that is terminated in the resistance load. 
c. Transmitting Antennas. When the wave an- 
tenna is used for transmitting, the r-f output of 
the transmitter is connected between the wire con- 
ductor of the antenna and ground (fig. 129). 
Ground can be considered as one conductor of a 
transmission line and the antenna wire as the 
other conductor. As the r-f energy travels down 
the line, the distance traveled by the energy along 

150 



the ground is less than the distance traveled by 
the energy along the wire, because of the lower 
velocity of propagation of the earth wave. This 
effect produces an out-of-phase relationship be- 
tween the wire wave and the ground wave. As a 
result, the wave front is caused to tilt forward 
(fig. 129) and the traveling wave off the end of the 
antenna contains both vertical" and horizontal 
components of wave energy with respect to a path 
of travel parallel to the ground. The radiated 
energy then is considered as elliptically polarized. 
At a distance from the transmitter, however, the 
predominant component of polarization is ver- 
tical, as a result of normal ground-wave propaga- 
tion effects. 

(1) Generally, the forward or desired radia- 
tion increases as the antenna is length- 
ened. As an example of the effect of 
difference in length on efficiency, data 
taken on 1,200-foot and 3,600-foot wave 
antennas at 250 kc and 500 kc erected 
over poor ground are indicated in the 
following chart: 



Length (ft) 


260 kc 


500 kc 


Length 
(wavelength) 


Gain 


Length 
(wavelength) 


Gain 


1,200 

3,600 


0. 306 
. 918 


db 
8 db 


0. 612 

1. 826 


db 
9 db 



Note. Gain of the 1,200-foot antenna is taken 
as db at each frequency. 

(2) Experimental data indicate that a wave 
antenna having a wire length equivalent 
to 2 wavelengths and erected over poor 
ground has a radiation efficiencjr in the 
forward direction equivalent to the radia- 
tion of a quarter-wave vertical antenna. 
A wave antenna having a wire length 
equivalent to 4 electrical wavelengths 
under the same conditions shows a radia- 
tion efficiency in the forward direction 
equivalent to a half-wave vertical an- 
tenna. Experiments in the range of 100 
to 200 kc with wave antennas of 0.6 to 
1.5 wavelengths, compared with stan- 
dard-type flat-top antennas mounted on 
180-foot towers, showed gains of approx- 
imately 10 db for the wave antenna. 
Highest gains were noted with the longer 



wave antennas erected over poor ground. 
(3) The efficiency of the wave antenna in- 
creases rapidly as the height is increased 
from to the range of 12 to 15 feet. 
Above 15 feet, there is little increase in 
efficiency. 

d. Receiving Antennas. Wave antennas also 
are used for receiving and, in this application, 
their performance also depends on wave tilt. 
However, when receiving, the radio waves ap- 
proaching the antenna already are tilted because 
of their propagation over poor soil in the locality 
of the antenna. 

(1) As the tilted wave moves in a direction 
from the terminating resistor toward the 
receiver, energy is induced along both the 
antenna wire and the ground . The effect 
of this induced energy is cumulative, 
since energy from the traveling wave is 
absorbed by the antenna, and a large cur- 
rent is produced at the input to the re- 
ceiver. Actually, the induction of energy 
along the ground is a continuing process 
throughout the entire travel of the wave 
and not only at the antenna location. 
When so regarded, the antenna wire can 
be considered as the medium of extracting 
energy from the space surrounding it, 
and guiding this energy to the receiver 
with the proper phasing with respect to 
the receiver ground, so that a high level 
input is obtained. 

(2) The polar pattern of the antenna is the 
same for transmitting and receiving, with 
maximum antenna gain in a direction 
from terminating resistor to receiver. 
When maximum gain is desired in the 
opposite direction, a special circuit ar- 
rangement can be used. This arrange- 
ment consists of using reflection trans- 
formers at each end of the antenna and 
placing the terminating resistor at the 
receiver location. 

(3) When bidirectional reception is desired, 
the normal antenna circuit is used except 
for omission of the terminating resistor. 

e. Feeding Methods. 

(1) Since the wave antenna is a grounded an- 
tenna with a wide frequency range, it 
usually is fed by means of an unbalanced, 
nonresonant transmission line. The in- 
put impedance of the single-wire antenna 
is approximately 500 ohms, so that the 



characteristic impedance of the line also 
must be 500 ohms. 

(2) The most common feeding arrangement 
is a single-wire transmission line con- 
nected to the end of the antenna. If co- 
axial line is to be used, an impedance- 
matching transformer is inserted between 
the transmission line and the antenna. 

89. V Antenna 

a. General Description. The V antenna con- 
sists of two horizontal, long wires arranged to 
form a V, and fed at the apex with currents of 
opposite polarity. Major lobes from each wire 
combine in such a way that maximum radiation 
occurs in the direction of a line that bisects the 
angle between the two wires. Figure 131 shows a 
V antenna with the individual radiation patterns 
of each of the wires. The shaded lobes produced 
by each individual leg of the V lie in exactly the 
same direction. These lobes combine to form the 
shaded lobes in the resultant pattern. Most of 
the other lobes are more or less attenuated. The 
pattern is bidirectional, and radiation occurs 
along a line that bisects the apex angle in both 
directions. 

b. Design. 

(1) As with other long-wire antennas, the 
greater the leg length the higher the gain 
and .directivity of the antenna. The 
gain of the V antenna is about twice that 
of a single long-wire antenna, since the 
radiation from the lobes of two wires 
combines to produce the radiation pat- 
tern of the V antenna. In actual prac- 
tice, the gain may be even higher than 
this value because of the effects of one 
leg of the V on the other. 

(2) The following chart shows the approxi- 
mate power gains of V antennas for 
various leg lengths, using the optimum 
value of apex angle in all cases. 



Antenna length 
(wavelengths) 


Power gain 


Antenna length 
(wavelengths) 


Power gain 


1 


2. 1 

2. 9 

3. 8 
5. 


6 


8.0 
11. 9 
17.8 


2 


8 


3 


10 


4 







151 



(3) The optimum apex angle for the V an- 
tenna is, ordinarily, twice the angle be- 
tween the lobe of maximum radiation and 
the wire itself when the wire is used as a 
conventional long-wire antenna. Here, 
the lobes of maximum radiation from the 
two long wires making up the V antenna 
are in the same direction so that they 
combine as shown in figure 131. In prac- 
tice, a somewhat smaller angle than this 
value is used when the V-antenna legs 
are shorter than about 3 wavelengths. 
This increases slightly the gain of the 
antenna. 

(4) The following chart shows the optimum 
apex angle for V antennas with equal legs 
of various lengths: 



various lengths. A height, above ground 
of a half-wavelength is assumed. 



Antenna length 
(wavelengths) 


Optimum apex 
angle (degrees) 


Antenna length 
(wavelengths) 


Optimum apex 
angle (degrees) 


1 


90 
70 

58 
50 


6 


40 
35 
33 


2 


8 


3 


10 


4 







(5) When the V antenna is to be operated 
over a wide frequency range, an average 
optimum apex angle should be used. 
Reasonably good results are obtained by 
noting the optimum apex angle for the 
antenna at its lowest operating frequency 
and the angle for its highest operating 
frequency, and then using the average of 
these two values. 

(6) The V antenna does not radiate the major 
portion of its energy along the surface of 
the earth. Instead, the energy is radi- 
ated upward at a certain vertical angle 
in respect to the earth. The size of this 
angle depends on the length of the an- 
tenna legs and the height of the antenna 
above ground. In general, as the an- 
tenna length is increased or as the height 
above ground is increased, the vertical 
angle at which maximum radiation oc- 
curs gradually becomes smaller. The 
vertical angle is measured in respect to 
the horizontal antenna wires. 

(7) The following chart gives the approximate 
value of the vertical angle at which maxi- 
mum radiation occurs for V antennas of 



Antenna length 
(wavelengths; 



1 

2 

3 

4. 



Vertical angle ij Antenna length 
(degrees) j (wavelengths) 



31 |! 6_ 
27 



23 
20 



10 



Verticle angle 
(degrees) 



16 
14 
13 



c. Feeding Methods Balanced lines are used to 
feed the V antenna. Resonant lines are used if a 
wide frequency range is to be covered. Nonreso- 
nant feeders can be used in conjunction with 
quarter-wave matching sections at the apex of the 
V antenna, but only a fairly narrow frequency 
band can be accommodated. 

d. Unidirectional V Antenna. 

(1) The V antenna can be made unidirec- 
tional by making the antenna non- 
resonant. This is accomplished by con- 
necting noninductive resistors of about 
500 ohms between the far end of each 
leg of the V antenna and ground (A of 
fig. 132). The resistors must be able to 
dissipate about one-third the power ap- 
plied to the antenna and must go to a 
good ground. Since no standing waves 
exist on the antenna, the length of the 
legs need not be a multiple of the half- 
wave. In a nonresonant antenna, maxi- 
mum radiation occurs in the direction of 
the terminating resistor. The lobe of 
maximum radiation then is directed 
toward the open mouth of the V antenna, 
whereas the radiation in the opposite 
direction is largely suppressed. 

(2) A unidirectional V antenna is shown in 
figure 132. The exact values for the 
terminating resistors can be found by a 
cut-and-try method in which various 
values of resistance are used until mini- 
mum standing waves appear on the an- 
tenna. The proper value should be in 
the vicinity of 500 ohms. A nonresonant 
open-wire line is used to feed the antenna. 
The unidirectional radiation pattern pro- 
duced by this antenna is shown in B of 
figure 132. 

e. Obtuse-Angle V Antenna. 

(1) If the angle between the legs of the V 
antenna is greater than 90°, the antenna 



152 




RESULTANT PATTERN 
AND DIRECTIVITY 




TM 666-133 



Figure 181. Formation of radiation pattern of V antenna. 




Antenna length 
(wavelengths) 


Angle between 
legs (degrees) 


Antenna length 
(wavelengths) 


Angle between 
legs (degrees) 


1 


90 
110 
122 
130 


6 


140 
145 
147 


2 


8 


3 


10 


4 








(2) 



RADIATION PATTERN 



Figure 188. Unidirectional V antenna. 

is an obtuse-angle V antenna (fig. 133). 
The value of the obtuse angle is obtained 
by subtracting from 180° the value of 
apex angle for the conventional V antenna 
with the same leg length. 
The following chart shows the correct 
angle for obtuse-angle V antennas of 
various leg lengths: 



(3) The obtuse-angle V antenna has the ad- 
vantage of maintaining the same direc- 
tivity over a wide frequency range. This 
is so because when the frequency is 
changed, the major lobe of radiation pro- 
duced by one leg shifts in one direction 
and the major lobe produced by the other 
leg shifts in the opposite direction. Al- 

MAXIMUM 
RADIATION 




TM 666-137 

Figure 188. Obtuse-angle V antenna. 



153 



though a broadened lobe occurs and the 
gain is reduced somewhat, the directivity 
is exactly the same. Obtuse-angle V an- 
tennas frequently are terminated by a 
resistor, however, which makes the an- 
tenna nonresonant and unidirectional. 
Such obtuse-angle V antennas are known 
as half-rhombic antennas. 
(4) The obtuse-angle V is objectionable be- 
cause it requires twice the distance of an 
ordinary V antenna and produces less 
gain. For this reason, this antenna is not 
too popular. 
/. Combination V Antennas. 

(1) A single unterminated V antenna radiates 
energy in two directions that are opposite 
to each other. Combinations of V an tennas 
can be used if it is desired to cover more 
directions. For example, nine 6-wave- 
length antenna wires can be erected radi- 
ally with angles of 40° between them 
like the spokes in a wheel. This forms 
nine V antennas, with all apexes meeting 
at a common point. Each one of the 
radial wires serves as a leg for two ad- 
jacent V antennas. Separate feeder lines 
are used for each V antenna. Any one 
of nine different directions can be covered 
by selecting the proper feeder lines 

(2) The radiated power produced by a V an- 
tenna can be doubled approximately by 
the use of two V antennas operating si- 
multaneously. There are, in general, 
three methods of arranging these two an- 
tennas. First, they can be erected paral- 
lel to each other in such a manner that 
one of the V's is a half-wavelength above 
the other. They now are said io be 
stacked. When the antennas are fed in 
phase, approximately twice the radiated 
power is produced. The vertical angle 
of maximum radiation is reduced. Sec- 
ond, the antennas can be erected in such 
a way that a W is formed. The two V's 
making up the W must be fed in phase. 
Third, the two antennas may be fit the 
same height above ground, but one V is 
located a quarter-wavelength in front of 
the other. When the antennas are fed 
so that their currents are 90° out of phase, 
unidirectional radiation occurs. The 
maximum radiation is in the direction of 
the antenna with the lagging current. 



These are called V beams. The same 
principles apply to combinations of V 
antennas to produce beams as apply to 
combinations of other antennas for the 
same purpose (pars. 94 through 109). 

90. Half-rh ombic Antenna 

a. General Description. The half-rhombic an- 
tenna is a terminated vertical antenna which 
resembles the obtuse-angle V antenna. With the 
obtuse-angle V, a balanced transmission line is used 
and ground does not form part of the radiating 
system. With the half-rhombic antenna, how- 
ever, an unbalanced transmission line is used and 
ground or a counterpoise is utilized. As a result, 
a vertically polarized radio wave is produced. 

b. Directional Characteristics. 

(1) The development of the radiation pattern 
produced by the half-rhombic antenna is 
shown in figure 134. In A, the half- 
rhombic antenna has not been termi- 
nated. Assume that each leg is 2 wave- 
lengths and that the angle between the 
two legs is correct. A transmitter is con- 
nected between the end of the antenna 
and a good ground. A single-wire count- 
erpoise frequently is used which extends 
for the entire projection of the antenna 
length on the ground. Current from the 
transmitter flows toward the untermi- 
nated end of the antenna where it is re- 
flected back along the antenna, as shown 
by the arrows. As a result of this re- 
flection, standing waves are set up on the 
antenna and lobes of radiation appear as 
shown. 

(2) Lobe 2 combines with lobe 5 to produce 
strong forward radiation from left to 
right. Lobe 4 combines with lobe 7 to 
produce strong rear radiation from right 
to left. These lobes exhibit bidirectional 
directivity along the direction of the 
antenna itself. The remaining lobes 
combine in various ways to produce 
several minor lobes in other directions. 
As a transmitting antenna, maximum 
energy is radiated in the directions shown 
by the large two-headed arrow, and as a 
receiving antenna best reception occurs 
.in these same directions. 

(3) When a terminating resistor of about 500 
ohms is connected between the far end of 



154 





the antenna and ground (or counter- 
poise), conditions become different. Cur- 
rent from the transmitter can flow only 
toward the resistor, as shown in B. This 
resistor absorbs any energy that is not 
radiated, and, in so doing, prevents any 
reflection of energy back along the an- 
tenna. As a result of using the termi- 
nating resistor, lobes 3, 4, 7, and 8 dis- 
appear and only the forward lobes re- 
main. Lobes 2 and 5 combine to produce 
intense radiation in the forward direction, 
from left to right, whereas lobes 1 and 6 
produce minor lobes. Consequently, 
when this half-rhombic antenna is used 



for transmission, it is unidirectional, and 
radiates maximum energy along the 
antenna in the direction of the terminat- 
ing resistor, as shown by the large arrow 
in B. 

(4) When the antenna is used for receiving, 
the antenna current will flow from the 
terminating resistor toward the receiver. 
Signals originating from the direction of 
the resistor will produce maximum effect 
on the receiver. Under these conditions, 
all arrows in B would be shown reversed. 
This is in accordance with the usual 
reciprocity of antennas. 

(5) Two important angles are illustrated. 



155 



T, commonly known as the tilt angle, is 
half the apex angle between the two legs 
of the antenna. It is made to have a 
certain value which is determined by the 
leg length. The wave angle, W, is be- 
tween either maximum lobe ot radiation 
and the antenna wire itself. The same 
value for this angle is obtained if meas- 
urement is made between lobe 5 and its 
antenna leg or between lobe 2 and its 
antenna leg. 
c. Design Information. 

(1) Assume that a unidirectional half-rhom- 
bic antenna using a single-wire counter- 
poise is to be designed. It is desirable 
that the legs of the antenna be many 
wavelengths in order to provide maxi- 
mum gain and directivity. For satis- 
factory performance, each leg of a half- 
rhombic antenna must be at least 1 wave- 
length at the lowest, frequency of oper- 
ation. In practice, a leg of at least 2 
wavelengths at the lowest frequency gen- 
erally is used, and some half-rhombic 
antennas use legs of 10 or 12 wave- 
lengths. The leg length usually is limited 
by the size of the available site and the 
directivity required. 

(2) The half-rhombic antenna maintains its 
characteristics over a wide frequency 
range. Frequency ranges of 2 to 1 and 
4 to 1 are common in practice. For 
example, a half-rhombic antenna de- 
digned for a frequency of 10 mc would 
operate satisfactorily to 20 mc and would 
be useful to 40 mc. Depending on the 
amount of change in gain and directivity 
that can be tolerated, an even greater 
frequency range can be accommodated. 
In general, as the frequency is raised, 
greater gain and directivity occur. 

(3) Once the leg length has been decided, it 
is necessary to determine the tilt angle 
required for that leg length. The op- 
timum value of tilt angle is a compromise 
between two sets of conditions. First, 
the tilt angle must have such a value that 
the lobes of maximum radiation from 
both legs are in exactly the same direc- 
tion. This is necessary so that the two 
forward lobes can combine properly to 
produce the unidirectional pattern. This 
value of tilt angle is simply 90° less than 



the wave angle. Second, the tilt angle, 
must have such a value that the radiation 
in the forward lobe of one leg of the half- 
rhombic antenna combines in phase with 
the radiation in the forward lobe of the 
other leg. This usually requires the pro- 
jection of either leg of the half-rhombic 
antenna on the ground to be a half- 
wavelength shorter than the actual length 
of the leg. When this condition is met, 
the tilt angle will be somewhat smaller 
than the first value obtained. In prac- 
tice, the actual size of the tilt angle is a 
compromise between these two values. 
(4) The following chart gives the value of tilt 
angle to be used in half-rhombic antennas 
of various leg lengths. As in most of the 
charts given so far, antenna length is ex- 
pressed in wavelengths. In order to 
convert these lengths to feet, the formula 
given previously can be used. The 
formula is: 

492(7/— 0.05)_ 

frequency (mc) 
where H is the number of half-waves on 
the antenna A somewhat more con- 
venient form is: 

, n , 984(A r -0.025) 

length (ieet)==j — — - 7 — ~ 

frequency (mc) 

where N is the number of full waves on 

the antenna. Because the length is not 

especially critical and the end effect is so 

small for long antennas, the factor 0.025 

may be neglected in practice. If the 

value of the angle between the two legs 

of the half-rhombic antenna is desired, 

simply double the value of the tilt angle. 



length (feet) — ; 



Antenna length 
(wavelenghts) 



1 
2 
3 

4 



Tilt ang!« 
(degrees: 



Antenna length ! Tilt angle 
(wavelengths) j (degrees) 



30 
50 
57 
62 



10 
12 



67 
70 
71 
73 



(5) Once the tilt angle and leg lengths are 
known, the height of the apex of the an- 
tenna above ground and the required 
counterpoise length can be calculated. 
In figure 135, a right triangle is formed 
by one of the legs of the antenna, L, 



156 



the height of the apex above ground, 
H, and one-half the length of the counter- 
poise, % C. The tilt angle, T, is one of 
the angles in the right triangle. The 
ratio of the height to the leg length, 
H/L, is equal to the cosine of the tilt 
angle, cos T. Assuming that the leg is 
2 wavelengths and the tilt angle is 50°, 
as shown in the preceding chart, the 
following relation can be written and 
solved : 



sin T= 



H 



cos T= 



cos 50°==^ 



0.643=^| 

H= 1.286 or 1.3. 

Hence, the height of the apex is 1.3 
wavelengths above ground. To convert 
this height into feet, it i3 necessary only 
to use the same formula ((4) above) for 
converting lengths in wavelengths to feet, 
except that end effect need not be con- 
sidered. Simplified, the formula is: 



length (ft) = 



984 N 



frequency (mc) 



where N is the number of full wave- 
lengths. 




COUNTERPOISE C 

TM 666-139 

Figure 185. Right triangle formed by half-rhombic antenna. 

(6) The ratio between one-half the length of 
of the counterpoise, % C, and the leg 
length, L, is equal to the sine of the tilt 
angle, sin T,. Using this relation, the 
length of the counterpoise ean be cal- 
culated for the example given above as 
follows: 



sin 50 
0.766 



2 



KC= 1.532 or 1.5 

Therefore, the counterpoise required is 3 
wavelengths. 

(7) The following chart gives the height of 
the apex and the length of the counter- 
poise required, both in terms of wave- 
lengths at the operating frequency, for 
half-rhombic antennas of various leg 
lengths. 



Leg length (wavelength) 


Apex height 
(wavelengths) 


Counterpoise 
length (wave- 
lengths) 


1 


0. 87 
1.3 

1. 6 
1.9 
2.3 
2.7 
3.3 
3. 5 


1 
3 
5 
7 

n 

15 
19 
23 


2 


3 


4 


6 


8 


io 


12... 





d. Practical Antennas. 

(1) The factor that most frequently limits 
the size of the half-rhombic antenna is 
the height of the apex above ground. 
If a very tall support is available for the 
apex, a large antenna can be erected. 
It is necessary that ? the single support 
required, be made of wood, or other non- 
conductor, so that the operation of the 
antenna is not affected. Steel masts, or 
wooden masts using metal guy wires, 
should not be used. 

(2) The typical military half-rhombic an- 
tenna shown in figure 136 consists of 
a 100-foot antenna wire erected over a 
single 30-foot wooden mast (supported 
by three rope guys) and an 85-foot 
counterpoise wire laid along the ground. 
The antenna and counterpoise are termi- 
nated in a 500-ohm resistor contained 



157 



Figure 186. Typical military half-rhombic antenna. 



in a small terminal box at the far end 
of the antenna. 

(3) The antenna shown can be used with 
low-power transmitters or receivers oper- 
ating at frequencies from 30 to 70 mc 
and equipped with either au r-f output 
impedance of 500 olims or a suitable 
antenna-matching network At 30 mc, 
the leg is 1% wavelengths, .and at 70 
mc it is 3% wavelengths. A power gain 
of 4 or 5 occurs at the lowest frequency, 
and a power gain of about JO occurs at 
the highest operating frequency. 

(4) Since the transmitter or receiver used 
with the half-rhombic antenna generally 
is located at the end of the antenna, 
direct connections can be made to the 
antenna. If a transmission line must 
be used between the antenna and the 
radio set, a two-wire line with a char- 
acteristic impedance of 500 ohms can 
be used 

(5) A large half-rhombic antenna designed 
for frequencies from 3 to mc, has a 
ground-projected length of 625 feet and 
an apex height of 225 feet. The antenna 
is supported by a hydrogen-filled balloon 
in low winds or by a kite in high winds. 
A ballon- or kite-supported half-rhombic 
antenna, designed for frequencies of 1 



to 8 mc, has a ground-projected length 
of 1,600 feet and an apex height of 560 

feet, 

91. Rhombic Antenna 

a. General Description. 

(1) The highest development of the long- 
wire antenna is the rhombic antenna 
(fig. 137). 1 1 consists of four conductors 
joined to form a rhombus, or diamond. 
Ail sides of the antenna have the same 
length and the opposite corner angles 
are equal. The antenna can be con- 
sidered as being made up of two V 
antennas placed end to end and termi- 
nated by a noninductive resistor to 
produce a unidirectional pattern. A 
rhombic antenna can be made of two 
obtuse-angle V antennas which are 
placed side by side, erected in a hori- 
zontal plane, and so terminated as to 
be made nonresonant and unidirectional. 

(2) In common with previous nonresonant 
antennas, the rhombic antenna radiates 
best in the direction of the terminating 
resistor and receives best from the direc- 
tion of the resistor. Maximum radiation 
does not occur in the same direction as 
the plane of the antenna, that is, hori- 




TOP VIEW 



\_ WAVE ANGLE 
W 



TRANSMISSION 
LINE 




TM •••-Ml 



Figure 1S7. Basic rhombic antenna. 



zontally. Instead, it occurs at some 
vertical angle above the horizontal plane, 
as shown by the wave angle, W. The 
tilt angle, T, is one-half the angle between 
the two legs making up one side of the 
antenna. 

b. Advantages. The rhombic antenna is used 
widely for long-distance high-frequency transmis- 
sion and reception, for reasons explained below. 
It is one of the most popular fixed-station anten- 
nas, being very useful in point-to-point work. 

(1) The rhombic antenna is useful over a 
wide frequency range, a range of 2 to 1 
being covered easily with excellent re- 
sults. Although it is true that some 
changes in gain, directivity, and charac- 
teristic impedance do occur with change 
in operating frequency, these changes are 
small enough to be neglected. A fre- 
quency range of 4 to 1 can be covered by 
a typical rhombic antenna with good 



results, and standard military rhombics 
cover a frequency range of 5 to 1 or 6 to 
1 satisfactorily. 

(2) Another advantage of the rhombic anten- 
na is that it is much easier to construct 
and maintain than other antennas of 
comparable gain and directivity. Only 
four supporting poles of common heights 
from 50 to 75 feet are needed for the 
antenna, which has a simple form, being 
made up of four straight lengths of wire. 

(3) The rhombic antenna also has the advan- 
tage of being noncritical so far as opera- 
tion and adjustment are concerned. 
This follows from the broad frequency 
characteristics of the antenna. 

(4) Still another advantage is that the volt- 
ages present on the antenna are much 
lower than those that would be produced 
by the same input power on a resonant 
antenna. This is particularly important 



159 



when high transmitter powers are used or 
\vhen high-altitude operation is required. 
The lower voltages mean less possibility 
of corona loss. 
Disadvantages. 

(1) The rhombic antenna is not without its 
disadvantages, probably the principal 
one being that a fairly large antenna site 
is required for its erection. Each leg is 
made at least 1 or 2 wavelengths at the 
lowest operating frequency, and when 
increased gain and directivity are re- 
quired, legs of from 8 to 12 wavelengths 
are used. Such requirements mean that 
high-frequency rhombic antennas have 
leg lengths of several hundred feet, and 
they are used then only when a large plot 
of land is available. 

(2) Another disadvantage is that the hori- 
zontal and vertical patterns depend on 
each other. If a rhombic antenna is made 
to have a narrow horizontal beam, the 
beam is also lower in the vertical direc- 
tion. Therefore, it is impossible to 
obtain high vertical-angle radiation ex- 
cept with a very broad horizontal pat- 
tern and low gain. Rhombic antennas 
are used, however, for long-distance sky- 
wave coverage at the high frequencies. 
Under these conditions, low vertical 
angles of radiation (less than 20°) are 
desirable. With the rhombic antenna, 




INDIVIDUAL RADIATION 
PATTERNS 



Figure 138. Formation o, 

160 



a considerable amount of the input 
power is dissipated uselessly in the ter- 
minating resistor. However, this resistor 
is required in order to make the antenna 
unidirectional, and the great gain of the 
antenna more than makes up for this 
loss. 

d. Operation. 

(1) Figure 138 shows the individual radia- 
tion patterns produced by the four legs 
of the rhombic antenna and the resultant 
radiation pattern. If the tilt angle, T, 
is properly chosen for the length of the 
legs used, the shaded lobes all add to- 
gether to form an intense forward lobe 
in the direction of the terminating re- 
sistor. The principle of operation is the 
same as for the V and the half-rhombic 
antennas. 

(2) Practically all rhombic antennas are 
erected in the horizontal plane. The 
two sides of the antenna are fed with 
currents of opposite polarity. As a 
result, the vertical electric field com- 
ponent of the radiated energy produced 
by one side of the antenna is largely can- 
celled by an equal and opposite electric 
field produced by the other side of the 
antenna. Lines of electric force are pro- 
duced from one side of the antenna to the 
other. Therefore, the polarization of the 



^^^^^^^^^^^^^^ 



RESULTANT RADIATION 
PATTERN 



B 

TM 66S-142 

rhombic antenna beam. 



radiated field produced by a horizontal 
rhombic antenna is mainly horizontal. 
(3) Very small rhombic antennas can be used 
at the very-high or ultrahigh frequencies. 
These antennas are erected in a vertical 
plane and a vertically polarized wave is 
radiated. Since the greatest percentage 
of rhombic antennas are used at high 
frequencies where the lengths of the legs 
are several hundred feet, most rhombics 
are horizontal. Therefore, the horizon- 
tal polarization is most common. 
e. Directivity and Gain. 

(1) Typical radiation patterns produced by 
rhombic antennas having various leg 
lengths are shown in figure 139. These 
radiation patterns clearly show an in- 
crease in gain and a reduction in the 
beam width of the main lobe as the 
lengths of the legs are increased. The 
wave angle, W, also is reduced as the 
leg lengths are increased. Information 
concerning the exact dimensions for 
optimum output from a rhombic antenna 
is given in paragraph 91i (4). 

(2) The gain of the rhombic antenna for a 
given leg length is considerably greater 
than for any of the other long-wire 
antennas discussed previously. The ap- 
proximate power gains of rhombic an- 
tennas of various leg lengths are shown 
in the following chart, which takes into 



account the power lost in the terminating 
resistor. 




Leg kngth 
(wavelengths) 


Power gain 


Leg length 
(wavelengths) 


Power gain 


1 


2. 5 
5. 4 
8. 3 
11. 2 


6 


17.0 
22. 4 
28. 2 


2 


8 


3 


10 


4 





/. Terminating Devices. 

(1) To operate properly, the rhombic antenna 
must be terminated correctly by correct 
value of resistance, which will make it 
unidirectional and nonresonant. The 
input impedance of the rhombic antenna 
then remains constant over a wide fre- 
quency range, and antenna coupling 
circuits need not be readjusted when the 
frequency applied to the antenna is 
changed. The proper value for the 
termination is about 800 ohms. When 
this is used at the far end of the antenna, 
the input impedance of the antenna is 
approximately 700 to 800 ohms. Thus, 
the terminating resistor is slightly higher 
in value than the input impedance of the 
antenna, because of the loss of energy by 
radiation as the traveling wave from the 
transmitter moves toward the terminat- 
ing resistor. 





LENGTH OF LEG - 2> 



LENGTH OF LEG = 3> 



LENGTH OF LEG = A\ 




Figure 139. Radiation patterns produced by various rhombic, antennas. 




TM 666-145 



161 



(2) The termination used with the rhombic 
antenna must be a pure resistance at all 
frequencies at which the antenna is to 
operate. If any reactance is associated 
with the termination, some reflection of 
energy occurs and standing waves are set 
up on the antenna, causing variations in 
characteristics and radiation pattern 
when the frequency applied to the an- 
tenna is changed. 

(3) The terminating resistance must dissipate 
a little less than one-half the power ap- 
plied to the input terminals of the an- 
tenna. For transmitter powers up to 
1,000 watts, noninductive carbon re- 
sistors generally are used. These re- 
sistors are available in power ratings up 
to approximately 100 or 200 watts each. 
Frequently, several resistors are paralled 
to provide for adequate power rating. 
The total rated wattage of the resistors 
should equal one-half the transmitter 
wattage. For example, five 150-watt, 
3,000-ohm carbon resistors connected 
in parallel provide a 600-ohm load (re- 
quired for special types of rhombics) 
which will handle a transmitter power of 
1,000 watts with a 50-percent safety 
factor. To reduce capacitance in the 
terminating load, several resistors some- 



times are connected in series, and t 
capacitances across each resistor thei. 
are in series so that a reduction in ter- 
mination capacitance results. The ter- 
minating resistors frequently are mount- 
ed within a weatherproof ed wooden box 
located atop the pole which supports the 
terminated end of the antenna. Con- 
necting leads to the terminating re- 
sistors are made as short as possible to 
minimize the amount of added reactance. 

(4) The insulators used with the rhombic 
antenna, and the supporting wires and 
fittings, sometimes introduce enough 
reactance to require precise adjustment 
of the terminating load to balance it out. 
It is more convenient, then, to mount 
the terminating resistors in a box that 
is near the ground rather than at the top 
of a pole. When this is done, the far 
end of the rhombic antenna is connected 
to the terminating resistors by means of 
an 800-ohm nonresonant transmission 
line. 

(5) For powers in excess of 1,000 watts, car- 
bon resistors are not available that will 
dissipate the necessary power. Lengths 
of transmission line constructed to have 
the proper impedance value, and made 
of wire that is a poor conductor, are used. 




Most of these dissipation lines are made 
of #14 AWG solid, annealed, stainless- 
steel wire, and take two general forms: 

(a) One, a two-wire stainless-steel trans- 
mission line, is spaced properly to pro- 
vide a correct termination for the 
rhombic antenna. The spacing is 
uniform along the entire length of the 
line. If sufficient length is provided, 
the energy is so attenuated by the high 
resistance of the wire that the far end 
can be grounded directly for lightning 
protection. One such dissipation line 
is 1,000 feet long and is run back and 
fourth four times between supporting 
poles, 250 feet apart. The entire dis- 
sipation line is mounted beneath the 
rhombic antenna which it terminates. 

(6) A more common dissipation line which 
requires less than one-third as much 
steel wire is shown in figure 140. This 
line is used with standard military 
rhombic antennas. 
(6) The dissipation line includes, all in- one 
length, the downlead from the end of the 



rhombic antenna. The downlead portion 
is made up as a two-wire line with each 
wire being made of two strands of the 
steel wire twisted together. The spacing 
between the two wires is 12 inches so that 
the characteristic impedance produced 
is about 650 ohms. The downlead be- 
comes part of the horizontal portion of 
the dissipation line by a right-angle bend, 
and at this point a modified exponential 
line begins (fig. 141). 
(7) The two-wire downlead is transformed 
into a four-wire dissipation line without 
the necessity of joining or splicing. The 
12-inch spacing starts diminishing and 
the two strands making up each line of 
the two-wire line now become separate 
spaced lines. In this manner, in a line 
length of 62.5 feet, the 12-inch spacing 
tapers down to a 5.5-inch spacing as the 
side members spread apart to 1.3 inches 
at the dissipation-line spreader insulator. 
From this point on, in a line length of an 
additional 62.5 feet, the 5.5-inch spacing 
tapers down to 1.3 inches, whereas the 




TM 666-145 



Figure HI. Dissipation line detail showing terminating assembly. 



1*3 




TM 666-146 

Figure H2. Dissipation line detail showing terminating assembly. 



side members remain spaced 1.3 inches 
apart. Then, the line continues as a 
1.3-inch, square-spaced, four-wire line. 
The modified exponential portion of the 
dissipation line transforms the approxi- 
mately 650-ohm impedance of the doA'n- 
lead to about 200 ohms. 
(8) The equally spaced four-wire portion of 
the dissipation line is fundamentally two 
400-ohm lines in parallel, one terminated 
in an open circuit and the other ter- 
minated in a short circuit. Such an 
arrangement improves the electrical 
balance and symmetry of the termination. 
The short-circuited end can be connected 
to ground (fig. 142) to provide for 
lightning protection. The three small 
pulleyr shown are used to equalize the 
tension on the individual wires making 
up the line. A wire rope at the end of 
the assembly is passed through the large 
pulley and made fast to a concrete 
weight which maintains tension on the 
line, preventing excessive sagging. 



g. Checking Termination. 

(1) Unless the correct value of termination 
for the rhombic antenna is used, unde- 
sired resonance effects occur. This 
makes the antenna coupling critical, and 
adjustments must be made when the 
applied frequency is changed. Lack of 
symmetry from improper termination 
causes an undesirable shift in the 
directivity pattern of the antenna, and 
the resultant unbalanced reactance may 
introduce undesirable resonance effects. 

(2) In checking the rhombic antenna for 
proper termination, advantage is taken of 
the fact that a properly terminated an- 
tenna has an input impedance that is 
purely resistive. With a perfectly bal- 
anced antenna having the correct value of 
terminating resistance, no reflection of 
energy occurs on the antenna and no re- 
actance appears at the input terminals. 
Therefore, if a balanced oscillator is con- 
nected to the input terminals of a prop- 
erly terminated rhombic antenna, no 



164 



change in oscillator frequency should 
occur. 

(3) The balanced (push-pull) oscillator used 
should have a high L-C ratio-tank circuit 
so that small values of reactance, which 
are usually capacitive with improperly 
terminated rhombic antennas, may have 
considerable effect on the oscillator fre- 
quency. To insure that the oscillator is 
not too heavily loaded, the correct tap 
points on the oscillator tank coil can be 
predetermined with an 800-ohm nonin- 
ductive resistor as load. 

(4) The oscillator first is set at a frequency 
within the operating range of the rhombic 
antenna. The oscillator frequency then 
is carefully measured with a heterodyne 
frequency meter or with a stable com- 
munication receiver having a beat-fre- 
quency oscillator, and the antenna is 
connected to the oscillator. If the rhom- 
bic antenna is perfectly terminated, the 
frequency meter or the receiver should 
indicate no change in oscillator frequency. 
In actual practice, since a perfect termi- 
nation is difficult to produce, a very small 
frequency shift can be tolerated. A 
possibility exists in which little or no 
change in oscillator frequency occurs, 
with the antenna not properly terminated. 
This happens if the antenna is resonant 
at the particular frequency used for the 
check. To eliminate this possibility, 
the frequency shift should be measured 
at a different oscillator frequency. If 
considerable frequency shift occurs when 
the antenna is connected to the oscillator, 
a recheck should be done with the value of 
the terminating resistor changed about 5 
or 10 percent, and this procedure should 
be continued until a resistance value is 
found that produces the least effect on 
the oscillator frequency when the antenna 
is connected. When a rhombic antenna 
is to be used over a wide frequency range, 
this procedure should be repeated for 
several frequencies within the desired 
frequency range. Then an average value 
for the proper terminating resistance can 
be used. 

(5) To use the procedure explained above, it 
is necessary that the test oscillator be 
battery-operated and portable, so that 



the oscillator can be carried up the pole 
which supports the input end of the 
antenna to permit direct connections at 
the input terminals. If the oscillator 
must be used on the ground below the 
rhombic antenna, a length of two-wire 
line is used to connect it to the input 
terminals of the antenna. This line 
must be a half-wavelength at the fre- 
quency at which the rhombic termination 
is to be checked. The frequency of the 
oscillator should be adjusted so that 
there is no change in oscillator frequency 
when the connecting line is connected or 
disconnected from the oscillator. Sev- 
eral separate half-wave connecting lines 
must be used to check the antenna ter- 
mination at several frequencies within 
the operating range. 
h. Design Information. 

(1) In designing a rhombic antenna, the 
first consideration is usually a determi- 
nation of the wave angle, W, needed to 
cover the required distance when a given 
frequency is used at a certain time of the 
day. Such information can be obtained 
from charts and information given in 
TM 11-499, Radio Propagation Hand- 
book. 

(2) The following chart shows some typical 
wave angles required to provide sky- 
wave communication over various great- 
circle distances. The wave angles used 
are invariably less than 30°. 



Great-circle distance (miles) 


Wave angle (degrees) 


1-hop E 
transmission 


l-hop F 
transmission 


2-hopF 
transmission 


250 


25 
13 
7 
4 






500.. 






750 


24 
17 
9 
4 




1,000 




1,500 


25 
17 
12 


2,000 




2,500 











(3) For a given vertical wave angle, a rhombic 
antenna produces maximum power out- 
put when its leg length, L, tilt angle, T, 
and height above ground, H, have 
certain definite values. These values 
are all interdependent and any change 



165 



from the optimum value results in a 
reduction in power at the desired wave 
angle. 

(4) The following chart gives the proper 
values for these factors at various wave 
angles. Wave angles less than 10° are 
not shown. This does not mean that 
such angles are not required but rather 
that the rhombic antenna designed for 
maximum output at such low angles is 
prohibitively large. 



Wave angle (degrees) 



10 
14 
18 
22 
26 
30 



Optimum 
tilt angle 
(degrees) 



80 
76 
72 
68 
64 
60 



Optimum 
leg length 
(wavelengths) 



17. 
8. 5 
5. 3 
3. 7 
2. 7 
2. 



Optimum 
height above 

ground 
(wavelengths) 



1. 45 
1. 04 
. 81 
. 67 
. 57 
. 50 



(5) Frequently, a sufficiently large antenna 
site is not available for the erection of a 
rhombic antenna of proper size to pro- 
duce maximum output. For example, 
according to the chart above, a leg of 
17 wavelengths is required for a wave 
angle of 10°. If such an antenna is to 
operate on a frequency of 8 .me, for 
example, each leg would have to be 
over 2,000 feet long. This antenna 
would require over a mile and a half 
of antenna wire for its construction, and 
the installation would prove difficult. 
Therefore, rhombic antenna dimensions 
are chosen which represent a compro- 
mise design. 

(6) When a rhombic antenna is designed 
with leg length definitely limited, the 
gain of the antenna is less than if the 
dimensions shown in the preceding chart 
are used. All of the advantages of the 
antenna given previously still apply, 
however, and rhombic antennas of a 
compromise design are used widely. 

(7) The following chart shows the dimensions 

to be used when constructing a rhombic 
antenna limited to legs of 2 wavelengths. 



Wave angle (degrees) 


Tilt angle 
(degrees) 


Height above 

ground 
(wavelengths) 


5 


52 

52. 5 

54 

55 

57. 5 
60 


3. 00 
1. 45 
1. 00 
. 75 
. 60 
. 50 


10 


15 


20 


25 


30 





(8) When the limit is 3 wavelengths, the 
dimensions in the following chart apply. 
The required heights above ground are 
the same as given in the previous table 
for similar wave angles. 



Wave angle 
(degrees) 


Tilt angle 
(degrees) 


Wave angle 
(degrees) 


Tilt angle 
(degrees) 


5 


59 
60 
62 


20 


63. 5 
65 


10 


25 _ 

30 


15 









A tilt angle is not given when a wave angle of 30° is required. No com- 
promise in design is needed, since a rhombic antenna with leg limited to only 
2 wavelengths and with a tilt angle of 60° can be used to produce maximum 
output. 

(9) When the leg is limited to 4 wavelengths, 
the dimensions given in the following 
chart apply. Here again the required 
heights above ground are the same as 
those given in the previous charts for 
similar wave angles. Where no tilt 
angle is given, no compromise in dimen- 
sions is required. 



Wave angle 
(degrees) 


Tilt angle 

(degrees) 


Wave angle 
(degrees) 


Tilt angle 
(degrees) 


5 


63. 5 

64. 5 
66. 5 


20 


68. 5 


10 


25 


15 


30 











i. Standard Designs. 

(1) Most rhombic antennas used for military 
applications are based on certain stand- 
ardized dimensions which make satisfac- 
tory operation possible over a frequency 
range of from 4 to 22 mc. This range 
includes the frequencies that commonly 
are used for long-distance point-to-point 



166 



sky-wave communication between fixed 
stations. 

(2) -The seven standard sizes used are desig- 
nated as rhombic antennas A through 
Q, inclusive. Antenna A is the largest 
rhombic and it is used when communica- 
tion is required between points over 
3,000 miles apart. The leg of this 
antenna is about 1% wavelengths at the 
lowest operating frequency (4 mc) and 
about 8% wavelengths at the highest 
operating frequency (22 mc). Antenna 
0, the smallest rhombic, is used when 
communication is required between points 
that are from 200 to 400 miles apart. 
The leg of this antenna is somewhat less 
than 1 wavelength at the lowest operating 
frequency and about 5 wavelengths at the 
highest operating frequency. Rhombic 



antennas B through F inclusive have 
intermediate ranges and leg lengths. 
(3) Complete kits are available which include 
all necessary material for the construction 
of standard military rhombic antennas. 
The four large poles or metal supports 
used are designated as side poles, front 
pole, and rear pole in the isometric view 
of the rhombic antenna (fig. 143). 
Terminating resistors, or a dissipation 
line, are connected at the corner of the 
antenna supported by the front pole. 
The transmission line which connects 
the transmitter or receiver to the antenna 
is attached to the corner supported by 
the rear pole. As shown in the plan 
view, the side poles and the front pole 
are located 3 feet from the corners of the 
antenna which they support, and the 



TERMINATING 



FRONT 



SIDE 
POLE 



TRANSMISSION 

LINE TO 
TRANSMITTER 
OR 
RECEIVER 



REAR 
POLE 




FRONT 
POLE 




PLAN VIEW 

Figure lJfi. Standard military rhombic antennas. 



TM «M-I4T 



1*7 



rear pole is located 8 inches from the cor- 
ner which it supports. These distances 
permit the installation of strain insulators 
and supporting harnesses which attach 
the antenna to the poles. 
(4) The chart given below indicates the essen- 
tial dimensions used in the seven standard 
rhombic antennas, along with the useful 
ranges of these antennas. The various 
letters designate dimensions that are indi- 
cated in the plan view of figure 143. All 
linear dimensions are given in feet. L 
refers to the leg length measured from 
corner to corner. This includes the length 
of the strain insulators used at the front 



and rear poles. T refers to the size of the 
tilt angle in degrees, as previously defined. 
H is the average height of the antenna 
above average ground level. The harness 
which ties the antenna corners to the 
poles usually is attached to the- pole? at a 
height 1 to 2 feet above H. W is the pole 
spacing along the minor axis of the an- 
tenna. X is the distance between the 
rear pole and the point at which the axes 
cross as measured along the major axis, 
and Y is the distance between the front 
pole and the point at which the axes cross 
as measured along the major axis. 







Range 


L 


T 


H 


w 


X 


Y 




Type 


(miles) 


(feet) 


(degrees) 


(feet) 


(feet) 


(feet) 


(feet) 


A 




3, 000 + 


375 


70 


65 


262. 4 


352. 7 


355 


B 




2, 000-3, 000 


350 


70 


60 


245. 6 


329. 5 


331. 8 


C 




1, 500-2, 000 


315 


70 


57 


221. 6 


296. 7 


299 


D 




1, 000-1, 500 


290 


67. 5 


55 


228 


268. 7 


271 


E 




60O-1, 000 


270 


65 


53 


234 


245. 4 


247. 7 


F 




400-600 


245 


62. 5 


51 


232 


219 


221. 3 


G 




200-400 


225 


60 


50 


231 


195. 7 


198 



j. Multiwire Rhombics. 

(1) A rhombic antenna will improve in per- 
formance if more than a single conductor 
is used to form each leg. The most com- 
mon of the multiwire rhombics is the 
three-wire type (fig. 144). The spacing 
between the three wires forming this an- 
tenna increases continuously as the side 
poles are approached. At this point, a 
separation of 6 feet exists between 
adjacent conductors. 

(2) When this type is used, the capacitance 
of the antenna per unit length increases 
as the separation between the two sides 
increases. Along the minor axis where 
the two sides of the antenna are spread 
farthest apart, the three conductors have 
their maximum capacitance, and the 
characteristic impedance of the antenna, 
therefore, does not vary along its length 
as it does in a single conductor. 

(3) Two advantages occur with the multi- 
wire rhombic. First, the input im- 
pedance of the antenna is held at a more 
constant value over a given range of 
frequencies. Second, the value of input 



impedance is reduced somewhat so that 
a better impedance match to ordinary 
two-wire line is possible. An ordinary 
single-wire rhombic antenna designed to 
operate over a frequency range from 4 
to 22 mc may have an input resistance of 
850 ohms at 4 mc, 700 ohms at 14 mc, 
and 625 ohms at 22 mc. If a three- 
wire rhombic is used instead, it will have 
an input resistance of 600 ohms plus or 
minus 50 ohms over this same frequency 
range. At the same time, a conventional 
600-ohm two- wire line can provide prac- 
tically an ideal impedance match. In 
addition, the three-wire rhombic has a 
slight gain (about i db) over the single- 
wire type. 
k. Methods oj Feeding, 

(1) The most common method of feeding a 
rhombic antenna is by means of a non- 
resonant two-wire line. With this line, 
the wide frequency range of the rhombic 
antenna is not restricted by transmission- 
line limitations. 

(2) When a single-wire rhombic is used, the 
transmission-line impedance required is 



168 




169 



approximately 700 to 800 ohms. A two- 
wire line having such a high value of 
characteristic impedance would have a 
fairly wide spacing. As a result of the 
wide spacing, considerable radiation loss 
occurs from the line. To avoid this loss, 
two alternatives are possible. First, a long 
tapering section of transmission can be 
used as an impedance-matching section. 
The spacing between the two conductors 
is greatest at the end of the section con- 
nected to the antenna; therefore the im- 
pedance is high enough to match the 
antenna. The spacing at the other end 
of the tapered section is least, resulting 
in a low value of impedance. If the 
tapered section, known as an exponential 
matching section, is designed to have an 
impedance of 600 ohms at its small end, 
ordinary 600-ohm two-wire line can be 
used between the low-impedance end of 
the matching section and the transmitter 
or receiver that is to be connected to the 
antenna. If the exponential matching 
section is designed to have a low im- 
pedance of 200 to 300 ohms, ordinary 
four-wire line can be used The second 
alternative involves the use of 600-ohm 
two-wire line directly connected to the 
end of the rhombic antenna. Since the 
standing-wave ratio is so small, the 
added loss resulting from a slight mis- 
match may be low enough to neglect in 
all cases except those in which peak 
efficiencies are required. In this method, 
the coupling to the transmitter may 
have to be readjusted slightly as the 
frequency is changed. 

(3) When a three-wire rhombic antenna is 
used, the transmission line impedance 
required is 600 ohms. It is then neces- 
sary only to connect an ordinary two- 
wire open line having a 600-ohm imped- 
ance between the antenna and the trans- 
mitter. 
I. Lobe Alinement. 

(1) When a rhombic antenna is designed 
according to the information given in the 
previous charts, it produces maximum 
output at the supposedly desired vertical 
wave angle. If the vertical radiation 
pattern of such an antenna is examined, 
it will be noted that the maximum output 



power actually is produced at a vertical 
angle which is a few degrees less than the 
desired wave angle. The peak of the 
maximum radiation lobe falls slightly be- 
low the wave angle for which the antenna 
is designed, as illustrated by lobe A in 
the vertical pattern of figure 145. 



1TO 2db-H 




HORIZONTAL PATTERN 

TM (66-14* 

Figure 1^5. Lobe alinemenl patterns. 



(2) Since no other combination of leg length, 
tilt angle, and height above ground will 
produce a rhombic antenna having a 
greater radiation at the desired wave 
angle, changing any of these factors 
would serve to reduce the gain of the 
antenna at the desired wavelength. It 
is entirely normal then for the peak of the 
lobe to occur at a vertical angle that is a 
few degrees less than the wave angle for 
which the antenna is designed. 

(3) When a rhombic antenna is designed so 
that the peak of the lobe occurs just at 
the desired wave angle, the radiation 
pattern produced is illustrated by lobe 
B. This may be desirable if a somewhat 
sharper vertical radiation pattern is re- 
quired along with a somewhat broader 
horizontal radiation pattern. Some 
rhombic receiving antennas are designed 
that produce this type of pattern in order 
to minimize noise originating near the 
ground level, and to improve the signal- 
to-noise ratio of the antenna. 

(4) When the pattern illustrated by lobe B 
is required, the lobe alinement method is 
used in designing the antenna. The 
dimensions given in the previous charts 



170 



can be used in the lobe alinement design, 
except that the length of each leg of the 
antenna is shortened to three-quarters of 
the leg length required to produce maxi- 
mum output. When the lobe alinement 
method is used, there is a reduction in 
gain of about 1 or 2 db at the desired 
wave angle. 
m. Oround Effects. 

(1) The effect of ground reflections on the 
radiation pattern of the rhombic antenna 
is exactly the same as with any horizontal 
antenna. There is one optimum height 
above ground at which reflection of 
radiated energy from the ground acts to 
produce maximum radiation at a given 
wave angle. This is the height which 
has been given in the preceding charts 
that show rhombic antenna dimensions. 

(2) As the height of the antenna above 
ground is increased, the wave angle at 
which maximum radiation occurs is 
reduced (fig. 146). 




10 15 20 25 
WAVE ANGLE (IN DEGREES) 

TM 666-00 

Figure 1^6. Graph showing variation in wave angle for 
different antenna heights. 

(3) The ground above which the rhombic 
antenna is installed should be flat and 
free from obstructions, and it should 
have a uniform conductivity. If these 
precautions are not observed, the actual 
required height of the antenna will be 
difficult to determine because the actual 
height required above ground may be 



considerably different for a given wave 
angle than is shown in the graph. If a 
rhombic antenna must be installed above 
uniformly sloping ground, it is possible 
to design the antenna in such a way that 
it compensates for the ground slope. 
For example, assume that a rhombic 
antenna is to be designed with a wave 
angle of 15°, to be erected over ground 
which has a 5° downward slope toward 
the front pole of the antenna. The 
required wave angle can be produced by 
designing the rhombic to produce a 20° 
wave angle and mounting the antenna 
parallel to the ground. 

n. Resonant Rhpmbics. 

(1) All of the rhombic antennas discussed so 
far have been terminated properly so 
that a unidirectional radiation pattern is 
produced. In practically every case 
where a rhombic antenna is referred to, 
this is the type used. If, however, the 
terminating resistance is removed from 
the antenna, a resonant rhombic is 
produced. 

(2) Since the resonant rhombic antenna is 
unterminated, energy traveling from the 
transmission line to the far end of the 
antenna is reflected back, and standing 
waves of voltage and current are set up 
along the conductors making up the 
antenna. 

(3) The input impedance of such an antenna 
is no longer in the vicinity of 800 ohms, 
but instead, is a much higher value. 
This means that it must be fed by resonant 
transmission lines, or that impedance- 
matching sections are necessary. As a 
result, the antenna cannot be used over a 
wide frequency range unless extensive 
retiming of the coupling system occurs 
along with considerable readjustment of 
the impedance-matching sections. 

(4) The radiation pattern of a resonant 
rhombic antenna is bidirectional along 
the major axis of the antenna. If it is 
desired to transmit or receive in only one 
direction, the resonant rhombic should 
not be used. This ma} 7 interfere with 
other communication when the antenna 
is used for reception, since undesired 
signals and noise will be received from 



171 



the direction that is opposite to the 
desired direction of reception. This low- 
ers the signal-to-noise ratio. 

(5) In addition to the undesired major iobe 
that is produced to the rear of the anten- 
na, the radiation pattern of the resonant 
rhombic antenna changes considerably 
when the applied frequency is changed. 
For example, a certain lobe pattern is 
produced when the antenna leg is 2 
wavelengths, and another pattern is 
produced when the frequency is raised 
so that the leg is 2){ wavelengths. 

(6) The gain of a resonant rhombic antenna is 
less than that of a nonresonant terminated 
rhombic. In the terminated antenna, 
less than one-half the input power is dis- 
sipated by the terminating resistance so 
that more than one-half the input power 
is radiated into space. In the unterm- 
inated or resonant rhombic antenna, the 
radiation divides equally; one-half the 
available input power is radiated toward 
the front and one-balf toward the rear. 

92. Summary 

a. Antenna directivity is the ability of an 
antenna to radiate or receive energy better in 
some directions than in others. 

6. An antenna is said to have gain when it pro- 
duces a greater field strength at a given distant 
receiving point than does a standard half-wave 
antenna. The standard antenna is assumed to be 
at the same position and height above the earth 
and is oriented to produce the same polarization. 

c. A long-wire or harmonic antenna is one with 
length greater than a half-wavelength and with 
current distribution such that there is a reversal 
of current flow in adjacent half-wave sections. 

d. As the length of a long-wire antenna is in- 
creased, the gain increases, the lobes of maximum 
radiation lie closer to the antenna itself, and a 
greater number of minor lobes is produced. 

e. In general, the effects of ground on long-wire 
antennas are the same as with the basic half-wave 
antenna. 

j. A nonresonant antenna is terminated in a 
resistance equal to its characteristic impedance. 
As a result, standing waves no longer exist on the 
antenna and the radiation pattern becomes 
unidirectional. 

g. The Beverage or wave antenna is a single 



terminated wire of 2 or more wavelengths, sup- 
ported on poles a short distance above ground. 
It is used to transmit or receive vertically polar- 
ized ground waves, particularly at low radio 
frequencies. The operation depends on a process 
known as wave tilt. 

h. The V antenna consists of two horizontal 
long wires arranged to form a V, and fed at the 
apex with currents of opposite phase. The radia- 
tion pattern is bidirectional along a line which 
bisects the apex angle. 

■i. The V antenna can be made unidirectional by 
connecting 500-ohm noninductive resistors be- 
tween the far ends of the legs of the V antenna and 
ground. 

j. The half-rhombic antenna is a terminated 
vertical antenna which has the form of an obtuse- 
angle V- This antenna works in conjunction with 
a ground or a counterpoise. An unbalanced 
transmission line is used to feed the antenna. 

k. The radiation from & half-rhombic antenna 
is unidirectional in the direction of the terminating 
resistor. A vertically polarized radio wave is 
produced by this antenna. 

I. The rhombic antenna is the highest develop- 
ment of the long-wire antenna. It is used widely 
for long-distance high-frequency point-to-point 
communication. 

m. The rhombic antenna is useful over a wide 
frequency range. It is easier to construct and 
maintain than are other antennas of comparable 
gain and directivity. The antenna is noncritical 
so far as operation and adjustment are concerned. 

n. Noninductive resistors are used to terminate 
the rhombic antenna used for receiving or for 
low transmitter powers. Dissipation lines are 
used for termination when high transmitter powers 
are applied to the rhombic antenna. 

o. A balanced oscillator can be used to check a 
rhombic antenna for correct termination. If the 
oscillator frequency does not change when it is 
connected to the antenna, no reactance is present 
at the input terminals of the antenna, and proper 
termination exists. 

p. Standard military rhombic antennas are 
designed for frequencies from about 4 to 22 mc 
at ranges from 200 to over 3,000 miles. 

q. A multiwire rhombic antenna has a lower 
and a more constant value of input impedance 
over a given frequency range than does the single- 
wire type. 

r. Resonant rhombic antennas have bidirec- 
tional radiation patterns, high input impedances, 



172 



varying characteristics over a given frequency 
range, and slightly less gain than do the conven- 
tional terminated rhombic antennas. 

93. Review Questions 

0. What is meant by antenna directivity? 

b. How is antenna gain measured and in what 
units is it expressed? 

c. Describe a long-wire antenna. 

d. In general, what happens to the radiation 
pattern of an antenna as its length is increased? 

e. Calculate the length in feet of a long-wire 
antenna which is to have two full waves of current 
distributed along its length at 4 mc. 

/. How is radiation resistance affected as the 
length of a long-wire antenna is increased? 

g. How can long-wire antennas be fed? 

h. Distinguish between resonant and nonreso- 
nant antennas. 

1. What is the wave antenna? 

j. What value of terminating resistance is 
required for the Beverage antenna? 
k. What is meant by the term wave tUit 
I. Describe the V antenna. 
m. Discuss the radiation pattern of the V 



antenna and describe in general how such a pattern 
is produced. 

n. Distinguish between the angles known as 
the tilt angle and the wave angle. 

o. Describe the half -rhombic antenna. 

p. What is the radiation pattern of the half- 
rbombic antenna? 

q. Distinguish between the half-rhombic an- 
tenna and the obtuse-angle V antenna. 

r. Give several methods of supporting the 
apex of the half-rhombic antenna. 

s. Give several advantages of a rhombic 
antenna. 

t. Give some disadvantages of a rhombic 
antenna. 

u. What is the polarization of a radiated wave 
transmitted by a rhombic antenna? How is this 
polarization produced? 

v. What is the purpose of a dissipation line? 

w. How can the termination of a rhombic 
antenna be checked? 

x. Describe the multiwire rhombic antenna 
and give some advantages of this type over the 
single-wire antenna. 

y. How can rhombic antennas be fed? 

z. Give some characteristics of the resonant 
rhombic antenna. 



173 



CHAPTER 5 
DRIVEN AND PARASITIC ARRAYS 

Section I. INTRODUCTION 



94. Multielement Arrays 

One means of attaining increased antenna gain 
and directivity is by use of the multielement 
array. The long wire, regardless of its length, is 
looked upon as a single radiating or receiving ele- 
ment; the array is a combination of elements 
which, considered separately, could be individual 
antennas. These elements act together or upon 
each other to produce a given radiation pattern. 
Various factors influence the choice of methods 
used to produce high directivity. Whereas the 
long-wire antenna often is preferred where recep- 
tion or transmission on more than one frequency 
is required and where gain or directivity require- 
ments are moderate, the more exact phasing and 
determination of element lengths in the array 
make for a more regular radiation pattern. Since 
fewer minor lobes are developed, available power 
is concentrated in the major lobe or lobes, and 
therefore there is greater gain and sharper direc- 
tivity in the favored direction. In a given avail- 
able space, the elements of an array can be so 
arranged as to provide greater gain than a long- 
wire antenna confined to the same space. 

95. Definitions 

a. Types of Elements. 

(1) As a special arrangement involving new 
factors and concepts, the array requires 
special terminology. It is made up of 
more than one element, but the basic 
element is, generally, the half-wave dipole. 
Sometimes it is made to have more or 
less than this length, but the deviation 
usually is not great. 

(2) A driven element is connected directly to 
the transmission line. It obtains its 
power directly from the transmitter or, 
in reception, it applies the received 
energy directly to the receiver. A para- 
sitic element, on the other hand, derives 



its power from another element in the 
same array. It is placed close enough to 
the other element to permit coupling and 
it is excited in this way. 

(3) If all of the elements in a given array are 

driven, the array is called a driven array. 
The term connected array sometimes is 
used to described this type. If one or 
more elements are parasitic, the entire 
system usually is considered to be a 
parasitic array. 

(4) A parasitic element sometimes is placed so 

that it will produce maximum radiation 
(in transmission) from its associated 
driver, and it operates to reinforce energy 
going from the driver toward itself. When 
so used, the parasitic element is referred 
to as a director. If a parasitic element is 
placed on the other side of the driven 
element 'and causes maximum energy 
radiation in the direction from itself to- 
ward the driven element, it is called a 
reflector. 

b. Directivity. Multielement arrays frequently 
are classified as to their directivity. A bidirectional 
array radiates in both opposite directions along the 
line of maximum radiation. A unidirectional arra 
radiates in only one direction. 

c. Types of Arrays. 

(1) Arrays have been described above with 
respect to their radiation patterns and 
the types of elements which comprise 
them. It is useful, however, to identify 
them by the physical placement of the 
elements and the direction of radiation in 
respect to these elements. Generally 
speaking, the term broadside array desig- 
nates any one in which the direction of 
maximum radiation is perpendicular to 
the plane containing the elements. In 
practice, however, this term is confined 
to those arrays in which the elements 



174 



themselves are also broadside or parallel 
in respect to each other. 

(2) A collinear array is one in which all the 
elements lie in the same straight line. 
The direction of propagation is broadside 
to the array. 

(3) An end-fire array is one in which the 
principal direction of radiation is along 
the plane of the array itself. 

(4) Sometimes a system is used incorporating 

characteristics of more than one of the 
three types mentioned above. For 
instance, some of the elements may be 
collinear, others may be parallel. Such 
an arrangement often is referred to as a 
combination array or an array of arrays, 
although, since maximum radiation 
occurs at right angles to the plane of the 
array, the term broadside array could be 
used. 

d. Front-To-Back Ratio. The front-to-back ratio 
is the proportion between energy radiated in 
the principal direction to the energy radiated in 
the opposite direction. 

96. Phasing 

a. Coupling in Space. 

(1) Various reflected and refracted compo- 
nents of the propagated wave create cer- 
tain effects of reinforcement and cancela- 
tion. At certain points distant from the 
transmitter, some of these components 
meet in space. Reception at these points 
is either impaired or improved. If the 
different components arrive at a given 
point in the same phase, they add, mak- 
ing a stronger signal available, and if they 
arrive out of phase, they cancel. 

(2) Effects of this kind are caused by factors 
operating some distance from the point 
of transmission. It is possible to cause 
somewhat similar effects to occur at the 
transmitting point itself. Consider an- 
tennas A and B, figure 147. They are 
two dipoles perpendicular to the plane of 
the page and are, therefore, shown as 
points. They are spaced a quarter- 
wavelength apart at the operating fre- 
quency. The radiation from either an- 
tenna, operating alone, is uniform in all 
directions in this plane. Consequently, 
the pattern produced by each antenna is 



a circle with the antenna at its center. 
Suppose, however, that current is being 
fed to both antennas from the same 
transmitter, but in such a way that the 
current fed to antenna B lags the current 
in antenna A by 90°, or the time required 
for a quarter of a cycle. Energy radiat- 
ing from antenna A toward receiving 
location X reaches antenna B after % 
cycle of operation. When it reaches an- 
tenna B, it meets the radiation from that 
antenna toward X in exactly the same 
phase. Therefore, radiation from both 
antennas add, and propagation toward 
X is strong. Radiation from antenna B 
toward receiving location Y reaches an- 
tenna A after % cycle. Since the energy 
in antenna A was % cycle behind that of 
antenna B to begin with, the radiation 
from both antennas toward receiving 
point Y are exactly 180° out of phase 
when they join. As a result, they cancel 
and no radiation occurs toward Y. At 
receiving points away from the line of 
radiation, indicated by the broken ar- 
rows, there are phase differences not 
quite so pronounced as to produce com- 
plete addition or outright cancelation. 
The over-all effect is indicated by the 
radiation pattern shown. The physical 
phase relationship caused by the quarter- 
wave spacing between the two elements, 



t 




TM S6S-I5I 

Figure 147. Phasing of antennas in space. 



US 



as well as the phase of the currents in 
them, has acted to change the radiation 
pattern. 
6. Stub Phasing. 

(1) In the case just discussed, it was men- 
tioned that the currents fed to both an- 
tennas from the same transmitter were 
90° out of phase. No explanation was 
given of the manner in which this is done. 
Sections of transmission line, called 
stubs, frequently are used for this pur- 
pose. These can be adjusted to produce 
any desired phase relationship between 
connected elements. 

(2) When two collinear half-wave elements 
are connected directly so that their cur- 
rents are in the same phase, the effect is 
that of a full-wave antenna (^4 of fig. 
148). The current in the first half-wave- 
length is exactly 180° out of phase with 
that in the second half-wavelength, as 
shown. This is the opposite of the 
desired condition. (In addition to the 
current waveform, arrows are used to 
indicate the direction of current flow 
which is more convenient for determining 
the phase of current on more complicated 
arrays.) 




TM 666-IB2 

Figure 148. Phasing of connected elements. 

(3) When the elements are connected by a 
quarter-wave stub as in B, current travels 
down one side of the stub and up the 
other. It travels a distance of a half- 
wavelength in the stub itself, and, as a 
result, it moves through % cycle of change. 



When the current reaches the next 
element, it is in the desired phase. Since 
the current in one side of the stub is equal 
and opposite to current on the other 
side, the fields produced cancel, and, 
consequently, there is no radiation from 
the stub itself. 

97. Mutual Impedance 

a Definition. 

(1) The impedance of an antenna at any 
point can be calculated by Ohm's law 
from the current and voltage at that 
point, but, in an isolated antenna element 
this impedance is known as self imped- 
ance. When another element is nearby 
the impedance is changed. Suppose that 
in figure 147, driven element A is parallel 
to and near unconnected (parasitic) ele- 
ment B. During operation, radiation from 
antenna A reaches antenna B, inducing 
current in the latter. The field asso- 
ciated with this current in turn influences 
antenna A. In other words, in addition 
to the current supplied to antenna A 
from the source, current is induced by 
antenna B. The current and voltage 
relationship at a given point on antenna 
A now will be different from the relation- 
ships that exist when no other element is 
nearby. Therefore, the impedance, cal- 
culated from current and voltage, will be 
different also. 

(2) When an antenna element is operating in 
association with other elements, a differ- 
ent value of impedance is measured 
which is referred to as its actual imped- 
ance. Mutual impedance is the imped- 
ance that results from the coupling 
between the two elements which is 
responsible for the difference between the 
self impedance and the actual impedance 
of a given element. 

6. Current Amplitude. When two parallel an- 
tenna elements are close together, the current 
induced in one by the other will be great. If one 
of these elements is driven and the other is not, 
the current in the driven element then is the cur- 
rent supplied by the transmitter simply added to 
that induced by the parasitic element, if there is 
no phase difference. Mutual impedance permits 



176 



greater gain in the array than in a single antenna, 
although there is no actual increase in transmitter 
power, and mutual impedance acts to decrease the 
actual impedance, since the current in the driven 
element has increased. As the parasitic element 
is moved farther from the connected element, 
there is less coupling. As a result, less current is 
induced in the driven element, and less gain is 
produced. The effect of mutual impedance de- 
creases and the actual impedance of the driven 
element approaches its self impedance. 
c. Current Phase. 

(1) The amplitude of the induced current is 
not the sole factor determining gain. In 
practice, antenna gain can be reduced to 
a smaller value even as the amplitude of 
the induced current becomes greater. 
This occurs when the induced and origi- 
nal currents are 180° out of phase, at 
which time mutual impedance acts to 
increase the actual impedance, since the 
current in the driven element is reduced. 

(2) The distance between two elements in 
terms of wavelength at the operating fre- 
quency determines the relative phase 
between them. Then, cancelation and 
reinforcement of signal resulting from 
phase difference are particularly notice- 
able when the distance between the two 
elements is a fraction of a wavelength. 

(3) Consider an antenna that is cut to reso- 
nance at a given frequency. Since the re- 
active components cancel out, its self im- 
pedance is purely resistive. When this 



antenna works with another element 
(parasitic) so that the current induced 
back into the former is exactly in phase 
with the original, there is increased ampli- 
tude with no change in phase. Although 
the actual impedance has decreased, it 
still is purely resistive. When the in- 
duced current is exactly 180° out of phase 
with the original, there is decreased am- 
plitude but still no change in phase. 
Again, the actual impedance remains re- 
sistive, although it is now greater than 
the self impedance. When the induced 
current is not exactly in phase or 180° 
out of phase with the original current, 
the phase of the total current shifts in 
respect to voltage. This change of rela- 
tive phase between current and voltage 
indicates that a reactive effect is present. 
As a result of this reactive effect, the an- 
tenna can be tuned off resonance by the 
presence of another element. It is cor- 
rect, therefore, to say that mutual im- 
pedance may contain both reactive and 
resistive components. 
(4) If a resonant antenna associated with a 
parasitic element is tuned off resonance, 
the phase of the current induced in the 
parasitic element is shifted. Therefore, 
the phase of the current induced back 
into the resonant antenna also is shifted. 
In other words, the tuning of a parasitic 
element also affects the reactive com- 
ponent of mutual impedance. 



Section II. DRIVEN ARRAYS 



98. General 

a. Description. 

(1) Driven arrays form a major subdivision 
of multielement arrays. The distinc- 
tive property of the driven array is the 
fact that all of the elements used derive 
their power from the same source, the 
transmitter. This property differenti- 
ates this group from the other major 
class of multielement systems, the para- 
sitic arrays, in which one or more ele- 
ments are driven directly whereas others 
are excited by these driven elements. 

(2) The driven array is preferred when 
high-power transmission is desired in 



addition to high directivity, because 
the driven array introduces less over-all 
power loss than occurs in other multiele- 
ment systems where loss of energy is 
caused by insufficient coupling between 
elements. 
b. Problems of Feeding. 

(1) Special attention must be given to 
certain factors involved in feeding driven 
arrays. Current distribution is one of 
these, and phasing is another. For 
example, in an array consisting of four 
elements, the same amount of power 
must be fed to each element, and the cur- 
rent in each element must be in phase 
with the current in each of the other 



177 



(2) 



three. Care must be exercised to main- 
tain these conditions as exactly as possi- 
ble. If deviations exist, undesired 
cancelations and reinforcements occur. 
As a result, undesired lobes can be intro- 
duced into the radiation pattern. De- 
aired lobes can be emphasized or deem- 
phasized, but beam width and directivity 
are affected and the advantages sought in 
the use of a given array can be nullified 
in this way. 

Care must be taken in interconnect 
ing the elements of an array since radia- 
tion from sections of transmission line 
used for interconnecting can reduce 
effectiveness. Pick-up by these sections 
when the array is used for receiving may 
cause interference. 



of B T at A A , as shown by the broken 
arrows. This effect is emphasized if the 
receiving antennas intercept energy well 
in all directions 




99. Directivity 



TM 666-153 

Figure l^d. Directivity and interference. 



a. Directivity and Gain. The directivity of an 
antenna or an array can be determined by exam- 
ining its radiation pattern. In an array propa- 
gating a given amount of energy, greater radiation 
takes place in certain directions than in others. 
The elements in the array can be manipulated in 
such a way that they change this pattern and 
distribute it more uniformly in all directions. 
They can be considered as a group of antennas, 
fed from a common source, facing in different 
directions. On the other hand, the elements can 
be disposed in such a manner that the radiation 
will be focused or concentrated in a single direc- 
tion. With no increase in power, the amount of 
radiation in a given direction is greater. Since 
there is no increase in the input power, this is 
achieved at the expense of gain in other directions. 
6. Directivity and Interference. 

(1) There are many applications in which 
sharp directivity is desirable although 
there is no need for added gain. Examine 
the physical disposition of the units 
shown in figure 149. A T and B T are 
transmitters. It is desired that they send 
information to receivers A R and B R , re- 
spectively, along the paths shown by the 
solid arrows. The distance between A T 
and A R , or between A T and B R , is not so 
great as to require high-power trans- 
mission. If the antennas of A r and B T 
propagate well in all directions, however, 
there is some pick-up of A T at B R and 



(2) The use of highly directional arrays as 
radiators from A T and B T , beamed along 
the paths of the solid arrows and with 
low radiation along the paths of the 
broken arrows, tends to solve the prob- 
lem. With this arrangement, consider- 
able power is available from the desired 
sources at the respective receiving points, 
and little power is available from the 
unwanted sources. Further improve- 
ment along these lines is obtained by the 
use of narrowly directed arrays as receiv- 
ing antennas at locations A R and B R . 

(3) The effect of this arrangement is to select 
a desired signal while discriminating 
against an interfering signal. The same 
approach can be used to overcome types 
of radiated interference other than un- 
wanted transmissions. In such cases, it 
is more important to -prevent radiation in 
certain directions than it is to produce 
greater gain in other directions. 

(4) The differences between the single-ele- 
ment antenna and the array are illus- 
trated in figure 1 50, in which A gives the 
relative field strength pattern for a hori- 
zontal polarized single antenna, and B 
shows the horizontal radiation pattern 
for one particular array. The antenna 
at A radiates fairly well in the desired 
direction, toward receiving point 1 

It radiates equally well, however toward 



178 



produced by the single dipole. The 
addition of another radiator, however, 
tends to intensify the pattern. A 
comparison of the two patterns in 
figure 151 shows that each consists of 
two major lobes in opposite directions 
along the same axis, Q to Ql. Along 
this line of maximum propagation, radi- 
ation is stronger with the added element. 
Moving toward the PP1 axis, this 
reinforcing effect falls off. The pattern 
in B is sharper or more directive. This 
is the same as saying that gain along 
the line of maximum energy propa- 
gation is increased, whereas the beam 
width is reduced. As more elements 
are added, the effect is heightened, 
although unimportant minor lobes are 
added. 

Q Q 



point 2 , although no radiation is de- 
sired in this direction. If the antenna at 
B is used in the same situation, it radiates 
strongly to point 1 but very little in 
the direction of point 2 Conse- 
quently, more satisfactory operation re- 
sults. 




TM -666-154 

Figure 160. Single antenna versus array. 

c. Major and Minor Lobes. The pattern shown 
in B has radiation concentrated in two lobes. 
The radiation intensity in the B 1 lobe is 
considerably stronger than in the B 2 lobe. 
B 1 is called a major lobe, B 2 a minor 
lobe. Since the complex radiation patterns 
associated with arrays frequently contain several 
lobes of varying intensity, it is convenient to 
adopt appropriate terminology. In general, 
major lobes are those in which the greatest 
amount of radiation occurs. Minor lobes are 
those in which the radiation intensity is less. 

100. Main Systems 

Within the family of driven arrays, there are 
three basic types — collinear, broadside, and end- 
fire. Any driven array is one of these types or 
represents a combination of more than one of them. 
a. Collinear Arrays. 

(1) Radiation from a half -wave antenna is 
represented by two broad lobes in 
opposite directions. A method of con- 
necting two such elements arranged in a 
straight line to operate in the same 
phase is shown in B of figure 148. 
Basically, the pattern radiated by this 
latter combination is similar to that 




01 A 01 B 

TM 666- tSS 

Figure 161. Single half-wave antenna versus two half-wave 
antennas in phase. 

(2) If all of the elements in a driven array lie 
in a single straight line, the array is 
known as a collinear array. The cur- 
rents in the various elements are always 
in the same phase. The elements are 
connected to each other by stubs adjusted 
to assure proper phasing. These ele- 
ments usually are a half-wavelength, but 
greater lengths also are used. To assure 
proper phasing, the connecting sections 
are a quarter-wavelength. However, 
when the length of the elements is in- 
creased, the length of the connecting sec 
tions must be decreased. 

(3) A method using four collinear elements is 
shown in figure 152. The arrows in A 
(side view) illustrate the distribution and 



179 



relative phase of current in the array. 
At the frequency for which it is designed 
to operate, any collinear array regard- 
less of the number of elements, produces 
a bidirectional pattern (top view in B) 
with axis perpendicular or broadside to 
the line of the elements. 

x x x x 



2 2 2 




FEEDER 
LINE 

SIDE VIEW 




TOP VIEW 



TM 666-15* 

Figure 152. Typical collinear array. 

b. Broadside Arrays. 

(1) Figure 153 shows an end view of two par- 
allel half-wave antennas, A and B, oper- 
ating in the same phase and located a 
half-wavelength apart. At a givsn point, 
P, far removed from the antennas, they 
appear as a single point. Energy radi- 
ating toward P from antenna <4 starts 
out in phase with energy radiating from 
antenna B in the same direction Prop- 
agation from each travels over the same 
distance to point P, arriving the r 'e in the 
same phase. In other words, the an- 
tennas reinforce each other in this direc- 
tion, making a strong signal available at 
P. Field strength measured at that 
point is greater than it would be if the 
total power supplied to both antennas 



had been fed to a single half -wave dipole. 
Radiation toward point PI is built up in 
the same manner. 



Q 




• Pi 



wl TM 666-157 

Figure 158. Parallel elements in phase. 

(2) Consider next a wavefront traveling 
toward point Q from antenna B. By 
the time it reaches antenna A, a half- 
wavelength away, half a cycle has 
elapsed. Therefore, energy from B meets 
the energy from antenna A 180° out of 
phase, with the result that energy from 
the two sources moving toward point Q 
cancels. In like manner, radiation from 
antenna A traveling toward point Ql 
meets and cancels the radiation from 
antenna B in the same direction. As a 
result, there is little propagation in either 
direction along the Q to Ql axis, most of 
it being concentrated in both directions 
along the P to PI axis. When both 
antenna elements are fed from the same 
source, the resiilt is the basic broadside 
array. 

(3) When more than two elements are used 
in a broadside arrangement, they are all 
parallel and in the same plane, as shown 
in figure 154. Current phase, indicated 
by the arrows in A, must be the same 
for all elements. The radiation pattern, 
shown in B, is always bidirectional. 
This pattern is sharper than the one 
shown in the previous illustration be- 



180 



SIDE VIEW OF ARRAY 




TOP VIEW OF ARRAY 
(END VIEW OF ELEMENTS) 



B 



TM 666-158 

Figure 154- Typical broadside array. 

cause of the addition of two elements. 
Directivity and gain depend on the 



number of elements and on the spacing 
between them. 
c. End-Fire Arrays. 

(1 ) The radiation pattern for a pair of parallel 
half-wave elements is shown in A of 
figure 155, fed 180° out of phase. The 
elements shown are spaced a half-wave- 
length apart, but, in practice, smaller 
spacings are used, for reasons to be dis- 
cussed later. Radiation from elements 

A and B traveling toward point 
P starts out with the 180° phase differ- 
ence. Moving the same distance over 
approximately parallel paths, the respec- 
tive wave fronts from these elements 
maintain the 180° phase difference. In 
other words, there is maximum cance- 
lation in the direction of P. The same 
condition holds true for the opposite 
direction. The P to PI axis, which is 
the line of propagation in the case of the 
broadside array, becomes the line of least 
radiation in figure 156, where the end- 
fire principle is used. 

(2) Consider what happens along the Q to 
Ql axis. Energy radiating from element 

B toward Q reaches antenna A about 
half a cycle or 180° after it leaves its 
source. Since radiation from element 




pi 




END VIEW 



SIDE VIEW 



B 

TM 666-159 



Figure 155. Parallel elements 180° out of phase. 



181 



A is originally 180° out of phase, the 
wave fronts are now approximately in 
the same phase moving toward! Q, and 
they reinforce. Similar reinforcement 
occurs along the same axis toward Ql. 

(3) In the example above, a bidirectional 
pattern is developed, which is not always 
true in end-fire operation. Figure 147 
is another application of the end-fire 
principle where elements are spaced a 
quarter-wavelength apart and phased 90° 
from each other to produce a unidirec- 
tional pattern. The importance of spac- 
ing and current phasing is apparent. 

(4) In A, figure 155, elements A and B 
are seen perpendicular to the plane 
represented by the paper and, therefore, 
only the ends of the antennas appear. 
If the antennas are rotated a quarter of 
a circle in space around the Q to Ql axis 
so that they are seen in the plane of the 
elements themselves, as shown in B, 
the jP to PI axis, now perpendicular to 
the page, is not seen as a line. The axis, 
R to Rl, now seen as a line, is perpen- 
dicular to P to PI as well as to Q to Ql. 
The end-fire array is directional in this 
plane also, although not quite so sharply. 
The reason for the greater broadness of 
the lobes can be seen by following the 
path of energy radiating from the mid- 
point of element B toward point S. 
This energy passes the A element at 
one end after traveling slightly more 
than the perpendicular distance between 
the dipoles. Energy from these, there- 
fore, does not combine in exact phase 
toward point S. Although maximum 
radiation cannot take place in this direc- 
tion, energy from the two sources com- 
bines closely enough in phase to produce 
considerable reinforcement. A similar 
situation exists for wave fronts traveling 
toward T, but the wider angle accounts 
for a greater phase difference with a 
resulting decrease in the strength of the 
combined wave. 

(5) To sum up, end-fire arrays consist of 
half-wave elements in which the currents 
are 180° out of phase. Directivity is 
off either or both ends of the array 
(along the axis of the array), as shown 
by the broken arrows in figure 156; hence 



the term end-fire is used. The patten* 
may be dibirectional or unidirectional, 
depending on the distance between 
elements and the relative phase of the 
currents flowing in them. Gain also 
depends on these two factors. Direc- 
tivity is achieved in two planes, but is 
sharper in one than in the other. The 
pattern in the plane shown in A is 
sharper than that for the plane in B. 



TOP VIEW OF ARRAY 

A 

* — O O — — G Q - 

SIDE VIEW OK ARRAY 
( END VIEW OF ELEMENTS) „ 

B 

TM 666160 

Figure 156. Typical end-fire array. 

101. Collinear Arrays 

a General Description. 

(1) The simplest type of collinear array is 
the center- fed "two half-waves in phase" 
arrangement shown in A, figure 157. 
The array in B also consists of two 
dipoles, but end feeding is used here, 
making necessary the use of a phase- 
reversing quarter-wave connecting stub 
between the elements. For purposes of 
feeding, notice that both arrangements 
mentioned so far are fed at voltage 
loops. In this connection, the terms end- 
fed and center-fed are likely to be mis- 
leading. In C, a three-element array is 
used. In D, the three-element array is 
center-fed with feed being introduced at 
the midpoint of one element. This is an 
instance of current feed. The system 
shown in E is the frequently used, 
balanced four-element collinear array. 

(2) More than four elements seldom are used, 
because, as more elements are added 
farther from the point of feeding, accu- 
mulated losses cause the farthest ele- 



182 



X 



2 



1 



X 



/ 
FEEDER 
LINE 



/ 

FEEOER 
UNE 



B 



/ 

FEEDER 
LINE 




Figure 157. Representative collinear arrays. 



ments to have less current than the 
nearest ones. This introduces an un- 
balanced condition in the system, which 
impairs its efficiency. Space limitations 
often provide another reason for limiting 
the number of elements. Since this type 
of array is in a single line, rather than in a 
stacked arrangement, the use of too many 
elements results in an antenna of several 
wavelengths. 

(3) The characteristic radiation pattern of a 
given array is obtained at the frequency 
or band of frequencies at which the sys- 
tem is made resonant; but the desired 
gain and directivity characteristics are 
lost when the antenna is not used at or 
near this frequency. The array then 
tunes sharply and acts as a simple long- 
wire antenna. However, it will be shown 
later that collinear arrays have higher 
radiation resistances than other types. 
If the resistance is higher, the Q is lower, 
and the antenna does not tune as sharply. 
A collinear antenna, then, is more effec- 
tive when used off its tuned frequency 
than an end-fire array. This feature is 
considered when transmission or recep- 
tion is to be over a wide frequency band. 
When more than two elements are used, 
this advantage largely disappears. 
b. Length and Phasing. 

(1) Although the half-wavelength is the basis 



for the collinear element, greater lengths 
often are used. Effective arrays of this 
type have been constructed in which the 
elements are 0.7 and even 0.8 wavelength 
which provides efficient operation at 
more than one frequency or over a wider 
frequency range. One frequent varia- 
tion uses elements cut to 0.64 wavelength. 
Whatever length is decided on, it is 
important that all of the elements in a* 
particular array closely adhere to it. 
If elements of different lengths are com- 
bined, current phasing and distribution 
are changed, throwing the system out of 
balance and seriously affecting the radia- 
tion pattern. 
(2) When the elements are made longer, it is 
necessary to decrease the size of the 
connecting stubs in order to maintain 
proper current phase (A of fig. 158). 
The arrows indicate the phase reversal 
at every half-wavelength. The first ele- 
ment, at the left, has current flowing in 
one direction for a quarter-wavelength 
and current flowing in the opposite 
direction for the remaining half-wave- 
length. The eighth-wavelength stub does 
not allow for a complete reversal of 
current flow. Instead, the change in 
direction takes place a quarter-wave- 
length along the length of the second 
element, making the distribution in this 



183 



length exactly like that in the first 
element. The current wave forms are 
shown to make this clear. In B, another 
possible distribution is shown for the 
same two elements. In each case, there 
are half-wave sections of both elements 
in the same phase. The distance be- 
tween one half-wave section as it exists 
on one element in the array and the 
corresponding similarly phased half-wave 
section on the adjacent element (the 
distance from 1 to 2) is greater in each 
of thesa examples than it is . when the 
elements themselves are exactly 1 half- 
wavelength long and are separated by 
quarter-wave connecting stubs. This 
affords an advantage, which will be 
discussed later. 





B 



TM 666-162 



Figure 158. Collinear elements longer than a half-wavelength 

(3) The length of the basic half -wave ele- 
ment used in this type of array and all 
driven arrays is determined from the 
following corrected formula: 



L = 



468 



where L is the length in feet, and / is the 
frequency in megacycles. The same 
formula applies whether the elements are 
of rigid construction or are made up of 
lengths of wire. For wire arrangements, 
the formula takes into account the ca- 



pacitive effects of spacers and support- 
ing insulators. When rigid elements are 
used, the greater thickness of these ele- 
ments can be considered to have much 
the same effect. 
c. Gain and Directivity. 

(1) Number of elements. As a general prin- 
ciple, increasing the number of elements 
in a collinear array also increases gain 
and directivity. Figure 159 shows the 
change in shape of the radiation patterns 
of various collinear arrays. The increase 
in gain produced is not shown in these 
patterns. Parts A, B, and C represent 
respectively the radiation patterns of 
typical two-, three-, and four-element 
arrays, all patterns being shown broad- 
side to the line of the elements. There 
is a practical limit, however, to the num- 
ber of elements. Availability of space is 
one limiting factor, the danger of unbal- 
ancing the array is another, and the third 
is a nonlinear gain over one array element 
less, which is shown in the chart below. 
The figures are for collinear arrays using 
half-wave dipoles with negligible spacing 
between the elements. Adding a third 
dipole to the basic two-element array 
provides an additional gain of 1.5 db. 
A fourth dipole affords another 1.2 db of 
gain; a fifth element provides only 0.8 
db of additional gain. As the number 
of elements is increased, the added gain 
thus achieved does not go up proportion- 
ally. When greater gain is important 
in a particular installation, good practice 
indicates the use of some type of array 
other than the collinear. The addition 
of elements to a collinear array increases 
loss resistance, both in the phase- 
reversing connecting stubs and in the 
elements themselves. Power consumed 
by this resistance is not radiated. The 
addition of dipole sections tends to un- 
balance, because some of the energy fed 
to the array is radiated by the elements 
nearest the point of feed before it can 
reach the sections farthest from this 
point. Consequently, the end elements 
radiate less than the center segments. 



184 



Number of elements 


Power gain 
(db) 


Gain over array 
using one element 
less (db) 


2 


1.8 

3. 3 

4. 5 
5.3 




3 


i. 5 
1. 2 
0. 8 


4 


5 





(2) Spacing. 

(a) The lower relative efficiency of collinear 
arrays of many elements compared 
with other multielement arrays relates 
directly to spacing and mutual im- 
pedance effects. Mutual impedance 
is an important factor to be consider- 
ed when any two elements are parallel 
and are spaced so that there is consid- 
erable coupling between them. Be- 
tween collinear sections there is very 
little mutual impedance, but, where it 
does exist, it is caused by the coupling 
between the ends of adjacent elements. 

(6) Constructional problems, especially 
where long lengths of wire are involved, 
frequently make it necessary to place 
the ends of the elements close together. 
Another limit on the spacing is the 
physical construction of the connecting 
stub. If the width of the stub is made 
too great or if the stub is not connected 
properly between the segments, un- 
desired radiation results. When rigid 
elements are used at the higher fre- 
quencies and correspondingly shorter 
wavelengths, the advantages of opti- 
mum spacing can be realized in a 
practical way. The graph of figure 
160 shows the relationship of spacing 
between adjacent ends and gain for two 
half-wave collinear elements. 

(c) Spacing often is referred to as the dis- 
tance between the center points of ad- 
jacent elements rather than the ab- 
solute distance between their ends. 
If the spacing is given as 3 quarter- 
wavelengths, center to center, where 
half-wave dipoles are involved it is the 
same as saying that the space between 
their ends is slightly greater than a 
quarter-wavelength. It is slightly 
greater because end effect accounts for 




90* C 



TM 666-163 

Figure 159. Free-space patterns for collinear arrays. 

185 



1 4 

<=* 
2 

X) _ 



the dipoles being shorter than a half- 
wavelength. 
(d) The effects of spacing, and the advan- 
tages of proper spacing, can be demon- 
strated by some practical examples. 




10 

20 

30 

40 

50 

60 

70 

80 

90 

!00 

110 

120 



.1 .2 .3 .4 .5 .6 .7 .8 .9 1.0 

SPACING BETWEEN ENDS OF 
ELEMENTS (IN WAVELENGTHS) 

TM 666-164 

Figure 160. Spacing and gain for collinear elements. 

A collinear array consisting of two 
half-wave elements with half-wave- 
length spacing between centers has a 
gain of 1.8 db. If the ends of these 
same dipoles are separated so that the 
distance from center to center is 3 
quarter-wavelengths, and they are 
driven from the same source, the gain 
increases to 1.2 db. Reference to the 
chart above shows that a three- 
dipole array with negligible spacing be- 
tween elements gives a gain of 3.3 db. 
In other words, when two elements are 
used with wider spacing, the gain thus 
obtained is approximately equal to the 
gain obtainable from three elements 
with close spacing. The spaced array 
permits simpler construction, since 
only two dipoles are used, and occupies 
less space. Reference to the graph of 
figure 160 shows that maximum gain 
is reached when spacing between the 
elements is in the vicinity of 0.4 or 0.5 
wavelength. However, constructional 



problems usually dictate smaller prac- 
tical spacing. 
(e) Optimum spacing is difficult to arrange 
in general, and it is particularly a 
problem when wire elements rather 
than rigid elements are used at fre- 
quencies having fairly long wave- 
lengths. The only spacing between 
wire elements generally is provided by 
the insulator between lengths of wire. 
There is a practical way, however, of 
achieving the desired effects of in- 
creased spacing without actually in- 
creasing the physical distance between 
the ends of the elements. This is done 
by making the elements themselves 
longer, which, although it gives the 
effect of increased spacing, simplifies 
construction. 
d. Feeding Methods and Adjustment. 

(1) Feeding. Collinear systems usually are 
fed at an end point between half-wave 
elements. Since there is a voltage loop 
and a current null at such a point, the 
impedance is relatively high, generally 
around 1,500 ohms. This impedance, 
together with the low Q of collinear 
arrays resulting from the relatively high 
radiation resistance and consequently 
lower standing-wave ratio, permits a 
certain amount of mismatch between the 
feed line and the antenna. For this 
reason, flat or nonresonant lines are used 
widely. The use of the fiat line provides 
another advantage if operation is to be 
on more than one frequency. The per- 
missible degree of mismatch makes it 
possible to feed a conventional collinear 
array with open-wire 600-ohm line in 
which the standing-wave ratio is 2 to 1 or 
less. In the less frequently encountered 
case of the current-fed collinear array 
(when feed is introduced at the center of 
one element as it sometimes is with the 
three-element arrangement to maintain 
balance), impedance at the feed point 
closely matches 300-ohm line. The use 
of 600 ohm line also is permissible here, 
since the standing-wave ratio is still low. 
Conventional matching devices can be 
used, however, if it is desired to connect 
line and antenna. 

(2) Adjustment. When only two elements are 



186 



used with center feed (between the ele- 
ments), both lengths must be the same 
and no great problem exists. When more 
than two elements are involved, adjust- 
ment to resonance should begin with the 
two connected directly to the feed line 
and with a constant input to the antenna. 
The length thus obtained may not cor- 
respond exactly to the calculated length. 
It is important, however, that all elements 
be of the same length to maintain balance. 
Once this measurement is determined, 
cut all elements to be used to the same 
size. The length of the phasing stubs 
must be determined by formula for the 
particular type of line used. Cut them 
slightly longer than calculation indicates. 
Effective adjustment for proper phasing 
can be made only during actual assembly. 
One additional element with its associated 
stub is added to one side of the basic 
two-element array. Next, using a 
shorting bar, determine the point along 
the stub at which current is maximum. 
Cutting at this point makes the length of 
the stub correct. If the array is not to 
be fed at its center (with respect to all its 
elements), add one element at a time 
with its associated stub. If the array is 
balanced, add two elements at a time, 
one to either end. In either case, the 
procedure just mentioned for phasing the 
stub or stubs is repeated. After the 
addition of each new element or pair of 
elements, the entire system must be 
checked to make certain that it is tuned 
properly. If a matching stub is used 
between antenna and line, it must be 
adjusted for maximum current also, but 
only after all elements have been con- 
nected and phased properly. 

1 02. Broadside Arrays 

a. General Description. Physically, a broadside 
array looks like a ladder. When the array and the 
elements in it are polarized horizontally, it looks 
like an upright ladder. When the array is polar- 
ized vertically, it looks like a ladder lying on one 
side. Horizontally polarized arrays using more than 
two elements are common, since the requirement 
that the bottom of the array be an appreciable 
distance above the earth presents constructional 



problems. Compared with collinear systems, 
arrays of this kind tune sharply, and therefore, 
lose efficiency rapidly when not operated on 
the frequencies for which they are designed. The 
higher Q resulting from the lower radiation resist- 
ance is responsible for this. 
6. Gain and Directivity. 

(1) Spacing. The physical disposition of di- 
poles operated broadside to each other 
allows for much greater coupling between 
them than can occur between collinear 
elements. Moving the parallel antenna 
elements closer together or farther apart 
materially affects the actual impedance 
of the entire array and the over-all 
radiation resistance as well. This critical 
effect of spacing may be seen from the 
graph of figure 161, in which the gain 
of two broadside elements is plotted 
against the spacing between them. 
Compare this with a similar graph for 
two collinear dipoles (fig. 160). Both 
curves follow the same general path, 
but the one for broadside elements is 
much sharper. For collinear elements, 
since spacing is varied between and 
1.0 wavelength, there is a variation in 
gain of about 1.5 db. For broadside 
dipoles, the range of variation in gain 
is nearly 5 db over the same range of 
spacings. In addition, the optimum 
spacing as far as gain alone is concerned 
occurs at a slightly greater spacing 
than is the case for collinear dipoles, 
approximately 0.65 wavelength. How- 
ever, to simplify phasing and feeding, 
half-wave spacing generally is used. 
Less than 1 db of gain is sacrificed by 
this expedient. Where more elements 
are used, the gain sacrificed is even 
greater. However, it can be recovered 
easily since the space saved can be 
used to accommodate one or more 
additional elements. As the spacing 
between broadside elements is increased, 
the effect on the radiation pattern is to 
sharpen the major lobes. When the 
array consists of only two dipoles exactly 
a half-wavelength apart, there are no 
minor lobes at all. Increasing the dis- 
tance between the elements beyond that 
point, however, tends to throw off the 
phase relationship between the original 



187 



current in one element and the current 
induced in it by the other element. The 
result is that, although the major lobes 
are sharpened, minor lobes are introduced 
even with two elements. These, how- 
ever, are not large enough to be of 
consequence. 



5.0 




.1 .2 3 .4 .5 .6 .7 .8 .9 1.0 

SPACING BETWEEN ELEMENTS 
(IN WAVELENGTHS) 



TM 666-165 

Figure 161. Spacing and gain, broadside elements. 

(2) Number of elements. The increase in 
gain of a broadside array as more 
elements are added is marked as com- 
pared to the increase with a collinear 
system. Reduced radiation resistance 
resulting from the efficient coupling 
between dipoles accounts for most of 
this. However, certain practical factors 
limit the number of elements that 
may be used. The constructional prob- 
lem increases with the number of ele- 
ments, especially when they are polarized 
horizontally (upright ladder). The fol- 
lowing chart shows the effect on gain 
by adding elements. With 3-quarter- 
wavelength spacing, the combined effects 
of optimum coupling and additional 
elements act to increase gain still fur- 
ther, and whenever more than two 
elements are used, minor lobes are 
developed regardless of the spacing. 
These lobes are greater than those 
developed by collinear arrays using 
the same number of elements. 



Number of elements 

. 


Gain in db (X/2 
spacing) 


Gain in db 
(3X/4 spacing) 




4. 

5. 0-5. 5 

6. 0-7. 

7. 0-8. 

8. 0-9. 


4. 5 
7.0 
8. 5 

10. 

11. 













(3) Radiation patterns. A representative pat- 
tern for a broadside array with four 
elements is shown in figure 154. The 
pattern is taken in free space in the 
plane perpendicular to the elements. 
Since the elements are vertical, the 
characteristic broadside pattern is devel- 
oped horizontally; the vertical pattern 
is that of the ordinary half-wave antenna. 
The characteristic pattern varies slightly 
from the illustrated free-space pattern 
when the elements are horizontal or 
stacked. This variation is a function 
of ground reflection. The resulting pat- 
tern can be developed by applying the 
reflection factor discussed in paragraph 
58. For this purpose, the height above 
the earth of the center element (or the 
average height of the entire array) is 
considered to be the height from ground. 
The resultant low angle of radiation is 
a feature of this type of polarization. 

c. Variations. When an antenna system con- 
sists of purely broadside, in-phase elements, 
there is little difference between one and another 
aside from the number of elements and the 
spacing, both of which have been discussed, and 
the means of feeding. The lazy H, Sterba 
curtain, and Bruce array all represent systems 
using collinear as well as broadside elements. 
There is also a popular four-element arrangement 
combining broadside with end-fire elements. 

d. Phasing and Feeding. 

(1) Phasing. 

(a) One frequently used means of supplying 
currents in the same phase to broadside 
elements is shown in figure 162. In 
A, the feed line is connected directly to 
the center of the phasing section 
(vertical) between the dipoles. Con- 
sider that leg of the vertical section 
connected directly to the dipoles and 
its associated side of the feed line. 



T88 



The two halves of this side of the 
section are parallel with respect to the 
feed point. Current traveling up this 
side of the feed line continues to move 
in exactly the same phase up one 
side of the section and down the other. 
Phase reversal in either portion of 
this line occurs at exactly the same 
distance away from the point of fe'ed. 
Electrically, the driven elements are 
merely series extensions of these ex- 
actly phased portions of line. There- 
fore, current in them must be also 
in the same phase. 

(6) Although the elements are represented 
as being exactly a half-wavelength 
apart, this need not be considered a rule 
when only two sections are involved. 
If the feed is introduced at the exact 
midpoint, correct phase is maintained 
regardless of the length of the phasing 
line. Advantage then may be taken 
of the larger optimum spacing, as indi- 
cated in the graph of figure 161. In 
B (fig. 162), where the array consists 
of more than two elements, adherence 
to half-wavelength spacing becomes 
necessary. The phasing of the two 
center dipoles is the same as in A. 
However, current traveling up the 
half-wavelength phasing section feed- 
ing the upper dipole undergoes a half- 
cycle of phase reversal. At the point 
where the section joins the top element, 
the current is 180° different from the 
desired phase. Connecting the ele- 
ment to the opposite side of the line, 
however, puts it exactly in the correct 
phase. The bottom element is driven 
in the same way. For the sake of 
clarity in showing the connections, the 
top and bottom dipoles have been made 
to appear slightly out of line with 
respect to the two in the center. In 
practice this is not necessary. The 
phasing lines may be transposed or 
dressed to accommodate the position- 
ing of the dipoles. Here, spacings 
greater than a half-wavelength are not 
possible since the phasing lines must 
be exactly that long. 

(c) Energy also can be introduced at the 
junction between the feed line (or 



2 




FEED 
LINE 



FEED 
LINE 



B 

TM 666-166 

Figure 162. Phasing broadside arrays, first method. 

matching device) and one of the ele- 
ments (A of fig. 163). Current phase 
in the lower element, shown by the 
arrow, is determined by the feed. At 
the other end of the vertical phasing 
line (180° or 1 half-wavelength away), 
the upper dipole is connected to the 
opposite side of the line. Phase re- 
versal in the line is canceled out, and 
the elements are kept in the same phase 
as desired. In B, phase is established 
for the upper element in the same way 
as for the upper dipole in A. The 
center dipole in B is connected just as 
is the lower element in A. The bottom 
element in B is connected and phased 
exactly as is the top dipole in the same 
array. Since they are exactly 1 wave- 
length apart, and exactly the same 
distance from the point of feed, they 
must be in the same phase. When 
three or any other odd number of 
broadside elements is used, this method 
of applying signal to the center of the 
array is desirable because it makes for 
better balance. If more elements are 
added to either side of the array, they 
are connected to alternating sides of 
the phasing line. In this way, every 
even dipole is connected to one side 
of the phasing line; every odd dipole is 

189 



FEED 
LINE 



FEED 
LINE 



B 



th cee-isT 

Figure 163. Phasing broadside arrays, second method. 

connected to the other side. The same 
spacing restrictions are applied, 
(rf) The widely used method of feeding a 
half-wave dipole at its current loop 
(center of the dipole) also can be ap- 
plied to the broadside array. In A of 
figure 164, the elements are a pair of 
dipoles, fed at their center points, and 
connected in parallel. At any spacing 
the elements are always in phase pro- 
vided that energy is introduced at the 
center of the line between them. If 



' 2 ' 



X 
2 




v ~ FEED LINE 



additional elements are added to this 
basic arrangement, the array assumes 
the appearance shown in C, and the 
extreme dipoles must be connected to 
that side of the phasing section which 
is opposite to the side feeding the cen- 
ter elements. Another method for 
center-feeding the elements is shown in 
B. Energy is introduced to one of the 
extreme dipoles as though it were an 
ordinary, center-fed, half-wavelength 
section. All additional elements are 
joined at their centers to the continu- 
ous phasing line ; the sides of the feed- 
ing line to which the respective halves 
of the elements connect are alternated 
throughout. The half-wavelength 
spacing restriction limits all of the ar- 
rays shown here just as it limits all of 
the multielement broadside arrays 
presented up to this point. 
(«) When space permits, full advantage of 
the gain attainable with 0.6- to 0.7- 
wavelength spacing can be taken. 
Phasing sections a full wavelength long 
are inserted between one dipole and 
another. These connecting lengths are 
bent or dressed as shown in figure 165, 
or in some similar fashion. The space 

L *~ 

I 2 



FEED LI N E - 



B 



1 




FEED LINE 



Ttt 666-168 



Figure 164- Phasing broadside elements, third method. 



190 



between elements in this example is 3 
quarter-wavelengths. Current moving 
in any phasing section undergoes a 
full, 360° cycle of change so that it is 
fed to an element at the same phase 
point. With full-wave connecting 
lengths, all dipoles connect to the same 
side of the line. Any spacing up to 1 
wavelength is made possible by this 
method. The feed line or matching 
device usually is connected where the 
phasing section joins an element. 
Feeding at the center section gives the 
best balance. 




TM S66-I69 

Figure 166. Phasing broadside elements, fourth method. 

(2) Feeding. Broadside arrays may be fed by 
either resonant feeders or flat lines with 
matching devices. A two-element array 
phased by the resonant feeders presents 
an impedance of less than 100 ohms at 
the point of feed. Although the phas- 
ing line between the elements may be 
longer than a half-wavelength, it is only 
with an exact half-wavelength that the 
feed impedance is purely resistive. With 
other lengths, there is a reactive compo- 
nent that must be tuned out. When an 
array phased in this manner has four 
elements (B of fig. 162), the impedance 
at the point of input is about 250 ohms 
with open-wire phasing line. In any 



case where the transmission line is intro- 
duced at the end of one element, the im- 
pedance is approximately 1,000 ohms. 
When the transmission line is introduced 
as shown in A of figure 164, the input im- 
pedance is about 3,000 ohms. Where 
more elements are used with open-wire 
line as in C, the input impedance is 
about 1,500 ohms. Matching to the in- 
put of broadside arrays has to be worked 
out for individual cases. 

103. End-fire Arrays 

a. General Description. 

(1) In appearance, an end-fire system does 
not look very different from a broadside 
array, the ladder-like appearance being 
characteristic of both. The currents in 
the elements, however, are 180° out of 
phase with each other. Because the 
plane in which the maximum radiation 
develops is in the plane of the elements, 
however, a vertically stacked arrange- 
ment is not likely to be operating as an 
end-fire array. Instead, the end-fire 
array is more likely to be constructed like 
a ladder lying on its side (elements 
vertical) or like one lying flat (elements 
horizontal). Moreover, the dipoles in 
an end-fire system are closer together 
(eighth-wavelength to quarter-wave- 
length spacing) than they are for broad- 
side radiation. 

(2) Closer spacing between elements permits 
compactness of construction. For this 
reason, an end -fire system is preferred 
to other types when high gain or sharp 
directivity is desired in a confined space. 
However, the close coupling creates 
certain disadvantages. Radiation re- 
sistance is extremely low, sometimes in 
the order of 10 ohms, making the 
possibility of antenna losses greater. 
Furthermore, this lower resistance is 
responsible for a higher Q, with the result 
that end-fire antennas are narrowly tuned 
affairs. This confines the array closely 
to a single frequency and introduces the 
danger of detuning with changes in 
climatic or atmospheric conditions. 

(3) The major lobe or lobes occur along the 
axis of the array. The pattern is sharper 



191 



5.0 




1.5 



1.0 
.5 



.1 .2 .3 .4 .5 .6 .7 .8 9 1.0 

SPACING BETWEEN ELEMENTS 
(IN WAVELENGTHS) 

A 

Figure 166. Spacing, gain, and radiation 

in the plane which is at right angles to 
the plane containing the elements. If 
the elements are not exact half-wave- 
length dipoles, operation is not affected 
materially. However, the required bal- 
ance of phase relationships and critical 
feeding makes it important that the 
array be symmetrical. Folded dipoles 
are used frequently because the im- 
pedance at their terminals is higher. 
This is an effective way of reducing Q and 
avoiding excessive antenna losses. 
Another expedient to reduce losses is the 
use of tubular elements of wide diameter. 
b. Gain and Directivity. 

(1) The gain available from a pair of dipoles 
phased 180° apart at different spacings 
is graphed in A of figure 166. A com- 
parison with a similar graph for the same 
pair of elements operated in phase, or as 
a broadside array (fig. 161) shows an 
inverse relationship between the two. 
This is caused by the inverse phase 
relationship. In B, the relationship 
between spacing and radiation resistance 
is shown. 

(2) In end-fire arrays, directivity increases 
with the addition of more elements and 
with spacings approaching the optimum. 
The directive pattern for a two-element, 




.1 .2 .3 .4 5 .6 .7 .8 .9 1.0 

SPACING BETWEEN ELEMENTS 
(IN WAVELENGTHS) 



B 

TM 666-170 

resistance with end-fire elements. 

bidirectional system is illustrated in 
figure 155, where A shows radiation 
along the array axis in a plane perpen- 
dicular to the dipoles, and B shows 
radiation along the array axis in the 
plane of the elements. These patterns 
were developed with 180° phase differ- 
ence between the elements. Additional 
elements introduce small, minor lobes. 
(3) With 90° phase difference in the energy 
fed to a pair of end-fire elements spaced 
approximately a quarter-wavelength 
apart, unidirectional radiation can be 
obtained. The pattern perpendicular 
to the plane of two elements is shown in 
A of figure 167. The pattern shown in B, 
taken in the same plane, is for a six- 
element array with 90° phasing between 
adjacent elements. Since both patterns 
show relative gain only, the increase in gain 
produced by the six-element array is not 
evident. End-fire arrays are the only ones 
wholly made up of driven elements that 
can be unidirectional. 
c. Variations. 

(1) End-fire elements are used frequently in 
combination with other types of elements 
to procure a particular kind of radiation 
pattern or to obtain extra gain. In figure 
168, a four-element antenna, arrows 



192 



90* 




90* 



90° 




B 

TM 666-171 

Figure 167. Unidirectional end-fire arrays. 

indicate the directions in which bidirec- 
tional radiation develops. This antenna 
system can be regarded as a pair of two- 
element end-fire arrays, operating broad- 
side to each other. The two top ele- 
ments can be considered as one end-fire 
array, the bottom pair as another. The 
radiation pattern developed in the plane 
shown in A is similar to that shown in 
A of figure 155, except that the lobes are 
sharper and the entire pattern is rotated 



m 3X 

m 4 



o'o 

END VIEW 



FRONT VIEW 



B 





TOP VIEW 



PICTORIAL 

D 

TM 666-172 

Figure W8. Combination end-fire and broadside elements. 

90 °. The pattern shown in C is like that 
of B, figure 155. 
(2) The dipoles in figure 168 are fed at their 
centers, the main feed line branching into 
four legs, one feeding each element. The 
top and bottom elements on the right, in 
A, are in the same phase, or broadside. 
The two elements to the left are also in 
the same phase, but they are 180° out of 
phase with the elements on the right, as 
indicated by the arrows in D. The ele- 
ments on the left are connected directly 
to the phasing section of the transmission 
line without transposition. Connections 
to the right-hand elements are trans- 
posed. Spacing between the left and 
right elements, shown as eighth-wave- 
length in the figure, actually can vary 
from that measurement up to a quarter- 
wavelength. Spacing between top and 
bottom elements, shown as 3 quarter- 



193 



wavelengths, can vary from 3 eighth- 
wavelengths to 3 quarter-wavelengths. 
Depending on spacing, gain for this 
combination array varies from about 6.5 
db to about 9 db. The 9-db gain is feed 
obtained with the spacing indicated in LINE 
A. Gain for any combination of spac- 
ings is obtained by adding the gain as 
read from the graph of figure 161 for the 
broadside spacing used to the reading 
from the graph of A of figure 166, for the feed 
end-fire spacing used. L,NE 
d. Phasing and Feeding. 
(1) Phasing. 

(a) Principal methods of operating parallel 
elements 180° out of phase are shown 
in figure 169. The method shown in 
A is acceptable when spacing is no 
more than an eighth-wavelength. 
Here, either side of the single-leg 
phasing section is no greater than a 
sixteenth-wavelength, and very little 
radiation takes place from these short 
lengths. The methods used in B and 
C appear similar to broadside phasing 
techniques. Two fundamental differ- 
ences, however, account for the 180° 
phase difference here as contrasted to 
to the in-phase relationship of broad- 
side elements. The phasing sections 
and spacing are always considerably 
less than a half-wavelength, and the 
point of feed can be different. The 
arrows indicate the direction of current 
flow at a given instant. In D, a 
method for center-feeding many ele- 
ments is shown. Whatever the spac- 
ing between elements, the lengths of 
the phasing lines between one element 
and the next must be exactly 1 half- 
wavelength to maintain the phase 
difference desired. Energy always 
must be applied at the junction of the 
phasing line with one of the elements. 

(b) Phasing of unidirectional end-fire arrays 
requires a 90° difference and must be 
arranged in a different manner. 
Quarter-wavelengths of phasing line 
can be used to provide the quarter- 
cycle phase difference required. 
Another popular method for a two- 
element array involves two separate 
transmission lines, one for each ele- 




D 

TM 666-173 

Figure 169. Phasing end- fire elements. 



ment. The transmission lines are 
matched individually to the elements, 
and one of the lines is exactly a quarter- 
wavelength shorter or longer to produce 
the desired phase relationship. Both 
lines are of the same type and are fed 
simultaneously from a coupling circuit 
at the transmitter. The advantage of 
this method is that it provides control 
at the transmitter of the amount of 
current fed to each element. In this 
mode of operation, mutual coupling is 
not the same for both elements, with 



194 



the result that each has a different 
radiation resistance. 
(2) Feeding. With eighth-wavelength spacing 
of center-fed elements, radiation resist- 
ance falls as low as 8 ohms, and the 
standing- wave ratio becomes very high. 
The ratio can be in the order of 30 to 1 or 
higher, and this accounts for the critical 
tuning of such arrays. If the array is 
reasonably close to the transmitter, a 
tuned, open-wire line can be used, but 
when longer transmission lines are used, 
they should be matched carefully at the 
operating frequency to reduce the high 
standing wave ratio (SWR). Folded 
dipoles as end-fire elements are used to 
take advantage of the higher impedance 
at their terminals. The folded elements 
are fed and matched individually through 
quarter-wave stubs to increase impedance 
still further. It then is possible to reduce 
the SWR to no more than 2 to 1. 

1 04. Other Driven Arrays 

All of the driven arrays discussed so far have 
been basic collinear, broadside, and end-fire types. 
Some special combination driven arrays are 
discussed in the following paragraphs. 

a. Extended Double-Zepp. 

(1) A of figure 170, shows a two-element 
collinear array, with wide spacing, in 
which each element is longer than a 
half-wavelength, the optimum length 
usually being 0.64 wavelength. 

(2) The greater than normal spacing between 
the half-wave sections at the ends of the 



wires carrying in-phase currents results in 
an increase in gain of slightly more than 
1 db over the gain of the basic collinear 
array with two half-wave dipoles. Un- 
desired radiation is small from the 
center of the array in which current flows 
in the opposite direction, because of the 
short sections of wire used (0.64 minus 
0.5 is only 0.14 wavelength). 
(3) The radiation from this array is bidirec- 
tional in a plane containing the two 
elements. A circular radiation pattern is 
produced in a plane that is at right angles 
to the elements. 
b. Lazy-H Antenna Array. 

(1) The lazy-H antenna array consists of two 
collinear arrays connected in parallel as 
shown in B of figure 170. Each collinear 
array, consisting of two half-wave ele- 
ments, usually is spaced a half-wave- 
length from its neighbor. The current 
in each of the four half-wave elements is 
in phase, so that maximum radiation 
occurs at right angles to the plane of 
the array. 

(2) The lazy-H array shown in B has a gain 
of about 6 db. Although smaller spac- 
ing between the two collinear portions 
can be used, a reduction in gain results. 
Larger spacing can be used also, which 
results in a slight increase in gain; how- 
ever, the input impedance is no longer 
purely resistive as it would be with 
half-wave spacing. 

(3) With the half-wave spacing, the input 
impedance is about 100 ohms (resistive). 
When a high input impedance is required 




TRANSMISSION 
LINE 

EXTENDED DOUBLE - ZEPP ARRAY 




TRANSMISSION 
LINE 

LAZY-H ARRAY 



B 

TM 6S6-IT4 



Figure 170. Extended double-Zepp and laty-H arrays. 



195 



that is also resistive, the transmission 
line is connected directly at the lower 
center insulator, and the connecting line 
to the upper elements is transposed once. 
This is required to put the currents in the 
upper elements in phase with the cur- 
rents in the lower elements. The input 
impedance then is in the vicinity of 
2,000 ohms. 
(4) The radiation pattern produced by this 
array is bidirectional in both horizontal 
and vertical planes. The beam width 
is not as broad as in the horizontal plane. 
c. Sterba Curtain. 

(1) The Sterba curtain, shown in A of figure 
171, consists of two collinear arrays, 
usually made up of a large number of 
elements stacked one above the other. 
The spacing between the two collinear 
arrays is a half-wavelength. 

(2) When the array is arranged as shown, 
only the horizontal elements produce 
useful radiation. The three twisted ver- 
tical sections of line do not radiate since 
one conductor of each pair carries current 
in one direction and the other conductor 
carries current in the opposite direction, 
and their fields, therefore, cancel. Also, 
the two single vertical half-wave con- 
ductors at either end of the array pro- 
duce little radiation since the current 
flow in the upper half is in the opposite 
direction to that flowing in the lower half. 

(3) Although the array shown in A consists 
of only three half-wave elements as the 



top section and a similar number as the 
bottom section (actually two half-wave 
elements in the center and two quarter- 
wave elements at the ends) it can be 
extended to include many more elements. 

(4) The gain of the array is increased as it 
is lengthened to include more collinear 
elements. In the array shown, the 
equivalent of three collinear elements 
provides a gain of over 3 db. To this 
must be added the gain provided by 
operating the additional set of collinear 
elements beneath the first. This pro- 
vides an additional gain of about 4 db, 
so that the total gain of the array is 
about 7 db. 

(5) When the Sterba curtain is fed as illus- 
trated, the impedance is about 600 ohms. 
This permits the use of a 600-ohm non- 
resonant two-wire line. Another com- 
mon feeding point is at the exact center 
of the lower collinear array between the 
two half-wave sections. At this point, 
the input impedance of the array is 
about 1,000 ohms. 

(6) A unique advantage of the Sterba curtain 
is that it provides a closed circuit for 
power-frequency alternating current or 
direct current. Such current sometimes 
is used to heat the elements to prevent 
ice formation. 

d. Bruce Array. 

(1) The Bruce array consists of a single wire 
folded as shown in B of figure 171. All 
vertical and horizontal lengths are a 




TRANSMISSION 
LINE 



STERBA CURTAIN 




TRANSMISSION 
LINE 




BRUCE ARRAY 



B 

TM 666-175 



Figure 171. Sterba curtain and Bruce array. 



196 



quarter-wavelength with the exception 
of two at the ends of the array which are 
only 1 eighth-wavelength. 

(2) The distance from the extreme left end 
of the array to point 1 represents a total 
distance of a half-wavelength (an eighth- 
wavelength plus a quarter-wavelength 
plus an eighth-wavelength). In this 
section of the array, current flows in the 
direction toward point 1 which is indi- 
cated by the current wave and the small 
arrows. In the next half-wave section, 
between points 1 and 2, a current re- 
versal occurs. The current now flows 
from point 2 toward point 1. In the next 
half-wave section, between points 2 and 
3, another current reversal occurs. The 
current now flows from point 2 to point 3. 
This portion of the array is unfolded in 
the dotted view so that the directions of 
current flow are made clear. Similar 
reversals occur throughout the length of 
the array. 

(3) Because of the folding of the wire, the 
direction of current flow is the same (that 
is, down) in each of the six vertical 
quarter-wave sections comprising the 
antenna. On the other hand, half of the 
small amount of current that flows in the 
horizontal sections of the array flows in 
one direction and half flows in the oppo- 
site direction. 

(4) As a result of the current distribution de- 
scribed above, the radiation produced by 
the vertical sections adds together, 
whereas the radiation produced by the 
horizontal sections is canceled out. Con- 
sequently, strong vertically polarized 
radiation results. 

(5) The Bruce array functions are similar to 
those of a simple broadside array made 
up of six vertical elements. However, 
since the vertical elements are only a 
quarter-wavelength instead of the usual 
half-wavelength, considerably less gain 
is obtained than from an ordinary six- 
element broadside array. As a result, 
the array must be made at least several 
wavelengths in order to produce a worth- 
while gain. The Bruce array usually is 
fed at a current loop as in the illustration. 

e. Flat-Top Array. 

(1) The flat-top array (A of fig. 172) consists 



of two pairs of collinear elements, each a 
half-wavelength, mounted in a horizontal 
plane and separated from each other by 
1 quarter- to 1 eighth-wavelength. Be- 
cause of the transposed transmission line 
connecting these pairs, the direction of 
current flow in one pair of collinear 
elements is opposite to that in the other 
pair. This is indicated by the small 
arrows in A. The array is not only 
collinear but is also end-fire, as indicated 
by the two large arrows which represent 
the directions of maximum radiation. 
These directions are in the plane of the 
elements. 




SOLID PATTERN 

e 



TM 666-176 

Figure 172. Flat-top array. 

(2) The gain of the array shown is under 6 db 
when 1 -quarter-wavelength spacing is 
used. With eighth-wave spacing, the 
gain increases slightly. The radiation 
pattern produced by this array is bi- 
directional in both horizontal and vertical 
planes. The beam width is narrower in 
the horizontal plane, as shown by the 
solid radiation pattern in B. 

197 



(3) The input impedance of the array is 
several thousand ohms. Therefore, reso- 
nant feeders generally are used. If a 
nonresonant line is to be used, a matching 
is required and the feeder must be con- 
nected properly on the matching section. 



(4) The flat-top array can be constructed also 
of two half-wave elements instead of four. 
Here the array is not collinear, but is 
merely a close-spaced end-fire array. The 
gain is about 3 to 4 db and the input 
impedance is a low value. 



Section III. PARASITIC ARRAYS 



105. Parasitic Elements 

a. General. 

(1) Parasitic arrays represent another method 
of achieving high antenna gains. A par- 
asitic array consists of one or more para- 
sitic elements placed in parallel with each 
other and, in most cases, in the same 
line-of-sight level. The parasitic ele- 
ment is fed inductively by radiated 
energy coming from the driven element 
connected to the transmitter. It is in 
no way connected directly to the driven 
element. 

(2) When the parasitic element is placed so 
that radiation is in the direction shown 
in A of figure 173, the element is a 
director. When the parasitic element is 
placed so that radiation is in the direction 
shown in B, the element is a reflector. 

(3) The directivity pattern resulting from the 
action of parasitic elements depends on 
two factors. These are the tuning, 
determined by the length of the parasitic 
element and the spacing between the 
parasitic and driven elements. To a 
lesser degree, it also depends on the 
diameter of the parasitic element, since 
diameter has an effect on tuning. 

b. Operation. 

(1) When a parasitic element is placed a 
fraction of a wavelength away from the 
driven element and is of approximately 
resonant length, it will reradiate the 
radiated energy it intercepts. The par- 
asitic element is effectively a tuned 
circuit coupled to the driven element 
much as the two windings of a trans- 
former are coupled together. The radi- 
ated energy from the driven element 
causes a voltage to be developed in the 
parasitic element which, in turn, sets up 
a magnetic field. This magnetic field 
extends over to the driven element, 



which then has a voltage induced in it. 
The magnitude and phase of the induced 
voltage depend on the length of the 
parasitic element and the spacing be- 
tween the elements. In actual practice, 
the length and spacing are arranged so 

i RADIATED 
f SIGNAL 



DIRECTOR 

DRIVEN 
ELEMENT 




FEEDER 



. RADIATED 
>' SIGNAL 



/Z 



FEEDER 



"^V. DRIVEN 
ELEMENT 



REFLECTOR 



3 



B 



TM 668-177 

Figure 17S. Position of the director and reflector with reference 
to the driven element. 

that the phase and magnitude of the 
induced voltage cause a unidirectional, 
horizontal radiation pattern, with an 
increase in gain. 
(2) In the parasitic array in B of figure 173, 
the reflector and driven elements are 
spaced a quarter-wavelength apart. The 
radiated signal coming from the driven 
element strikes the reflector after a 
quarter-cycle. The voltage developed in 



198 



the reflector is 180° out of phase with the 
driven element voltage. The magnetic 
field set up by the reflector induces a 
voltage in the driven element a quarter 
of a cycle later, since the spacing between 
the elements is a quarter-wavelength. 
The induced voltage is in phase with the 
driven element voltage, causing an in- 
crease in voltage in the direction of the 
radiated signal indicated. This forms 
the horizontal pattern in B of figure 174. 
(3) Because the voltage induced in the para- 
sitic element is 180° out of phase with 
the signal produced by the driven ele- 
ment, there is a substantial reduction in 
signal strength behind the reflector. In 
practice, since the jnagnitude of an in- 
duced voltage never quite equals that of 
the inducing voltage even in very closely 
coupled circuits, the energy in the minor 
lobe is not reduced to zero. In addition, 
very little radiation is produced in the 

HORIZONTAL PATTERNS 



t RADIATED 
SIGNAL 




PATTERNS OBTAINED 



direction at right angles to the plane of 
the elements. 

(4) It can be shown also that when the para- 
sitic element is a director, the horizontal 
and vertical radiation patterns will be 
as in A, figure 174. 

(5) The radiation patterns shown have several 
advantages and disadvantages. The two 
main advantages of a parasitic array are 
increased gain and unidirectivity. There 
is a reduction of transmitted energy in 
all but the desired direction. This makes 
the parasitic array useful in antenna 
systems that can be rotated to a given 
direction. These are known as rotary 
arrays. Size for size, the gain and 
directivity of a parasitic array are 
greater than for a driven array. The 
disadvantages of these arrays are that 
their adjustment is critical and they do 
not operate over a wide frequency range. 

VERTICAL PATTERNS 



t RADIATED 
SIGNAL 




X 



DIRECTOR 




TM («6-tTS 

Figure 174- Pattern* obtained using a parasitic element. 



199 



c. Relative Power Gain. 

(1) Figure 175 is a graph of the relative power 
gain for various element spacings. The 
curve was plotted under the special con- 
dition that both elements are a half- 
wavelength long. The dotted curve 
shows the power gain when a reflector 
is used as the parasitic element. The 
zero reference line shows the power gain 
from the driven element alone. 



(3) The gain of a parasitic array depends on 
the length of the parasitic element as 
well as the spacing between elements. 
Figure 176 shows the gain when the 
length of the parasitic element is ad- 
justed to obtain the most possible gain 
at any given spacing. There is, then, 
an increase in gain at any spacing be- 
tween elements. 




-3 



-4 1 I I I I I I I I I I I I I I 

.05 .1 .15 .2 .25 .3 .35 
ELEMENT SPACING (WAVELENGTHS) 

TM 666-179 

Figure 175. Relative gain for various spacings between 
elements. 

(2) From the curves, it is seen that when the 
director is placed about 0.1 wavelength, 
or when the reflector is placed about 0.2 
wavelength from the driven element there 
is power gain of 3 to 4 db. The curves 
also show that the director provides a 
greater power gain with less spacing be- 
tween elements than does a reflector. 
However, the curves show that the gain 
does not change as rapidly when the 
spacing of a reflector is changed. Con- 
sequently, with only a single parasitic 
element the reflector is used, since its 
spacing is not so critical. 




-1 .51 1 1 1 1 1 1 1 1 1 1 1 I l_J 

.05 .1 .15 .2 .25 .3 .35 

ELEMENT SPACING (WAVELENGTHS) 

TM 666-180 

Figure 176. Relative gains of a parasitic array for variou* 
spacing between elements when the parasitic element is 
adjusted for greatest gain. 

(4) In figure 176, it can be seen that the maxi- 
mum gain with a director is almost 6 db 
at a spacing of slightly more than 0.1 
wavelength. With a reflector, the maxi- 
mum gain is 5.5 db at a spacing of 0.15 
wavelength. The same power gain is 
obtained when the director is placed 0.15 
wavelength from the driven element. 
Comparison of figures 175 and 176 indi- 
cates that more gain is obtained at any 
spacing when the length of the parasitic 
element is changed by tuning. 
d. Front-to-Back Ratio. 

(1) The front-to-back ratio of an array is the 
proportion of energy radiated in the 
principle direction of radiation (or re- 
ception) to the energy radiated (or re- 
ceived) in the opposite direction. It is 
desirable to have a high value of front-to- 



200 



back ratio because this means that the 
minimum amount of energy is radiated 
in the undesired direction. Since it is 
impossible to suppress all such radiation 
completely, an infinite ratio cannot be 
achieved, but rather high values can be 
attained in practice. It is usual to 
adjust the length and spacing of the 
parasitic element so that a maximum 
front-to-back ratio is obtained rather 
than maximum gain in the desired direc- 
tion. 

(2) In general, as the length of the parasitic 
element is reduced from a half-wave- 
length, it has a greater capacitive react- 
ance. As the length is increased from a 
half-wavelength, it has a greater induc- 
tive reactance. The variation of the 
front-to-back ratio for different reactance 
values of parasitic elements is shown in 
figure 177. Here the front direction is 
assumed to be in a line from the parasitic 
element to the driven element. The 
spacing between elements in this partic- 
ular case is held constant at 0.1 wave- 
length. At all negative reactance values 
(capacitive reactance) and at low values 
of positive reactance up to about +8 
ohms, the parasitic element acts as a 
director with this value of spacing. When* 
the inductive reactance increases beyond 
+8 ohms, as the parasitic element length 
is increased, this element acts as a re- 
flector and the front-to-back ratio is 0.5. 
When the length of the parasitic element 
equals that of the driven element — that 
is, when both are tuned to the resonant 
frequency — the parasitic element is purely 
resistive and its reactance is zero. The 
parasitic element then behaves as a direc- 
tor and only half as much energy is 
radiated from the parasitic element 
toward the driven element as is radiated 
in the opposite direction. When the 
reactance of the parasitic element is +8 
ohms, the front- to-back ratio is 1. This 
means that there is equal radiation in 
both directions and the parasitic element 
behaves neither as a director nor as a 
reflector. 

(3) Tuning a parasitic element for a maxi- 
mum front-to-back ratio does not mean, 
necessarily, that the array provides a 



maximum power gain. With the director 
that is tuned for maximum power gain 
(6 db, fig. 176), the resonant length of 
the director can be changed by experi- 
menting so that the front-to-back ratio 
is increased 10 times with a reduction in 
the power gain of less than 1 db. 



3.0 



2.5 



S 20 



1C 

CD 
I 

g 
• 1 
t- 

z 

o 



1.5 



.0 

















1/ 


s 


























\ 


























v 
































































f 






































0IR 


ECT 


OR- 








IEFI 


-EC" 


roF 














i— 











































































-60 -40 



-20 



+20 +40 +60 



CAPACITIVE 
-REACTANCE- 
(0HMS) 



INDUCTIVE 
-REACTANCE 
(OHMS) 



TM 66S-I8I 



Figure 177. The front-to-back ratio of a parasitic array with 
varying reactance of the parasitic element. 

e. Radiation Resistance. 

(1) The elements of a parasitic array must 
be spaced a fraction of a wavelength 
from each other to obtain high power 
gain and high front-to-back ratio. When 
this is done, however, the radiation re- 
sistance of the driven element is altered 
considerably. The closer the spacing 
between elements, the smaller the radia- 
tion resistance becomes; the greater the 
spacing, the larger the radiation resist- 
ance (approaching the free-space value). 
The resulting low values of radiation 
resistance in parasitic arrays have sev- 
eral effects, not only limiting the number 
and type of feeding systems that can be 
used, but also reducing the radiation 
efficiency of the array. 

(2) The reason for the low values of radiation 
resistance in parasitic arrays is explained 
as follows: Since both elements are 
placed extremely close to one another, 
the voltage induced back in the driven 



201 



element as a result of the parasitic ele- 
ment is almost equal in magnitude to the 
original driven element voltage. Since 
these voltages are out of phase with each 
other, a cancelation effect exists at the 
driven element. This reduces the voltage 
at the driven element which, therefore, 
causes a reduction in its feed-point re- 
sistance with the same input power. The 
closer the elements, the greater is the 
cancelation effect, and the lower is the 
radiation resistance. 

(3) The manner in which radiation resistance 
varies with different element spacings is 
shown in figure 178, where both driven 
and parasitic elements are resonant. 
Note from the curve that the radiation 
resistance increases as either the reflector 
or the director is moved farther away 
from the driven element. The lowest 
radiation resistance is slightly greater 
than about 8 ohms when the elements 
are very close together. For spacings as 
great as 0.3 wavelength, when a reflector 
is used, the radiation resistance goes as 
high as 66 ohms. When a director is used 
at this spacing, the resistance is about 
56 ohms. This indicates that larger 
radiation resistance values can be ac- 
quired when using the parasitic element 
as a reflector. 

(4) The maximum power gain with a reflector 
and a director occurs at element spacing 
of 0.2 and 0.1 wavelength, and the para- 
sitic elements are the same length as the 
driven element. At these spacings, the 
radiation resistance values with the re- 
flector and director are 36 ohms and 14 
ohms, respectively. These values are 
very small compared with a radiation re- 
sistance of about 73 ohms for the driven 
element alone. So, although the power 
gain of a parasitic array is large, the 
radiation efficiency may be low. In the 
selection of the type of array that is to be 
used for an antenna system, it first must 
be determined whether maximum power 
gain or radiation efficiency is desired. 
For maximum radiation efficiency, the 
driven array is used, and for maximum 
power gain the parasitic array is used. 





























































64 












y 

L 










w 
























X 






















- U 


48 
















v 






a 
z 






















? 

«i 


40 




















</> 
bJ 


















u 




- a 
z 


32 


















v 




inn 


















\ 




«i 

5 






















K 


ti 






— 

J 














\ 




•16 






/ 
























f 






















/ 












-Dl 


*EC 


TOF 




-8 








•CTl 


)R- 










-Rl 












- 4 















.3 .2 .1 .1 .2 .3 



ELEMENT SPACING (WAVELENGTH) 

TM 666-182 

Figure 178. Curve showing radiation resistance at the driven 
element for various spacings of driven and parasitic 
elements. 

f. Radiation Patterns. 

(1) The radiation patterns of a parasitic 
array depend on the spacing between 
elements and the length of the parasitic 
element. The reactance reflected by the 
parasitic element is a direct function of 
its length. When the parasitic element 
is longer than a half-wavelength, it be- 
haves as an inductive reactance. The 
induced parasitic voltage leads the in- 
duced current, and the parasitic phase 
angle is positive. When the parasitic 
element is shorter than a half-wavelength, 
it behaves as a capacitive reactance. 
The induced parasitic voltage lags the 
induced current, and the parasitic phase 
angle is negative. These angles are indi- 
cated in figure 179, which shows several 
radiation patterns in the horizontal (xz) 
plane for various element spacings and 
parasitic phase angles. The power input 
to the driven element is the same in each 
case. The radiation resistance is shown 
above each pattern. 



DRIVEN 
ELEMEN1 



RADIATED. 
SIGNAL 



I — REFLECTOR 



-S-U--X 

I 
I 

Z 



DRIVEN 
ELEMENT" 



-DIRECTOR 



- -6- -l- -6- -X 



RADIATED 
SIGNAL 



23.8 OHMS 11.7 OHMS 14.9 OHMS 31.6 OHMS 54.3 OHMS 

, ■ 0- -B- 



tn 
x 

H 
O 

z 

UJ 



\ 



\ 



41.6 OHMS 25.6 OHMS \t9.5 OHMS 25.6 OHMS 41.6 OHMS 



REFLECTOR 



\ 

£ ....... \ 



Z 

o 
< 



64 OHMS 9 OHMS 



37 OHMS \ 33.3 OHMS 
\ 



.25 



DIRECTOR 



40 OHMS 



i'-Q- Or o?- w 

tu \ 

2 \ 

-J 80.6 OHMS 70.9 OHMS 58.8 OHMS 49.9 OHMS 48.7 OHMS 
Ul 



0- -0- 0- 

+ 40° +20° 0° -20° -40° 



ANGLE BY WHICH INDUCED PARASITIC VOLTAGE LEADS OR LAGS INDUCED CURRENT 



TM 666-183 

Figure 179. Radiation patterns of a parasitic array for various element spacings and phase angles. 



(2) At 0.15-wavelength element spacing and a 
phase angle of 0° (with parasitic element 
resonant), the radiation pattern is bi- 
directional. When the spacing is 0.15 
wavelength and the angle is +20°, the 
parasitic element acts as a reflector, but 
at a spacing of 0.15 wavelength and an 
angle of —20°, the parasitic element acts 
as a director. The diagonal dotted line 
indicates the dividing line between re- 
flector and director action. If the spac- 
ing between elements is other than 0.15 
wavelength, making the length of a para- 
sitic element shorter than a half-wave- 
length (resonant length) does not neces- 
sarily make it behave as a director, as 
shown by the pattern obtained at a spac- 
ing, of 0.25 wavelength and a phase angle 
of —20°. Since the angle is negative, 



the parasitic element is shorter than its 
resonant length which ordinarily makes 
it appear as a director. However, the 
radiation pattern indicates that it ac- 
tually behaves as a reflector. 
g. Frequency Response. 

(1) As mentioned previously, the parasitic 
array has a comparatively narrow fre- 
quency range. Well outside this range, 
the effect of the parasitic element is small 
and the driven element alone controls 
the over-all characteristics. Figure 180 
shows the frequency response of a driven 
element alone contrasted with a parasitic 
array using a director and a reflector. 
The reflector curve is that obtained when 
the length of the reflector produces max- 
imum power gain with a spacing between 
elements of 0.2 wavelength. The di- 



203 



+6 

I +4 

Z +2 
< 

iij 

* -2 

> 
_i 

or ° 
-10 

-12 

































































































A 




V 




^-DIRECTOR 






















1 — 'i 


t 

/. 






\ 
























REFLECTOR- 








» — 




A 








































\ 


























P 
















\ 




















/,< 






-DRIVEN ELEMENT 
ONLY 






\ 


















7 

/ 

















































































.8 .9 1 1.1 

FREQUENCY RATIO ( fx /f ) 



12 



1.3 



1.4 



1.5 



TM 666-184 

Figure 180. Frequency coverage of a driven element contrasted with an array using a director and a reflector. 



rector curve is that obtained when the 
physical length of the director produces 
maximum power gain, with the spacing 
between elements 0.1 wavelength. The 
diameter of the elements is about 0.006 
wavelength. Ratio J x /f is the frequency 
applied to the array compared with the 
resonant frequency of the driven element. 

(2) In the curve showing the frequency range 
of the driven element alone, a maximum 
power gain of db (reference gain) is 
obtained when /*//„ equals 1; that is, 
when the frequency of the radiated signal 
equals the resonant frequency of the 
driven element. As the frequency of the 
applied signal decreases, the j x \j ratio 
decreases, and the relative power gain 
decreases. As the frequency of the 
applied signal increases, the j x \j ratio 
increases, and the relative power gain 
decreases. In other words, a decrease 
or an increase in the applied frequency 
causes a decrease in the power gain. 
This curve tapers off gradually as the 
frequency is changed, which indicates 
that the frequency response of the driven 
element alone is fairly broad. 

(3) Consider the reflector curve. Maximum 
power gain ( + 5 db) results when the 
applied frequency equals the resonant 
frequency of the driven element. How- 
ever, the power gain decreases sharply 
as the applied frequency is increased or 
decreased . This indicates the narrow fre- 



(4) 



quency response. When the applied fre- 
quency is increased by only 10 percent 
(Jxlfo— 1-1), the power gain drops 2 db — 
that is, from +5 to +3 db. When the 
applied frequency is decreased 10 percent 
(fx/f o=0.9), the power gain drops 4.5 db, 
from +5 to +0.5 db. At the ratios 
of 0.88 and 1.23, there is a no-power gain 
compared with that of the driven ele- 
ment acting alone. The points where 
the driven element curve and the reflector 
curve intersect (J x /f = 0.84 and 1.45) in- 
dicate that the same power gain results 
at this frequency ratio regardless of 
whether the reflector is used in the array. 
At all ratios lower than /r// o =0.84, a 
greater power gain is obtained when using 
the driven element alone. 
In the director curve, a similar action 
results. When/,//,, ratio equals 1, max- 
imum power gained is obtained. As the 
frequency ratio increases or decreases, 
the curve drops off rapidly, showing a 
sharp decrease in power gain. The direc- 
tor curve tapers off much more rapidly 
for ratios greater than 1 compared to the 
reflector curve. With ratios less than 1, 
the curve tapers off slightly less rapidly 
than the reflector curve. Therefore, the 
frequency response of the array with a 
director is very much narrower than that 
of the array with a reflector. 



204 



1 06. Multiparasitic Arrays 

a. A multiparasitic array is one which contains 
two or more parasitic elements with the driven ele- 
ment. If the array contains two parasitic elements 
(a driven element, a reflector, and a director), it 
usually is known as a three-element beam. If three 
parasitic elements are used, the array then is 
known as & Jour- element beam, and so on. Gen- 
erally speaking, if more parasitic elements are 
added to a three-element beam, each added ele- 
ment is a director. For example, a five-element 
beam contains one driven element, one reflector, 
and three directors. Most parasitic arrays do not 
use a greater number of elements than this. 

b. The parasitic elements of a multiparasitic 
array usually are positioned as shown in figure 181. 
A shows a three-element beam, B a four-element 
beam, and C a five-element beam. Although the 
spacing between elements is typical of those nor- 
mally encountered, many variations may be found. 
Frequently, the best spacings are found experi- 
mentally. In a three-element beam, the director 
usually is made slightly shorter and the reflector 
is made slightly longer than the driven element. 
The length of jthe second director in a four-element 
beam usually is shorter than the one nearest the 
driven element, and each additional director is 
made shorter than the previous one. A folded 
dipole can be used as the driven element to obtain 
greater values of radiation resistance. 

c. The reason for using a multiparasitic array is 
to obtain greater power gain and unidirectivity. 
In addition, a larger front-to-back ratio can be ob- 
tained with proper parasitic tuning. In general, 
the more parasitic elements used, the greater is the 
power gain. However, a greater number of such 
elements causes the array to have a narrower 
frequency response characteristic and become 
critical to adjust. The gain of a parasitic array 
does not increase directly with the number of 
elements used. For example, the three-element 
beam shown in the figure has a relative power gain 
of 5 db. Adding another director results in only a 
2-db increase, or a total gain of about 7 db. If 
another director is added to the array, an increase 
of less than 1 db results, making the total gain less 
than 8 db. It is seen that the director closest to 
the driven element has the greatest effect on the 



gain; the one farthest away has the least effect. 
Consequently, as more directors are added, the 
effect on radiation resistance becomes smaller and 
smaller. Regardless of how many directors are 
added to the array, only a single reflector is used, 
because very little radiation goes behind this 
reflector. 



REFLECTOR 
R 



RAOIATEO SIGNAL 

ORIVEN 

ELEMENT DIRECTOR 
OR 
-.1-1 5> — 



-.IV- 



OR 



01 



02 OS 
»» « , 1-.16> — *+ 



TM 666-185 

Figure 181. Examples of spacing between elements of multi- 
parasitic arrays. 

d. When there are two or more parasitic ele- 
ments in a multiparasitic array, it sometimes is 
referred to as a yogi array. A typical yagi array 
used for receiving and transmitting radio energy 
is shown in figure 182. This antenna, used by the 
military services, operates at frequencies from 12 to 
50 mc and consists of two separate yagi arrays 
mounted on one frame (one high-frequency and 
one low-frequency antenna array.) The various 
elements are indicated in the figure. The high- 
frequency array consists of one reflector, one 
driven element and two directors; the low-fre- 
quency array has the same arrangement with one 
less director. The lengths of the elements in the 
high-frequency array are shorter than those in the 
low-frequency array. The physical lengths of the 
elements in the individual arrays are equal, but 
the electrical lengths can be varied by means of 
the tuning -stubs at the center of the elements. 
The array can be rotated in any desired direction 
by a remotely controlled, electrically driven, an- 
tenna rotator. Some of the characteristics of this 
array are given in the chart below. 



205 



Range 


Number 
of ele- 
ments 


Power 
gain 


Front-to 
back ratio 


Feeding of driven element 


1-f— ... 


3 


7db 


25 db 


52- to 600-ohm trans- 










mission line fed 










directly to T-match- 










ing section of driven 










element. 


h-f 


4 


9db 


30 db 


52- to 600-ohm trans- 










mission line using 










inductive, coupling 










to driven element. 



(2) A simple field intensity meter or high- 
input impedance vacuum-tube voltmeter 
may be used as the adjustment indicator. 

(3) No transmitter adjustments are necessary. 

(4) The r-f voltage induced in the antenna 
elements is of such low magnitude that 
the elements, including stubs and T- 
match sections, may be touched and 
handled without endangering personnel 
with r-f burns. 

(5) Fewer personnel are needed to make the 
adjustments. 





Figure 18$. A typical parotitic array used for receiving and transmitting. 



107. Tuning, Adjusting, and Feeding 

a. There are many methods of tuning and 
adjusting parasitic arrays. Regardless of which 
method is used, and whether it is to be used for 
transmitting or receiving, the parasitic array 
usually is set up as a receiving array when it is 
adjusted. Some of the reasons for doing this are 
given below. 

(1) The parasitic array is immersed in a 
radiated field of constant intensity which, 
in turn, gives truer indications. 



b. In many instances, the elements of a para- 
sitic array contain telescopic sections which permit 
the physical length of the elements to be length- 
ened or shortened. It is customary first to adjust 
physical lengths approximately before attempting 
final adjustments. The following formulas can be 
used: 

468 

length (in feet) of driven element==-y- 

492 

length (in feet) of reflector=— ^r- 

450 



length (in feet) of director =-j 



206 



where/ is in megacycles in all cases. As an exam- 
ple, if a three-element beam is to operate at a 
frequency of 100 mc, the approximate physical 
lengths of its driven element, reflector, and di- 
rector, respectively, are 468/100 or 4.68 feet, 492/ 
100 or 4.9 feet, and 450/100 or 4.5 feet. 

c. The method by which the driven element of a 
parasitic array is fed determines the method used 
in adjusting it. 

d. A typical arrangement used for tuning and 
adjusting the elements of a four-element beam is 
shown in figure 183. A half-wave exciting dipole 
is fed by a low-power transmitter. This dipole 
radiates energy to the parasitic array. The power 
output of the transmitter must be held constant 
during the procedure. The spacing between the 
parasitic array and the exciting dipole is 2 or more 
wavelengths. The driven element of the parasitic 
array is connected through a 600-ohm balanced 
line to a field intensity meter or a high-impedance 
vacuum-tube voltmeter by means of a ^-matching 
section. These meters need not be calibrated, 
since only relative values of r-f voltage are required 
to be measured. The lengths of all of the elements 
are adjusted electrically (tuned) by the adjustable 
tuning stubs. 



e. The method and points to remember in tuning 
and adjusting the array are as follows: 

(1) The height of the array should be a rea- 
sonable distance from the ground so that 
the element adjustments may be made 
conveniently. Very large arrays should 
be placed as much as 10 to 12 feet off the 
ground. In this case, the adjustments 
can be reached by a ladder. 

(2) The height of the exciting array must be 
the same as the parasitic array so that a 
constant large amount of radiated energy 
may be received by the array. 

(3) The low-power transmitter must be as 
close to the exciting dipole as possible. 
This means that the transmission line 
connecting the two must be as short as 
possible. In this way, it is less likely 
that the parasitic array will pick up radi- 
ation from the transmission line or from 
the transmitter directly. 

(4) The spacing between the exciting dipole 
and parasitic array must be 2 or more 
wavelengths for the reason given above. 
If the distance is made too great, on the 
other hand, the parasitic array might not 
pick up sufficient energy to actuate the 
meter. 



EXCITING 
DIPOLE 



2 OR MORE WAVELENGTHS 



REFLECTOR 

DRIVEN 



DIRECTOR 
I 





LOW- POWER 
TRANSMITTER 
10 TO 50 WATTS 



TUNING STUBS 
T- MATCHED SECTION 



FIELD INTENSITY METER 
OR 

HIGH-IMPEDANCE VTVM 



TM 668-187 

Figure 188. Typical arrangement for tuning and adjusting a four-element beam. 



207 



(5) The T-matching section and the tuning 
stub of the driven element are adjusted 
for a maximum meter reading, and the 
stub of the director element closest to the 
driven element is adjusted for maximum 
meter reading. The stub of director No. 
2 then is adjusted for a maximum reading. 



(6) The antenna is rotated 180° so that the 
reflector element is closest to the exciting 
dipole. The stub of the reflector then is 
adjusted for a minimum meter reading, 
to insure that the reflector produces a 
maximum attenuation in the unwanted 
direction of radiation. 



Section IV. SUMMARY AND REVIEW QUESTIONS 



108. Summary 

a. An array is a combination of half-wave ele- 
ments operating together as a single antenna. 
Arrays provide more gain and greater directivity 
than single-element antennas. 

b. In a driven array, all elements derive their 
power directly from the source. In a parasitic 
array, one or more (parasitic) elements derive 
power by coupling from another element or other 
elements. 

c. Arrays can be bidirectional or. unidirectional. 
A bidirectional array radiates equally in two 
opposing directions along a line of maximum 
radiation. A unidirectional array radiates well 
in a single direction. 

d. The elements in a collinear array lie in the 
same straight line, and maximum radiation occurs 
at right angles to this line. The currents in all 
elements must be in the same phase. 

e. The elements in a broadside array are all 
parallel and in the same plane. Maximum radia- 
tion develops in the plane at right angles to the 
plane of the elements. The currents in all ele- 
ments are in the same phase. 

f. The elements in an end-fire array also are 
parallel to each other and in the same plane, but 
maximum radiation occurs along the axis of 
the array. Currents in adjacent elements are 
never in the same phase. A 180° phase difference 
from element to element is most common and 
produces bidirectional radiation. A 90° phase 
difference results in a unidirectional pattern. 

g. At points distant from the array, radiation 
is either stronger or weaker than from a single 
element, depending on the direction of the point 
with respect to the array. At some angles, com- 
bined radiation from the various elements results 
in reinforcement; at other angles, combined 
radiation results in cancelation. The degree of 
cancelation of reinforcement is a function of the 
relative phase of radiation from different elements 
combining in space. 



h. Matching stubs or sections are used between 
elements to maintain current in the desired phase. 

i. The impedence of an antenna element acting 
alone (selfimpedance) differs from its impedance 
when it is acting with other elements in an array. 
Mutual impedance accounts for the difference 
between these two. 

j. When collinear elements are a half-wave- 
length, the length of the stub between them is a 
quarter-wavelength. When the elements are 
longer, the length of the stubs is reduced cor- 
respondingly. In any one array, all elements 
must be of the same length. 

k. The gain of a collinear array depends on the 
number of elements used and the space between 
elements. Gain is greatest when this spacing 
is 0.4 to 0.5 wavelength. Gain also increases as 
the number of elements is increased. 

I. Collinear antennas can be fed adequately 
by resonant lines. Feed usually is introduced 
at a point between the ends of two elements, 
where the impedance is in the order of 1,500 ohms. 
Feed also can be introduced at the center of one 
of the elements. For best balance, the trans- 
mission line is introduced at the center of the 
array. 

m. Optimum gain is obtained from broadside 
arrays when the elements are spaced 0.65 wave- 
length apart. However, half-wave spacing, which 
provides good gain, simplifies phasing. 

n. When a broadside array is stacked vertically, 
height requirements usually limit the number of 
elements used; when the dipoles are vertical, 
more than six elements rarely are used. 

o. There are two principal means for bringing 
currents into broadside elements in the same 
phase. When the phasing line between dipoles 
is a half-wavelength, alternate elements are con- 
nected to opposite sides of the line. When the 
lengths are 1 wavelength, all elements are con- 
nected to similar sides of the line. 



208 



p. The optimum spacing between end-fire 
elements is an eight-wavelength. 

q. The radiation resistance of end-fire arrays is 
very low. 

r. Radiation resistance can be increased and 
feeding can be simplified by the use of folded 
dipoles and impedance step-up matching devices. 

s. The extended double-Zepp antenna consists 
of a two-element collinear array in which the 
length of each element is somewhat longer than 
a half-wavelength. 

t. The lazy-H antenna array consists of two 
collinear arrays connected in parallel and spaced 
a half -wavelength apart in the vertical plane. 

u. The Sterba curtain consists of two collinear 
arrays, usually made up of a large number of 
elements, stacked one above the other at a spac- 
ing of a half-wavelength. The form of the array 
is such that a closed loop is formed by the con- 
ductors making up the array. 

v. The Bruce array consists of a single wire 
folded into horizontal and vertical quarter-wave 
sections. 

w. The flat-top array consists of two pairs of 
collinear elements, mounted in a horizontal plane 
and separated from each other by a small spacing 
(approximately an eighth- to a quarter-wave- 
length). 

x. A parasitic array consists of one or more 
parasitic elements along with a driven element. 

y. The director usually is made shorter and the 
reflector usually is made longer than the driven 
element to obtain a large power gain. 

z. The amount of power gain and directivity 
of a parasitic array depends on the lengths of the 
parasitic elements and the spacing between 
elements. 

aa. When the length of a parasitic element 
equals that of the driven element, a maximum 
power gain exists when the parasitic element is 
used as a director and is placed 0.1 wavelength 
from the driven element, or when it is used as a 
reflector and placed 0.2 wavelength from the 
driven element. 

ab. When the length of a parasitic element is 
tuned for a maximum power gain at any given 
spacing between elements, it is found that maxi- 
mum power gain for the director results at 0.1 
wavelength spacing and for the reflector at 
0.15 wavelength spacing. 

ac. Some power gain must be sacrificed when a 
parasitic array is tuned to obtain a maximum 
front-to-back ratio. 



ad. The radiation resistance of the driven ele- 
ment of a parasitic array is very much less than 
that of a driven array. This, in turn, reduces its 
radiation efficiency. 

ae. The parasitic array has a comparatively 
narrow frequency range of optimum operation. 

af. The frequency response of a parasitic array 
using a reflector is wider than the frequency re- 
sponse of an array using a director. 

ag. Multiparasitic arrays are used to obtain 
greater power gain, directivity, and front-to-back 
ratios. 

ah. When tuning and adjusting a parasitic 
array, it usually is set up as a receiving array. 

ai. The approximate physical lengths of the 
elements of a parasitic array are as follows: 

length (driven element) = 468 

/ 

length (reflector) =492 
/ 

length (director) =450 

T~ 

where the length is in feet and / is in mc. 

aj. When tuning a parasitic array, the elements 
are tuned in this order: the driven element, the 
director nearest the driven element, all other 
directors, and the reflector. 

109. Review Questions 

a. What is an array? A driven array? A 
parasitic array? 

b. What are the advantages and disadvantages 
of arrays as compared with other antennas? 

c. Describe a collinear array, with special atten- 
tion to the following features: arrangement and 
number of elements, phase relationship, radiation 
pattern. 

d. In what ways is a broadside array likely to 
differ from an end-fire array in appearance? 

e. Explain the difference in operation between 
broadside and end-fire arrays. Compare them as 
to the following features: plane of directivity, 
radiation resistance, phase relationship, spacing, 
feeding. 

/. What is mutual impedance? 

g. What is the gain of three collinear elements 
with negligible spacing between the elements? 

h. What is the gain of two collinear elements 
with 3-quarter-wavelength center-to-center spac- 
ing? 

209 



i. What is the radiation resistance of two col- 
linear elements with 0.35-wavelength spacing 
between their adjacent ends? 

j. What is the best point in a collinear array at 
which to introduce the feed? Why? 

k. For purposes of calculating reflection effects, 
what is considered to be the height of a broadside 
array above ground? 

I. Two parallel elements, half a wave apart, 
must be phased for broadside operation. Describe 
two methods by which this can be done. At what 
points can the transmission line be introduced? 

m. Three parallel elements with half-wavelength 
spacing must be phased for broadside operation. 
Describe one method by which this is done. Do 
not use a method explained in answer to the 
previous question. 

n. Explain how four parallel elements must be 
phased for broadside operation when full-wave 
spacing is used. 

o. With what spacing will a broadside array be 
most efficient? 

p. Explain two methods for phasing a pair of 
parallel elements for bidirectional end-fire opera- 
tion. Choose one method using center feed. 

g. At what spacing do end-fire elements provide 
the greatest gain? What problems arise at this 
spacing and how can they be overcome? 

r. What is the input resistance of a pair of end- 
fire elements operated 180° apart with quarter- 
wavelength spacing? 

s. What is the optimum length of the elements 
in the extended double-Zepp antenna? 

t. Why is the lazy-H antenna array so named? 



if. To what use can the closed-loop structure of 
the Sterba curtain be placed? 

v. Why does the Bruce array illustrated in 
figure 171 produce a vertically polarized radio 
wave? 

w. Describe the radiation pattern produced by 
the flat-top array. 

x. What is a parasitic array? 

y. Why is the physical length of a director 
shorter, and that of a reflector longer than the 
length of the driven element? 

z. Name three factors that the gain and direc- 
tivity of a parasitic array depend upon. 

aa. Compare the characteristics of a parasitic 
array with those of a driven array. 

aft. When a parasitic array is tuned for maxi- 
mum power gain, is the front-to-back ratio also a 
maximum? 

ac. Why is the radiation resistance of a parasitic 
array always lower than that of a driven array? 

ad. What is a two-element beam? A three- 
element beam? A four-element beam? 

ae. What are the advantages of using multi- 
parasitic arrays? 

af. Name the common methods for varying the 
length of parasitic elements. 

ag. Why are parasitic arrays operated usually 
as receiving arrays when they are adjusted? 

ah. What are the formulas used to determine the 
approximate physical lengths of a director, a 
reflector, and a driven element? 

ai. Give the order in which the elements of a 
five-element beam are tuned. 



210 



CHAPTER 6 
RADIO DIRECTION FINDING ANTENNAS 



110. Directional Reception 

Since the early days of radio, it has been known 
that a radio wave, when traveling between a 
transmitter and a receiver, almost always follows 
the shortest line along the surface of the earth, 
the Great Circle Path. For example, if a radio 
station located in New York City is radiating 
energy, that portion of the energy picked up by a 
receiver in Los Angeles, Calif., will travel diago- 
nally across the United States. Thus, theproblem 
of radio direction finding is reduced to that of 
determining the direction of arrival of a radio 
wave at the receiver. This determination is ac- 
complished by using a directional receiving anten- 
na, such as a loop or an Adcock antenna. 

111. Loop Antenna 

a. General. A loop antenna consists of one or 
more turns of conductor, either self-supporting or 
wound on an insulated frame. The most com- 
monly used styles are diamond-shaped, square 
(fig. 184), or circular loops. These antennas all 
have a figure-8 response pattern. Thus, if a small 
transmitter is moved about a loop, unequal 
responses will be obtained at equal transmitter 
distances. Conversely, if a loop is rotated about 
its central vertical axis and the transmitter is held 
in one position, unequal responses will be obtained 
at different loop positions. 

(1) The output voltage of these antennas is 
the result of phase differences between 
the voltages induced in opposite sides of 
the antenna. Consider that the single- 
turn loop of figure 185 is placed in the 
path of a vertically polarized wave. 
When the plane of the loop is in line with 
the direction of wave travel, as indicated, 
the wave front (fig. 11) reaches the sides 
of the loop at slightly different times 
(one side of loop is nearer the transmitter 
than the other side), producing a phase 



difference between the voltages induced 
in the two sides and giving rise to a 
resultant voltage across the antenna 
coupling coil. The resultant voltage is 
maximum when the plane of the loop is 
in line with the direction of wave travel. 
No voltage is induced in the horizontal 
sides of the loop because the vertically 
polarized wave travels parallel to them. 
(2) If the loop then is rotated about its 
central vertical axis until the plane of the 
loop is perpendicular to the direction of 
wave travel, the wave front will reach the 
sides of the loop at the same time, the 
voltages induced in the two sides will be 
equal in magnitude and of the same 
phase, and, being directed in opposition 
through the coupling impedance, will 
cancel, thus resulting in a minimum 
output. The point of minimum response 
is called a null, 
b. Pattern with Normal Polarization. When the 
incoming radio waves have vertical polarization, 
which is the normal condition under which a 
vertical loop would be used for direction finding, 
the loop antenna has & figure-8 response pattern 
(fig. 186). In this illustration, the loop appears 
in the 90° to 270° position; any signal received 
from either of these directions will induce maxi- 
mum signal into the receiver. As the loop is 
turned away from the direction from which the 
wave is arriving (90° to 270° position), the 
received signal decreases, reaching a minimum 
when the loop is in either the 0° or the 180° 
position. The line of direction of a transmitter 
can be determined by rotating the loop about its 
vertical axis until either a null or a maximum 
signal is produced. Then the transmitter direc- 
tion is broadside to the loop at the null, or edge- 
wise to the loop at the maximum, and the appro- 
priate direction (azimuth) can be indicated by a 
pointer attached to the loop. It is customary to 



211 



TM 666H89 



Figure 184. Loop antenna. 



use the minimum rather than the maximum output 
of a loop when finding ah azimuth. This permits 
a sharply defined indication and greater accuracy. 
The response pattern in figure 187 shows why this 
is so; maximum response of 100 microvolts is 
obtained with the loop edgewise to direction A 
(toward the transmitter) . With the loop pointing 
toward B, a 10° rotation, there is only a 1.5-micrb^ 



volt change in signal strength; this difference is 
not noticeable in the output. But with the loop 
broadside to a transmitter at D, a null position, a 
similar 10° rotation of the loop causes a 17.4- 
microvolt change in signal intensity. 

c. Pattern with Abnormal Polarization. Since 
vertical polarization is considered normal for loop 
operation, any wave containing a horizontally 



212 




(•-AXIS OF 
ROTATION 



ANTENNA 
COUPLING COIL 

c 



/ TO RECEIVER 
y - INPUT CIRCUIT 



TM 666-190 

Figure 186. Simple loop antenna circuit, schematic diagram. 

polarized component is considered as being abnor- 
mally polarized. A horizontally polarized wave 
has no effect on vertical conductors, but it will 
induce, voltage in horizontal conductors such as 
those at the top and bottom of a loop. The 
response -pattern oj a loop for horizontal polarization 
is a figure 8, with maximum broadside and null 
endwise to the horizontal conductors. If the wave 
travels exactly horizontally, voltages induced in 
the top and bottom conductors will be equal and 
in phase; they will cancel each other in the loop 
output, just as do the voltages induced in the two 
sides by a (vertically polarized) wave arriving 
from the normal null direction (broadside) . Thus, 
the horizontally polarized component of a wave 
produces little or no loop output when arriving at 
a low angle, but its effect becomes important when 
the wave is steeply downcoming (or upcoming). 
There will be an error of ±90° in the azimuth 
determined by a simple loop direction finder if the 
polarization is assumed to be vertical when it 



270 

MAXIMUM — - 
VOLTAGE 




actually is horizontal. If the incoming wave has 
both vertically and horizontally polarized com- 
ponents, the figure-8 responses to the two compo- 
nents may combine to produce a figure 8 with 
sharp nulls, rotated in azimuth by some angle less 
than 90° from the normal position; this happens 
when the two components produce output voltages 
either in phase or 180° out of phase. When the 
two components produce voltages having other 
phase relations, the resulting response pattern 
may have its nulls filled in so that they become 
broad minima rather than zeros. In the special 
case where the two components produce equal 
output voltages 90° out of phase, the resulting 
response pattern becomes a perfect circle. 

d. Ambiguity. Unless the general direction of 
the transmitter is known, a direction finder 
equipped with a simple loop antenna cannot 
determine whether a transmitter lies forward or to 
the rear of the direction finder. This 180° 




98.5 UV 



Figure 186. Loop antenna, figure-8 response pattern. 



TL 9711 



Figure 187. Change in loop position versus change in signal 
voltage. 

ambiguity is caused by the two null positions of 
the loop, both of which indicate the same line of 
direction. There is no indication as to which of 
the two is the correct azimuth. 

112. Loop and Sense Antennas 

a. Purpose. The sense antenna is usually a 
vertical whip or monopole placed at the vertical 
axis of the loop. It is omnidirectional in azimuth. 
Both the circular response pattern of the sense 
antenna and the figure 8 pattern of the loop are 
symmetrical; but, when properly combined, the 
two antennas produce a lopsided or unidirectional 
pattern (cardioid pattern in fig. 188). The big 
end of this pattern lies to the right of one maxima, 
and to the left of the other maxima of the figure 8 



213 




Figure 188. Cardioid response pattern obtained by com- 
bination of circular and figure 8 voltages. 



pattern. By observing the relative positions of 
the unidirectional pattern, the two nulls can be 
distinguished, thus resolving the 180° ambiguity 
of the simple loop. One null is designated arbi- 
trarily the front or direction null; the other is 
called the back or reciprocal null. 

b. Application. After the loop has been rotated 
to a null, the direction it faces may be read from 
an azimuth scale or observed directly. Then the 
sense antenna is placed in operation, changing the 
response pattern from figure 8 to unidirectional. 
The null vanishes because of this change, and, 
upon turning the loop 90° to either side from the 
former null position, the response is found to be 
greater on one side than on the other. Which side 
is greater depends on whether the loop originally 
has its direct null facing toward or away from the 
transmitter. - As a rule, if the response increases 
as the loop is turned clockwise, which increases 
the azimuth scale reading, or if the response 
decreases as the azimuth reading decreases, the 
direct null was toward the transmitter; if the re- 
sponse changes in the opposite direction, the 
reciprocal null was toward the transmitter. This 
relation is not always used. It can be reversed 
by transposing connections in the antenna circuit, 
to reverse the relative polarity of loop and sense 
antennas, or by a 180° shift in position of azimuth 
scale or pointer relative to the loop, and there are 
cases in which such a reversal has been made 
intentionally. 



1 1 3. Cardioid Theory 

When the voltages from loop and sense antennas 
are combined with the proper phase and amplitude 
the resulting pattern is a heart-shaped curve known 
to mathematicians as a cardioid. A tvpical case 
is illustrated in figures 189 and 190, and is ex- 
plained, step by step, as follows: 

a. A radio wave, traveling past the loop, as 
indicated in A of figure 189, strikes leg No. 1 a 
short time before it strikes leg No. 2. 

b. The voltages induced in the two vertical legs 
are connected in series opposition, so that the net 
output of the loop depends on their difference. 

c. As shown in B of figure 189, the voltage in 
leg No. 1 is starting to rise at time zero (< ) ; the 
voltage induced in leg No. 2 starts to rise a short 
time later (2 2 ). However, so far as the output of 
the loop is concerned, the voltage induced in leg 
No. 2 is out of phase and begins to subtract from 
the voltage in leg No. 1 at this time (fe). 

d. Resultant voltage E x is developed across the 
output of the loop. This voltage is directly pro- 
portional to the time delay (phase shift) between 
the voltages induced in \the legs of the loop. The 
greater the separation between t and t 2 in B of 
figure 189, the greater the resultant loop voltage. 

e. It is apparent in B of figure 189, that the 
resultant voltage leads the voltage induced in leg 
No. 1 approximately 90° and lags the voltage 
induced in leg No. 2 by the same amount. 

/. Voltage Ei induced in the vertical sense an- 
tenna is intermediate in phase between the voltage 
induced in the two legs of the loop, and therefore 
lags the resultant loop voltage, E\, by 90°. To 
compensate for this phase difference (to have either 
an in-phase or an out-of -phase relation between 
resultant loop voltage E x and sense voltage Ez), it 
is necessary to advance or retard the phase of the 
loop voltage by 90° with a phase shifter. Re- 
tarded loop voltage Ej is shown in C of figure 189. 
If the loop voltage had been advanced, it would be 
shifted 180° in phase from that shown in C of 
figure 189. 

g. Notice that retarded loop voltage E 2 and 
sense voltage E 3) beginning at the same instant 
(ti), are in phase. These two voltages add in the 
input transformer; the receiver voltage E R is maxi- 
mum (E of fig. 189). 

h. If the antenna is rotated on its vertical axis 
through 180°, the electromagnetic wave strikes 
leg No. 2 before it strikes leg No. 1 (fig. 190). 



214 



sense 



OIRECTIOH Of 
WAVE TRAVEL 



LEG I 



LEO 2 




PHASE SHIFTER 




i. The voltages across both legs are induced in 
the same manner, producing a resultant again pro- 
portional to the separation between the legs. 
However, because of the loop rotation, the volt- 
ages of the two legs are interchanged, and resultant 
output voltage E x is shifted 180° in phase (B of 
fig. 190). 

j. Retarded loop voltage E R is, therefore, out of 
phase with the sense voltage, and minimum signal 
E is applied to the receiver (C, D, and E of fig. 
190). 

k. At intermediate points between the maximum 
and minimum positions of the loop, the following 
conditions exist. Assume that the transmitter 
azimuth is 0° as shown in figure 191. 



(1) When the loop is rotated from 0° to 90°, 
loop voltage gradually decreases (dis- 
tance between loop legs along the direc- 
tion of wave travel becomes less). Sense 
voltage is constant and in phase with the 
loop voltage. Resultant receiver volt- 
age is decreasing. 

(2) When the loop is rotated from 90° to 
180°, loop voltage gradually increases 
(distance between loop legs along the 
direction of wave travel becomes greater). 
Sense voltage is constant and 180° out 
of phase with the loop voltage. Result- 
ant receiver voltage decreases because of 
the out-of -phase condition. 



215 



SENSE 



DIRECTION OF 
WAVE TRAVEL 



LEG 2 



LEG I 



LOOP 









-90' 











ER 

It 



RECEIVER 
INPUT 



PHASE SHIFTER 



t,t.t, INDUCED 
* I VOLTAGE 
I LEG 2- 



INDUCED 
VOLTAGE 
LEG 




/RESULTANT 
S LOOP VOLTAGE 



LOOP VOLTAGES 



B 



-90* PHASE SHIFTER C 
E 2 



SENSE VOLTAGE 
E 3 



RECEIVER 
Er 



VOLTAGE E 
TM 666-193 



Figure 190. Relationship of voltages in loop and sense antenna system when 
loop has been rotated 180° from position shown in figure 189. 



(3) 



When the loop is rotated from 180° to 
270°, loop voltage gradually decreases. 
Sense voltage is constant and 180° out 
of phase with the loop voltage. Result- 
ant receiver voltage increases. 
(4) When the loop is rotated from 270° to 
360°, loop voltage gradually increases. 



Sense voltage is constant and in phase 
with the loop voltage. Resultant re- 
ceiver voltage increases. 
I. Sense Voltage. In practice, sense circuits sel- 
dom are adjusted to the ideal condition just de- 
scribed, and the resulting unidirectional pattern is 
not a perfect cardioid. If the sense voltage is too 



216 



TRANSMITTER 



DIRECTION OF 
WAVE TRAVEL 




SENSE PATTERN 
(ORCLE) 



270* * 



LOOP PATTERN 
(FIGURE 8) 



TL I3SZ3 



Figure 191. Cardioid response pattern. 

small, the resultant pattern is a slightly lopsided 
figure 8. Increasing the sense voltage then makes 
the figure 8 more and more lopsided (fig. 192), until 
the sense voltage equals the maximum loop volt- 
age; then one lobe disappears completely, making 
the pattern a perfect cardioid (fig. 191). Further 
increase of sense voltage increases both maximum 
and minimum of the resultant pattern (fig. 193), 
making it more and more like a circle. Either too 
little or too much sense voltage makes sense de- 
termination difficult, but any of the patterns just 
mentioned would be usable. The lopsidedness 
of the resultant pattern is readily distinguishable 
so long as the sense voltage is within ± 50 percent 
of the maximum loop voltage. If the sense volt- 
age is a little out of phase with the loop voltage, 
the resultant pattern becomes nearly circular, and 
the amplitude relation must be kept closer to the 
ideal for satisfactory operation. Figure 194 shows 
a case in which both amplitude and phase relation 
are far from the ideal. Here, the sense voltage has 
about half the amplitude shown in figure 192, and 
is 40° or 50° out of phase with the loop voltage. 
The lopsidedness of the resulting pattern could be 
detected by comparing its two maxima on a visual 
indicator, but the difference is too small to be 




LOOP PATTER N- 



SENSE PATTERN 



COMBINED 
LOOP AND 
SENSE 
PATTERN 



TL 15071 S 



Figure 192. Response pattern, low sense voltage and correct 
phase relation. 



SENSE 
PATTERI 



LOOP 
PATTERN 




COMBINED LOOP 
AND SENSE 
PATTERN 



TM 666-194 



Figure 19S. Response pattern, high sense voltage and correct 
phase relation. 



detected by listening. If necessary (for example, 
if part of the sense antenna is lost), such a pattern 
can be used by observing which way the null shifts 
when the sense switch is operated ; both nulls shift 
toward the small end of the lopsided figure 8 
pattern. 



217 




AZIMUTH INDICATED 



TRUE AZIMUTH 



TM 666-198 



Figure 194. Response pattern, low sense voltage and incorrect 
phase relation. 

114. Loop Construction 

a. Balance. The most important single factor 
in the physical construction of a loop is its sym- 
metry. When the loop is symmetrical physically, 
electrical balance is obtained, and if the balance is 
good enough, antenna effect (stray pick-up of sense 
voltage) is reduced to a minimum. Any conduct- 
ing material near the loop, such as the top of the 
radio receiver case, should be placed symmetri- 
cally; otherwise the loop might be balanced at 
some positions but unbalanced at others. In the 
case of a square loop, it is preferable to place a 
corner rather than a side at the bottom; this 
arrangement keeps the body of the loop farther 
away from the ground so that any irregularities 
there (including metal items worn by the operator) 
are less likely to affect the loop balance. 

b. Electrostatic Shield. Multiturn loops often 
are inclosed in an electrostatic shield. This is a 
metal case, or a film of metal on a case of some 
other material which surrounds the loop winding 
almost completely (fig. 195). At one point (the 
top) there is a gap in the metal, and opposite sides 
of the gap are insulated from each other to prevent 
the shield from forming a closed (short-circuited) 
loop. The shield is grounded near the loop termi- 
nals as far as possible from the insulated gap. As a 
vertical antenna, therefore, the shield is short-cir- 
cuited. Consider the voltages induced by a pass- 
ing radio wave as made up of two components, one 
corresponding to the desired loop voltage, the 
other to antenna effect. The former can produce 
very little current in the shield because of the in- 
sulated gap, but is fully effective in the loop which 
does have conductors crossing that gap. The 
latter component causes a flow of current over the 
shield to ground. Since the ground connection is a 
short circuit, this current produces in the shield 
counter electromotive force just equal to the origi- 




— TL 97I3A 

Figure 195. Shielded loop. 

nal induced voltage, and it induces in each loop 
conductor a voltage which very nearly cancels the 
original antenna-effect voltage induced there by 
the radio wave. The reduction ratio is substan- 
tially the same as the reactance/resistance ratio 
which the shield would have if connected as a loop ; 
since the Q is seldom less than 10, the residual 
antenna effect is seldom more than one-tenth the 
antenna effect with no shield. In addition to re- 
ducing antenna effect directly, the shield helps in- 
directly; unless very close to the gap in the shield, 
an external object cannot affect the capacitance 
from loop to ground and thus cannot change the 
capacitive balance of the loop. However, the 
shield affords no protection against inductive un- 
balance, such as might be caused by a scrap of 
metal placed too close to one side of the loop. The 
shield eliminates precipitation static of the type 
caused by electrically charged raindrops striking 
the antenna, and it provides mechanical protection 
for the loop. 

c. Loop Size. Except in some special vhf 
loops, which resemble groups of dipoles more than 
they do ordinary loops, the largest dimension of a 
loop antenna is usually a very small fraction of a 
wavelength (.1 wavelength at most). The voltage 
picked up by such a small loop is proportional to 
the total area inclosed by its turns; that is, to the 



218 



product of the number of turns by the average 
areas of each turn. For maximum pick-up, the 
turns should be as large as possible, subject to 
mechanical limitations, such as portability. The 
number of turns should be as large as possible, sub- 
ject to electrical limitations. Most receivers will 
not operate efficiently from a loop which is self- 
resonant at any point within the operating fre- 
quency range; consequently, the number of turns 
must be small enough to keep the natural resonance 
higher than the highest operating frequency. 




ANTENNA 
ROTATING 
HANDLE 

AZIMUTH DIAL 



TM 666 -199 



Figure 196. Simple Adcock antenna. 

115. Adcock Antenna 

a. General. An Adcock antenna (fig. 196) con- 
sists of two spaced vertical antennas connected in 
opposition. Theoretically, it responds only to the 
vertically polarized component of an incoming 
radio wave, and therefore is not subject to polari- 
zation error. In practice, some polarization error 
is caused by various imperfections, but usually 
much less than in a loop receiving the same signal. 
The Adcock antenna is preferable to a loop in 
radio D/F (direction finding) when medium- or 
high-frequency signals must be received at a point 
beyond ground-wave range. 




MAGNETIC COMPONENT Of 
VERTICALLY POLARlZEO 
WAVE 



TM 666-196 



Figure 197. Adcock antenna, effects of vertically polarized 

wave. 

b. Principle. The action of an Adcock antenna, 
as far as vertically polarized waves are concerned, 
is identical with that of the loop antenna. A re- 
sultant current in output coil L is proportional to 
the vector difference of the voltages induced in the 
vertical members, exactly as in the case of the loop. 
Horizontally polarized components of radio waves 
do not affect the antenna because of the absence of 
the upper and lower horizontal members, and be- 
cause the crossed arrangement of the center mem- 
bers effectively cancels the voltages induced in 
them. The response pattern of an Adcock an- 
tenna is the same figure 8 pattern typical of the 
loop antenna. Minimum and maximum response 
points are present in the Adcock pattern in the 
same respective positions as in the loop pattern. 
Thus, the directional properties of the Adcock and 
loop antennas are the same with respect to verti- 
cally polarized waves. The effect of various types 
of wave polarization on the Adcock circuit are as 
follows: 

(1) Vertical polarization. The horizontal mag- 
netic lines of the vertically polarized 
wave cut the two vertical antenna ele- 
ments. Induced currents, although they 
are induced in phase in the vertical 
elements, oppose each other in the an- 
tenna coupling coil, producing a resultant 
voltage which leads the radiation field by 
90°. If the wave strikes both vertical 
elements simultaneously (antenna ele- 
ments broadside to direction of wave 



219 



travel as shown in fig. 197), the resultant 
voltage is a minimum. At other angles 
of arrival, the resultant voltage is pro- 
portional to the separation between the 
two antenna elements along the direction 
of wave travel. Thus, the action of the 
Adcock antenna system is identical with 
the action of the loop system and can be 
used in conjunction with a sense antenna 
to obtain a unidirectional pattern. 

(2) Horizontal -polarization. As shown in 
figure 198, only the horizontal members 
(transmission lines) are in a position to 
respond to horizontally polarized waves. 
In a well designed radio direction finder, 
efficient use is made of shielding and 
balancing to prevent any voltages in- 
duced in the horizontal members from 
reaching the input tube of the receiver. 
The residue is small in comparison with 
the response of a loop under similar cir- 
cumstances, but has the same directivity 
pattern. 

(3) Elliptical polarization. Radio waves usu- 
ally contain both vertical and horizontal 
components of polarization which, in com- 
bination, produce an elliptically polarized 
wave. Although the vertical and hori- 
zontal components can be viewed as 
acting independently, their effect on the 
antenna results from the combined action 
of both components. Because the re- 
sponse of an Adcock antenna is relatively 



small for the horizontally polarized com- 
ponent, its polarization error is likewise 
smaller than that of a loop antenna. 
This is true as long as the vertically 
polarized component does not entirely dis- 
appear; when the vertically polarized 
component predominates, the Adcock 
antenna has scarcely any polarization 
error. 




TM666-I9T 

Figure 198. Adcock antenna, effects of horizontally polarized 

wave. 



220 



INDEX 



Paragraph Page 

Absorption of electromagnetic energy.. 29a; 31d 45,47 

Adcock antenna 115 219 

Angle: 

Critical 226 30 

Of incidence 226; 27d 30, 39 

Of tilt... 156 21 

Anisotropic radiator 53a 89 

Antenna: 

Adcock.... 115 219 

Basic theory 36 54 

Bent 94 174 

Beverage 88 148 

Coaxial 66 114 

Conical 67 115 

Directivity 82c; 84c 141, 145 

Feeder systems 47; 86 79, 146 

Folded-dipole 65 112 

Folded-top 75 128 

Gain 316; 82a; 84c 47, 

139, 145 

Grounded: 

Bent 74 126 

Counterpoises 73d 125 

Current and voltage distribu- 
tion 70 119 

Folded-top 75 128 

Ground rods 73c 123 

Hertz 69h 119 

Marconi. 69A 119 

Mast and tower radiators 77 130 

Polarization 71 120 

Quarter-wave 69 118 

Radial grounds., 736 123 

Radiation characteristics 72 120 

Top-loaded 76 128 

Types of grounds 73 122 

Whip . 79 133 

Ground-plane 78 132 

Half-wave 56a 97 

Harmonic 83c 142 

Height 13a;31e;58a 18, 

47,98 

Hertz 69A 119 

Image 57g 98 

Impedance 43c; 846 67, 143 

Length 426 65 

Loop x 111 211 

Marconi 69A 119 

Microwave 68 117 

Quarter- wave 69 118 

Receiving 88d 151 

Reciprocity 63 109 



Antenna — Continued Paragraph 

Resistance — 436; 59 

Resonance 43a 

Sense 112 

Single- wire 64 

Top-loaded 76 

V 89 

Vertical half-wave 58c 

Wave 88 

Whip 79 

Arrays: 

Adjusting 107 

Bidirectional- 956 

Broadside 1006; 102 

Bruce 104d 

Center-fed- 101a 

Collinear 836; 100a; 101 

Connected 95a 

Directivity 956; 99 

Driven 98 

End-fed half-wave 101a 

End-fire. 100c; 103 

Extended double-Zepp 104a 

Feeding 107 

Flat-top 104e 

Lazy-H 1046 

Multielement 94 

Multiparasitic 106 

Parasitic 95a 

Parasitic elements 105 

Parasitic reflector 105a 

Sterba curtain 104c 

Tuning... 107 

Types 95c 

Unidirectional 956 

Yagi 106a" 

Atmosphere, propagation in 9 

Atmosphere, radio 7 

Atoms 20a 

At tenuation 45d 

Atmospheric refraction 96 

Bazooka . 49a 

Beam width 55c 

Beverage antenna 88 

Broadside arrays 102 

Bruce array 104d 

Cardioid theory 113 

Center-fed half -wave antennas 50d 



Page 
66, 104 
66 
213 
110 
128 
151 
102 
148 
133 

206 
174 
180, 
187 
196 
182 
141, 
179, 182 
174 
174, 
178 
177 
182 
181, 191 
195 
206 
197 
195 
174 
205 
174 
198 
198 
196 
206 
174 
174 
205 
12 
10 
25 
69 
13 

81 
96 
148 
187 
196 

214 
82 



Paragraph Page 



Characteristic impedance 


45a; 


68 69 


45c; 466; 46/; 51* 


77 79 






87 


Characteristics, transmission-line 


45 


68 


Coaxial antennas 


56c; 66 


97, 114 


Coaxial lines 


46/ 


79 


Collinear arrays 


836; 100a; 101141, 179, 






182 


Communication, successful radio 


16 


1 


Conical antennas 


67 


115 


Counterpoise 


73d 


125 


CouDlinff line to antenna 


496 


82 


CouDlinir. transmitter-to-line 


48 


79 


Critical angle _ _ 


226; 27d 


30, 39 


Critical frequency 


22a; 27d 


28, 39 


Current and magneticfield.. 


38 


55 


Current, displacement 


3c 


4 


Current-fed antenna 


50d 


82 


Db power loss. 








21a 


27 


Dellimrer fade 


20d; 246 


25,34 


DeltA match 


51d; 646(4) 


85, 110 


Diff r*u*t.ion 




12, 14 


Dinole 


54a 


94 


RAHiAt.ion nAt.tAra 


54c 


96 


Dinnle fol HpH 


65c 


113 


DirA#»i.iAnAl tAntAnnftM* 

*SIA OvvlVUOl 'AUVCUUoO « 








956 


174 


RroAflsiHe aitavr 


1006; 102 180, 187 


Rpiif*p B.i*m.v 


104d 


196 


Collinear arrays 


836; 100a; 101141, 179, 






182 


Driven aitavr 


98 


177 


End-fire arrays 


100c; 103 181, 191 


Flat-toD arrav 


104e 


197 




1046 


195 


Multielement arrays 


94 


174 




95a 


174 


R.hninHi<* 


91 


158 




956 


174 


V 


....... 89 


151 


Directional gain 


84c 


145 


DirefttifinAl reeer*t,ioTi 


110 


211 


Director 


95a 


174 


Direct wava 


13a 


18 


Displacement current 


3c 


4 


Distance. trAnsmifurian 


27a 


137 


Distributed constants 


456 


69 


Diurnal variation 


236 


33 


Doublet, elementary 


54a 


194 


Radiation pattern 


546 


194 


Driven arrays 


98 


177 


Ducts, atmospheric 


18a 


23 


E field. 


4a 


7 


E layer 


216 


28 


E. SDoradic 


24a 


34 


Electric field 3c: 4a; 46; 5; 37a 4, 7, 8, 






54 


Combined electric and magnetic fields. 39 


56 


Intensity 


4a; 136 


4,20 


Magnetic component 


3a 


4 


Near and far 


3a 


4 



Parmrrapk Pft 

Electromagnetic field 4a; 36; 39 7, 54 

56 

Electromagnetic wave 4a 7 

Diffraction 9; 9c 12 

Direction 46 7 

Energy la 1 

Polarization 5; 62; 77 8,107 

130 

Reflection 9a; 226 12, 30 

Refraction 96; 17a; 226 13, 22, 

30 

Eleven-year variation of ionosphere 23d 33 

Electrostatic shield 1146 218 

Elements, types 95a 174 

Elliptical polarization 1156 219 

End effect 42d 66 

End-fed antenna 50e; 86 82, 146 

End-fire arrays 100c; 103 181,191 

Energy, electromagnetic la; 2a 1 

Energy, reflection 45e 69 

Extended double-Zepp 104a 195 

Extraordinary wave 28c 44 

Flayers.. 21c 28 

Fading... 32 48 

Far field 6c 10 

Feed, methods... 49 81 

Bazooka. 49a 81 

Long-wire antennas. 86; 88d 146, 151 

Feed: 

Tuned methods 50 82 

Center-fed 50d 82 

Current-fed 50d 82 

End-fed 50e 82 

Zepp-fed 50« 82 

Untuned methods 51 84 

Artificial-line matching... Slj 89 

Coaxial-line. — 51* 89 

Off-center 516 84 

Single-wire 516 84 

Twisted-pair 51c 85 

Two-wire, using delta match 51d 85 

Two-wire, using J match 51/ 86 

Two-wire, using Q match 51a 87 

Two-wire, using stub matching.. 51jr 86 

Two-wire, using T match 51e 86 

Feeder methods for collinear arrays lOld 186 

Feeder systems 47 79 

Field: 

EandH 4a 7 

Electric 3c; 4a; 46(1); 5; 37a 4,7, 

8,54 

Electric component 3a 4 

Electromagnetic 2; 3; 4; 36 1, 4, 

7,54 

Induction 2a; 3a; 46; 36 1, 4, 

7,54 

Intensity 4a; 6; 31c 7, 8, 47 

Direction 46 7 

Free space 58a 98 

Magnetic 386 56 

Radiation 2a; 3a; 46 1, 4,7 

Strength pattern.. 556; 58a 96,98 



222 



Paragraph Page 

Flat-top array 104e 197 

Flutter fading 24a* 35 

Flux lines'. 36; 37a 4, 54 

Folded dipole - 56c 97 

Free space 2d 3 

Field strength 58a 98 

Propagation formulas 2c 2 

Frequency: 

Critical 22a 28 

Classifications 8 12 

Characteristics of ground waves 16a 22 

Wave 2c 2 

Gain, antenna 316; 82a; 84c 47, 

139, 145 

Great circle path of radio wave 276 39 

Ground: 

Effects 57; 85 97,146 

Losses 606 105 

Rods 73c 123 

Screens 61 107 

Wave 12 17 

Frequency characteristics 16a 22 

Grounded antennas 69 118 

Ground-plane antennas 78 132 

Ground reflected wave 14a 20 

Grounds: 

Practical 60 105 

Types 73 122 

H field 4a 7 

Half-rhombic antenna 90 154 

Half-wave antenna 56 97 

Center-fed 50d 82 

Coaxial 56c; 66 97, 114 

Conical 56c; 67 97, 115 

Current distribution 41 59 

Current-fed _.- 50a 82 

Delta-matching section 51<f; 646 85, 110 

Dipole 54a; 54c 142, 145 

Effects of practical grounds 60 105 

Electrical length 426 65 

End-fed 50e 82 

Feed 49 81 

Folded dipole 56c; 65 97, 112 

Ground effects 57 97 

Ground screens 61 107 

Horizontal 58a; 586 98, 99 

Impedance 43c 67 

Methods of feed 49 81 

Microwave 68 117 

Length.. 426 65 

Polarization 62 107 

Radiation 3a 4 

Radiation resistance 436; 59a; 60d 66, 

104, 106 

Reflection factor 58o 98 

Resistance 436 66 

Resonance 43a 66 

Standing waves of voltage and current. 416 62 

Single-wire 56c; 64 97, 110 

Untuned methods of feed 51 84 

Vertical 58c 102 

'oltage distribution 41 59 





Paragraph 


Page 


Harmonic antenna... 


83a 


141 


Heaviside, Oliver 


19 


24 


Height: 






Effect . 


31e; 58a; 


47, 98, 




60e 


106 


Virtual 


22c 


31 


Horizontal half-wave antenna 


586 


99 


Horizontal polarization. . 


c 

- O 


Q 

s 


Image antenna. . . . 


57jp 


98 


Impedance: 






Antenna . . 


43c; 846 


44, 143 


Characteristic 


45a; 45c; 


68,69, 




466; 46/; 


77, 79, 




51h 


87 


Input and output 


45a; 45c 


68, 69 


Matching 


45i 


76 


Mutual 


97 


176 


Incidence, angle. 


226; 27a* 


30,39 


Induction field .. 


2a; 3a; 46 


1,4,7 


Insulated two-wire line . 


46c 


78 


Intensity, field.. 


4a; 136; 31c 


7, 20, 47 


Inversion, temperature 


176 


23 


Ionization 


206 


25 


Layers 


21 


27 




7c 


12 






Characteristics 


22 


28 


Formation 


20 


24 


General 


19 


24 


Irregular variations 


24 


34 


Layers 


21 


27 




25 


36 


Reflections 


26 


37 




23 


31 


Storms.. 


24c 


34 


Sudden disturbance 


246 


34 


Isotropic radiator 


... . 53a; 82a 


89, 139 




51/ 


QA 




19 


24 




tlh 




Length, electrical, of antenna 


426 


65 


Length, half- wave antenna 


426 


65 


Lobe 


53c; 99c 


92, 179 




82 


139 




88 


148 




89/ 


154 




82c; 84c 


141, 145 




85 


146 


Feeding 


86; 88e; 89c 


146, 






151, 152 




82a; 84c 


139, 145 




.. 826 


139 


General characteristics 


83 


141 




90 


154 




83a 


141 




82a 


139 




87 


148 


Radiation patterns — 


84d 


145 




88d 


151 




91 


158 




89 


151 




88 


148 



223 



Paragraph 

Loop antenna 115 

Loop construction 114 

Loops 40c 

Lowest useful frequency 29 

Magnetic component of electrical field 3c 

Magnetic field 46; 386 

Magnetic and electric field combined 39 

Magnetic intensity 4o 

Mast and tower radiators 77 

Matched lines 45j 

Matching, impedance 451 

Matching section, exponential 91k (2) 

Matching stubs 51g 

Matter 20a 

Maximum usable frequency 28 

Microwave antenna 68 

Mismatch 45/ 

Multielement arrays 94 

Mutual impedance 97 

Node 40c 

Noise figure 33c 

Noise, radio 336 

Nonresonant long- wire antennas. . 87 

Null 53c; 111a 

Off-center feed 516 

Open two-wire line 466 

Optimum working frequency 30 

Ordinary wave 28c 

Parasitic arrays 95 

Pattern: 

Polar.. 53c 

Radiation, ground affected 58 

Rectangular-coordinate 536 

Phasing 96 

Phasing stubs _. 966 

Polarization 5; 62; 71 

Elliptical 1156 

Grounded quarter- wave antenna 71 

Half -wave antenna.. 62a 

Horizontal 5; 62d; 111c; 1156 

Loop-antenna patterns. 1116; 111c 

Requirements for various fre- 
quencies 626 

Vertical 5; 62c; 1116; 1156 

Power, radiated. 66 

Propagation 7; 9; 12 

Characteristics of terrain 15a 

Line-of-sight 13a 

Sky-wave 26 

Velocity 2c; 42; 45wi 

Q match 51h 

Quarter-wave antenna. 69 

Radiation: 

Collinear arrays. 101c 

Dipole 54c 

Directional and nondirectional 546 



211 
218 
59 
45 

4 

7, 56 
56 
7 

130 
74 
76 

168 
86 
25 
43 

117 
74 

174 

176 

159 
50 
49 

148 
92,211 

84 
77 

46 
44 

174 

92 
98 
91 
175 
176 
8, 

107, 120 
219 
120 
107 
8, 109, 
212,219 
211,212 

108 
8, 108, 
211,219 
8 
10, 
12, 17 
21 
18 
37 

2, 65, 76 

87 
118 



188 
96 
94 



Radiation — Continued Paragraph 

Doublet 54b 

Field 2o; 3a 

Folded-dipole 65c 

Ground-affected 58 

Grounded quarter-wave antenna 72 

Horizontal half-wave antenna 586 

Patterns 53; 84d; 1026; 105/ 

Resistance 59; 60a"; 72c; 846; 105e 



Use 

Vertical half- wave antenna . 
Radio communication, elements- 
Radio noise 



55 

58c 

16 

336 

Reactance, antenna 846 

Received signal strength 31 

Receiving antennas 88d 

Reciprocity of antenna 63 

Reflection 9a; 226(1) 

Reflection factor 58a 

Reflector _ 68e; 95a(4) 

Refraction 96; 17a; 226 (1) 

Refractive index 9b (2) ; 22b (2) 

Resistance: 

Antenna 43b 

Radiation 59; 60a* 

Resonance, antenna 43a 

Resonant antenna 87 

Rhombic antennas 91 

Advantages 91b 

Directivity and gain 91e 

Design Qlk 

Feeding 91*; 

Ground effects 91m 

Lobe alinement 9 1Z 

Multiwire 91/ 

Resonant 91n 

Terminating devices i 91/ 

Scattered reflections 24a" 

Screens, ground 61 

Seasonal variation of ionosphere 23c 

Sense antenna 112 

Shielded pair 46d 

Signal reception, strength 31; 33 

Single- wire antenna 56c; 64 

Single-wire feed 516 

Single-wire line 46a 

Skip distance 226; 27 

Skip zone 27 

Sky wave _ 26 

Modes 27a 

Transmission paths. 27 

Space, free 2d 

Spacers 46b 

Spacing, collinear arrays... 101c(2) 

Sporadic-E layer 24a 

Spreaders. 466 

Standing-wave ratio 45/ 

Standing waves 40; 41a(2); 45g; 84b 



Page 
94 

1,4 

113 
98 
120 

99 

89, 145, 
187, 202 
104, 
106, 121, 
143, 201 
96 
102 
1 
49 
143 
47 
151 
109 
12,30 
98 

117, 174 
13, 22, 
30 
13, 31 



104, 106 
66 
148 
158 
159 
161 
165 
168 
171 
170 
168 
171 
161 

35 
107 
33 
213 
79 
47,49 
97, 110 
84 
76 
30,37 
37 
37 
37 
37 
8 

77 
185 
34 
77 
74 
57, 59, 
70, 143 



224 



Paragraph Pat* 

Sterba curtain array 104c 196 

Stratosphere 76 10 

Stubs, matching - 51? 86 

Sunspot cycle --- 20d(l);23d 26 

8un8pots, effect.. -- 20d(l) 26,33 

Surface wave — -- --- 15* 21 

T match 51« 86 

Temperature inversion 176 23 

Terminations, basic line -.- 45p 70 

Tilt: 

Angle 156; 906(5); 91fc(3) 21,155, 

165 

Wave 886 149 

Top-loaded antenna 76 128 

Transmission line: 

Attenuation 45a* 69 

Characteristic impedance 45c; 466(2); 69, 77, 

46/(2) 79 

Characteristics - 45 68 

Distributed constants 456 69 

Impedance matching — 45? 76 

Reflection of energy 45e 69 

Resonance 45fc 76 

Standing-wave ratio 45/ 74 

Terminations . - - 45 g 70 

Types — 46 76 

Velocity of propagation 45m 76 

Transmitter-to-line coupling. 48 79 

Troposphere 7o 10 

Abnormal effects 18 23 

Tropospheric propagation 18a 23 

Tropospheric refraction 17a 22 

Tropospheric-wave component 17 22 

Twenty-seven day variation of ionosphere. 23e 33 

Two-wire insulated line 46c 77 

Two-wire open line 466 78 

Unidirectional arrays 956 174 

V antenna 89 151 

Combination 89/ 154 



V antenna — Continued Pmugrapk 

Design 896 

Feeding 89c 

Obtuse angle 89c 

Unidirectional 89a" 

Velocity of propagation 2c; 42; 45m 

Vertical antenna 66; 69;77;7Ji; 79 

Half-wave 58c 

Vertical polarization l \; 886 

Virtual height 22c 

Voltage and current distribution on half- 
wave antenna 41 

Voltage and electric field 37 

Wave: 

Angle 906(5) ;91*(2) 

Antenna 88 

Front j.. 4 

Incident and reflected.. 406 

Motion 2 

Tilt 886 

Wavelength 2c 

Waves: 

Diffraction 9; 9c 

Direct 13 

Electromagnetic 2a 

Great circle path 276 

Ground 12a 

Propagation 7 

Reflection T 9a; 226(1) 

Refraction 96; 17a; 226(1) 

Sky 26 

Standing 40;41a(2); 45jr; 846 

Surface 12J; 15a 

Yagi 106d 

Zepp, extended double 104a 

Zepp-f ed antenna 50e 



Page 

151 
152 
152 
152 
2, 65, 76 
114, 118, 
130, 132, 
133 
102 
8, 149 
31 

54 
59 

155, 165 
148 
7 
57 
1 

149 
2 

12, 14 
18 
1 
39 
17 
10 
12,30 
13,22, 
30 
37 
57, 58, 
70. 143 
17,21 

205 

195 



-&U.S. GOVERNMENT PRINTING OFFICE: 1969-390-947/358 



225